MAX5974AETE+T - Maxim - Product.Live

_______________________________________________________________ Maxim Integrated Products 1

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,

or visit Maxim’s website at www.maxim-ic.com.

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

19-5331; Rev 2; 6/11

Ordering Information/Selector Guide

General Description

The MAX5974_ provide control for wide-input-voltage,

active-clamped, current-mode PWM, forward converters

in Power-over-Ethernet (PoE) powered device (PD) appli-

cations. The MAX5974A/MAX5974C are well-suited for

universal or telecom input range, while the MAX5974B/

MAX5974D also accommodate low input voltage down

to 10.5V.

The devices include several features to enhance supply

efficiency. The AUX driver recycles magnetizing cur-

rent instead of wasting it in a dissipative clamp circuit.

Programmable dead time between the AUX and main

driver allows for zero-voltage switching (ZVS). Under light-

load conditions, the devices reduce the switching fre-

quency (frequency foldback) to reduce switching losses.

The MAX5974A/MAX5974B feature unique circuitry to

achieve output regulation without using an optocoupler,

while the MAX5974C/MAX5974D utilize the traditional

optocoupler feedback method. An internal error amplifier

with a 1% reference is very useful in nonisolated design,

eliminating the need for an external shunt regulator.

The devices feature a unique feed-forward maximum

duty-cycle clamp that makes the maximum clamp volt-

age during transient conditions independent of the line

voltage, allowing the use of a power MOSFET with lower

breakdown voltage. The programmable frequency dither-

ing feature provides low-EMI, spread-spectrum operation.

The MAX5974_ are available in 16-pin TQFN-EP pack-

ages and are rated for operation over the -40°C to +85°C

temperature range.

Features

S Peak Current-Mode Control, Active-Clamped

Forward PWM Controller

S Regulation Without Optocoupler (MAX5974A/

MAX5974B)

S Internal 1% Error Amplifier

S 100kHz to 600kHz Programmable Q8% Switching

Frequency, Synchronization Up to 1.2MHz

S Programmable Frequency Dithering for Low-EMI,

Spread-Spectrum Operation

S Programmable Dead Time, PWM Soft-Start,

Current Slope Compensation

S Programmable Feed-Forward Maximum Duty-

Cycle Clamp, 80% Maximum Limit

S Frequency Foldback for High-Efficiency Light-

Load Operation

S Internal Bootstrap UVLO with Large Hysteresis

S 100µA (typ) Startup Supply Current

S Fast Cycle-by-Cycle Peak Current-Limit, 35ns

Typical Propagation Delay

S 115ns Current-Sense Internal Leading-Edge

Blanking

S Output Short-Circuit Protection with Hiccup Mode

S Reverse Current Limit to Prevent Transformer

Saturation Due to Reverse Current

S Internal 18V Zener Clamp on Supply Input

S 3mm x 3mm, Lead-Free, 16-Pin TQFN-EP

Applications

PoE IEEE® 802.3af/at Powered Devices

High-Power PD (Beyond the 802.3af/at Standard)

Active-Clamped Forward DC-DC Converters

IP Phones

Wireless Access Nodes

Security Cameras

Note: All devices are specified over the -40°C to +85°C operating temperature range.

+Denotes a lead(Pb)-free/RoHS-compliant package.

*EP = Exposed pad.

IEEE is a registered service mark of the Institute of Electrical and Electronics Engineers, Inc.

EVALUATION KIT

AVAILABLE

PART TOP MARK PIN-PACKAGE UVLO THRESHOLD (V) FEEDBACK MODE

MAX5974AETE+ +AHY 16 TQFN-EP* 16 Sample/Hold

MAX5974BETE+ +AHZ 16 TQFN-EP* 8.4 Sample/Hold

MAX5974CETE+ +AIA 16 TQFN-EP* 16 Continuously Connected

MAX5974DETE+ +AIB 16 TQFN-EP* 8.4 Continuously Connected

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional

operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute

maximum rating conditions for extended periods may affect device reliability.

IN to GND (VEN = 0V) ...........................................-0.3V to +26V

EN, NDRV, AUXDRV to GND .....................-0.3V to (VIN + 0.3V)

RT, DT, FFB, COMP, SS, DCLMP, DITHER/SYNC

to GND .................................................................-0.3V to +6V

FB to GND (MAX5974A/MAX5974B only) ..................-6V to +6V

FB to GND (MAX5974C/MAX5974D only) ..............-0.3V to +6V

CS, CSSC to GND ...................................................-0.8V to +6V

PGND to GND ......................................................-0.3V to +0.3V

Maximum Input/Output Current (continuous)

IN, NDRV, AUXDRV ......................................................100mA

NDRV, AUXDRV (pulsed for less than 100ns) .................. Q1A

Continuous Power Dissipation (TA = +70NC) (Note 1)

16-Pin TQFN (derate 20.8mW/NC above +70NC) .......1666mW

Operating Temperature Range .......................... -40NC to +85NC

Maximum Junction Temperature .....................................+150NC

Storage Temperature Range ............................ -65NC to +150NC

Lead Temperature (soldering, 10s) ................................+300NC

Soldering Temperature (reflow) ......................................+260NC

ELECTRICAL CHARACTERISTICS

(VIN = 12V (for MAX5974A/MAX5974C, bring VIN up to 17V for startup), VCS = VCSSC = VDITHER/SYNC = VFB = VFFB = VDCLMP =

VGND, VEN = +2V, NDRV = AUXDRV = SS = COMP = unconnected, RRT = 34.8kI, RDT = 25kI, CIN = 1FF, TA = -40NC to +85NC,

unless otherwise noted. Typical values are at TA = +25NC.) (Note 2)

ABSOLUTE MAXIMUM RATINGS

Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-

layer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial.

Junction-to-Ambient Thermal Resistance (BJA) ..............48NC/W

Junction-to-Case Thermal Resistance (BJC) .....................7NC/W

PACKAGE THERMAL CHARACTERISTICS (Note 1)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

UNDERVOLTAGE LOCKOUT/STARTUP (IN)

Bootstrap UVLO Wakeup Level VINUVR VIN rising

MAX5974A/

MAX5974C 15.4 16 16.5

MAX5974B/

MAX5974D 8 8.4 8.85

Bootstrap UVLO Shutdown

Level VINUVF VIN falling 6.65 7 7.35 V

IN Clamp Voltage VIN_CLAMP IIN = 2mA (sinking) 17 18.5 20 V

IN Supply Current in

Undervoltage Lockout ISTART

VIN = +15V (for MAX5974A/

MAX5974C);

VIN = +7.5V (for MAX5974B/MAX5974D),

when in bootstrap UVLO

100 150 FA

IN Supply Current After Startup ICVIN = +12V 1.8 3 mA

ENABLE (EN)

Enable Threshold VENR VEN rising 1.17 1.215 1.26 V

VENF VEN falling 1.09 1.14 1.19

Input Current IEN 1 FA

OSCILLATOR (RT)

RT Bias Voltage VRT 1.23 V

NDRV Switching Frequency

Range fSW 100 600 kHz

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

ELECTRICAL CHARACTERISTICS (continued)

(VIN = 12V (for MAX5974A/MAX5974C, bring VIN up to 17V for startup), VCS = VCSSC = VDITHER/SYNC = VFB = VFFB = VDCLMP =

VGND, VEN = +2V, NDRV = AUXDRV = SS = COMP = unconnected, RRT = 34.8kI, RDT = 25kI, CIN = 1FF, TA = -40NC to +85NC,

unless otherwise noted. Typical values are at TA = +25NC.) (Note 2)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

NDRV Switching Frequency

Accuracy -8 +8 %

Maximum Duty Cycle DMAX fSW = 250kHz 79 80 82 %

SYNCHRONIZATION (SYNC)

Synchronization Logic-High

Input VIH-SYNC 2.91 V

Synchronization Pulse Width 50 ns

Synchronization Frequency

Range fSYNCIN 1.1 x

fSW 2 x

fSW kHz

Maximum Duty Cycle During

Synchronization

DMAX x fSYNC/

fSW %

DITHERING RAMP GENERATOR (DITHER)

Charging Current VDITHER = 0V 45 50 55 FA

Discharging Current VDITHER = 2.2V 43 50 57 FA

Ramp’s High Trip Point 2 V

Ramp’s Low Trip Point 0.4 V

SOFT-START AND RESTART (SS)

Charging Current ISS-CH 9.5 10 10.5 FA

Discharging Current

ISS-D VSS = 2V, normal shutdown 0.65 1.34 2 mA

ISS-DH

(VEN < VENF or VIN < VINUVF),

VSS = 2V, hiccup mode discharge for

tRSTRT (Note 3)

1.6 2 2.4 FA

Discharge Threshold to Disable

Hiccup and Restart VSS-DTH 0.15 V

Minimum Restart Time During

Hiccup Mode tRSTRT-MIN 1024 Clock

Cycles

Normal Operating High Voltage VSS-HI 5 V

Duty-Cycle Control Range VSS-DMAX DMAX (typ) = (VSS-DMAX/2.43V) 0 2 V

DUTY-CYCLE CLAMP (DCLMP)

DCLMP Input Current IDCLMP VDCLMP = 0 to 5V -100 0 +100 nA

Duty-Cycle Control Range VDCLMP-R

VDCLMP = 0.5V 73 75.4 77.5

DMAX (typ) =

1 - (VDCLMP/2.43V)

VDCLMP = 1V 54 56 58

VDCLMP = 2V 14.7 16.5 18.3

NDRV DRIVER

Pulldown Impedance RNDRV-N INDRV (sinking) = 100mA 1.9 3.4 I

Pullup Impedance RNDRV-P INDRV (sourcing) = 50mA 4.7 8.3 I

Peak Sink Current 1 A

Peak Source Current 0.65 A

Fall Time tNDRV-F CNDRV = 1nF 14 ns

Rise Time tNDRV-R CNDRV = 1nF 27 ns

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

ELECTRICAL CHARACTERISTICS (continued)

(VIN = 12V (for MAX5974A/MAX5974C, bring VIN up to 17V for startup), VCS = VCSSC = VDITHER/SYNC = VFB = VFFB = VDCLMP =

VGND, VEN = +2V, NDRV = AUXDRV = SS = COMP = unconnected, RRT = 34.8kI, RDT = 25kI, CIN = 1FF, TA = -40NC to +85NC,

unless otherwise noted. Typical values are at TA = +25NC.) (Note 2)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

AUXDRV DRIVER

Pulldown Impedance RAUX-N IAUXDRV (sinking) = 50mA 4.3 7.7 I

Pullup Impedance RAUX-P IAUXDRV (sourcing) = 25mA 10.6 18.9 I

Peak Sink Current 0.5 A

Peak Source Current 0.3 A

Fall Time tAUX-F CAUXDRV = 1nF 24 ns

Rise Time tAUX-R CAUXDRV = 1nF 45 ns

DEAD-TIME PROGRAMMING (DT)

DT Bias Voltage VDT 1.215 V

NDRV to AUXDRV Delay

(Dead Time) tDT

From NDRV falling

to AUXDRV falling

RDT = 10kI40 ns

RDT = 100kI300 350 410

From AUXDRV rising

to NDRV rising

RDT = 10kI40 ns

RDT = 100kI310 360 420

CURRENT-LIMIT COMPARATOR (CS)

Cycle-by-Cycle Peak

Current-Limit Threshold VCS-PEAK 375 393 410 mV

Cycle-by-Cycle Reverse

Current-Limit Threshold VCS-REV Turns AUXDRV off for the remaining

cycle if reverse current limit is exceeded -118 -100 -88 mV

Current-Sense Blanking Time

for Reverse Current Limit

tCS-BLANK-

REV From AUXDRV falling edge 115 ns

Number of Consecutive Peak

Current-Limit Events to Hiccup NHICCUP 8 Events

Current-Sense Leading-Edge

Blanking Time tCS-BLANK From NDRV rising edge 115 ns

Propagation Delay from

Comparator Input to NDRV tPDCS

From CS rising (10mV overdrive) to

NDRV falling (excluding leading-edge

blanking)

35 ns

Minimum On-Time tON-MIN 100 150 200 ns

SLOPE COMPENSATION (CSSC)

Slope Compensation Current

Ramp Height Current ramp’s peak added to CSSC

input per switching cycle 47 52 58 FA

PWM COMPARATOR

Comparator Offset Voltage VPWM-OS VCOMP - VCSSC 1.35 1.7 2 V

Current-Sense Gain ACS-PWM DVCOMP/DVCSSC (Note 4) 3.1 3.33 3.6 V/V

Current-Sense Leading-Edge

Blanking Time tCSSC-BLANK From NDRV rising edge 115 ns

Comparator Propagation Delay tPWM Change in VCSSC = 10mV (including

internal leading-edge blanking) 150 ns

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

ELECTRICAL CHARACTERISTICS (continued)

(VIN = 12V (for MAX5974A/MAX5974C, bring VIN up to 17V for startup), VCS = VCSSC = VDITHER/SYNC = VFB = VFFB = VDCLMP =

VGND, VEN = +2V, NDRV = AUXDRV = SS = COMP = unconnected, RRT = 34.8kI, RDT = 25kI, CIN = 1FF, TA = -40NC to +85NC,

unless otherwise noted. Typical values are at TA = +25NC.) (Note 2)

Note 2: All devices are 100% production tested at TA = +25NC. Limits over temperature are guaranteed by design.

Note 3: See the Output Short-Circuit Protection with Hiccup Mode section.

Note 4: The parameter is measured at the trip point of latch with VFB = 0V. Gain is defined as DVCOMP/DVCSSC for 0.15V <

DVCSSC < 0.25V.

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

ERROR AMPLIFIER

FB Reference Voltage VREF VFB when ICOMP = 0,

VCOMP = 2.5V

MAX5974A/

MAX5974B 1.5 1.52 1.54

MAX5974C/

MAX5974D 1.202 1.215 1.227

FB Input Bias Current IFB VFB = 0 to 1.75V

MAX5974A/

MAX5974B -250 +250

MAX5974C/

MAX5974D -500 +100

Voltage Gain AEAMP 80 dB

Transconductance gM

MAX5974A/

MAX5974B 1.8 2.55 3.2

MAX5974C/

MAX5974D 1.8 2.66 3.5

Transconductance Bandwidth BW Open loop (typical gain

= 1) -3dB frequency

MAX5974A/

MAX5974B 2

MHz

MAX5974C/

MAX5974D 30

Source Current VFB = 1V, VCOMP = 2.5V 300 375 455 FA

Sink Current VFB = 1.75V, VCOMP = 1V 300 375 455 FA

FREQUENCY FOLDBACK (FFB)

VCSAVG-to-FFB Comparator

Gain 10 V/V

FFB Bias Current IFFB VFFB = 0V, VCS = 0V (not in FFB mode) 26 30 33 FA

NDRV Switching Frequency

During Foldback fSW-FB fSW/2 kHz

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Typical Operating Characteristics

(VIN = 12V (for MAX5974A/MAX5974C, bring VIN up to 17V for startup), VCS = VCSSC = VDITHER/SYNC = VFB = VFFB = VDCLMP =

VGND, VEN = 2V, NDRV = AUXDRV = SS = COMP = unconnected, RRT = 34.8kI, RDT = 25kI, unless otherwise noted.)

TEMPERATURE (°C)

IN UVLO WAKE-UP LEVEL (V)

603510-15

15.8

15.9

16.0

16.1

16.2

16.3

15.7

-40 85

IN UVLO WAKE-UP LEVEL

vs. TEMPERATURE

MAX5974A/B/C/D toc01

MAX5974A/MAX5974C

TEMPERATURE (°C)

IN UVLO WAKE-UP LEVEL (V)

603510-15

8.1

8.2

8.3

8.4

8.5

8.6

8.0

-40 85

IN UVLO WAKE-UP LEVEL

vs. TEMPERATURE

MAX5974A/B/C/D toc02

MAX5974B/MAX5974D

IN UVLO SHUTDOWN LEVEL

vs. TEMPERATURE

MAX5974A/B/C/D toc03

TEMPERATURE (°C)

IN UVLO SHUTDOWN LEVEL

603510-15

6.9

7.0

7.1

7.2

7.3

6.8

-40 85

EN RISING THRESHOLD

vs. TEMPERATURE

MAX5974A/B/C/D toc04

TEMPERATURE (°C)

EN RISING THRESHOLD (V)

603510-15

1.212

1.214

1.216

1.218

1.220

1.210

-40 85

EN FALLING THRESHOLD

vs. TEMEPRATURE

MAX5974A/B/C/D toc05

TEMPERATURE (°C)

EN FALLING THRESHOLD (V)

6035-15 10

1.143

1.144

1.145

1.146

1.148

1.147

1.149

1.150

1.142

-40 85

UVLO SHUTDOWN CURRENT

vs. TEMPERATURE

MAX5974A/B/C/D toc06

TEMPERATURE (°C)

UVLO CURRENT (µA)

603510-15

100

120

140

-40 85

MAX5974A/MAX5974C

MAX5974B/MAX5974D

SUPPLY CURRENT vs. SUPPLY VOLTAGE

(MAX5974A/MAX5974C)

MAX5974A/B/C/D toc07

TEMPERATURE (°C)

SUPPLY CURRENT (µA)

18161412108642

100

1000

10,000

0 20

TA = -40°C

TA = +85°C

SUPPLY CURRENT vs. SUPPLY VOLTAGE

(MAX5974B/MAX5974D)

MAX5974A/B/C/D toc08

TEMPERATURE (°C)

SUPPLY CURRENT (µA)

18161412108642

100

1000

10,000

0 20

TA = -40°C

TA = +85°C

SUPPLY CURRENT

vs. SWITCHING FREQUENCY

MAX5974A/B/C/D toc09

SWITCHING FREQUENCY (kHz)

SUPPLY CURRENT (mA)

700600500400300200100

0.4

0.8

1.2

1.6

2.0

2.4

0 800

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Typical Operating Characteristics (continued)

(VIN = 12V (for MAX5974A/MAX5974C, bring VIN up to 17V for startup), VCS = VCSSC = VDITHER/SYNC = VFB = VFFB = VDCLMP =

VGND, VEN = 2V, NDRV = AUXDRV = SS = COMP = unconnected, RRT = 34.8kI, RDT = 25kI, unless otherwise noted.)

SOFT-START CHARGING CURRENT

vs. TEMPERATURE

MAX5974A/B/C/D toc10

TEMPERATURE (°C)

SOFT-START CHARGING CURRENT (µA)

603510-15

9.98

9.99

10.00

10.01

10.02

10.03

10.04

10.05

10.06

9.97

-40 85

SWITCHING FREQUENCY

vs. RRT VALUE

MAX5974A/B/C/D toc11

RRT VALUE (kΩ)

SWITCHING FREQUENCY (kHz)

100

1000

10010

SWITCHING FREQUENCY

vs. TEMPERATURE

MAX5974A/B/C/D toc12

TEMPERATURE (°C)

SWITCHING FREQUENCY (kHz)

6035-15 10

245

246

247

248

250

249

251

252

244

-40 85

FREQUENCY DITHERING

vs. RDITHER

MAX5974A/B/C/D toc13

RDITHER (kΩ)

FREQUENCY DITHERING (%)

900800700600500400

300 1000

MAXIMUM DUTY CYCLE

vs. SWITCHING FREQUENCY

MAX5974A/B/C/D toc14

SWITCHING FREQUENCY (kHz)

MAXIMUM DUTY CYCLE (%)

700600100 200 300 400 500

0 800

MAXIMUM DUTY CYCLE

vs. TEMPERATURE

MAX5974A/B/C/D toc15

TEMPERATURE (°C)

MAXIMUM DUTY CYCLE (%)

6035-15 10

80.3

80.4

80.5

80.6

80.8

80.7

80.9

81.0

80.2

-40 85

MAXIMUM DUTY CYCLE

vs. SYNC FREQUENCY

MAX5974A/B/C/D toc16

SYNC FREQUENCY (kHz)

MAXIMUM DUTY CYCLE (%)

450400350300

250 500

VSS = 0.5V

MAXIMUM DUTY CYCLE

vs. VSS

MAX5974A/B/C/D toc17

VSS (V)

MAXIMUM DUTY CYCLE (%)

2.01.51.00.5

100

0 2.5

MAXIMUM DUTY CYCLE

vs. VDCLMP

MAX5974A/B/C/D toc18

VDCLMP (V)

MAXIMUM DUTY CYCLE (%)

2.01.51.00.5

100

0 2.5

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Typical Operating Characteristics (continued)

(VIN = 12V (for MAX5974A/MAX5974C, bring VIN up to 17V for startup), VCS = VCSSC = VDITHER/SYNC = VFB = VFFB = VDCLMP =

VGND, VEN = 2V, NDRV = AUXDRV = SS = COMP = unconnected, RRT = 34.8kI, RDT = 25kI, unless otherwise noted.)

DEAD TIME vs. RDT VALUE

MAX5974A/B/C/D toc19

RDT VALUE (kΩ)

DEAD TIME (ns)

908020 30 40 6050 70

100

150

200

250

300

350

400

10 100

DEAD TIME vs. TEMPERATURE

MAX5974A/B/C/D toc20

TEMPERATURE (°C)

DEAD TIME (ns)

11085603510-15

100

102

-40

PEAK CURRENT-LIMIT THRESHOLD

vs. TEMPERATURE

MAX5974A/B/C/D toc21

TEMPERATURE (°C)

PEAK CURRENT-LIMIT THRESHOLD (mV)

603510-15

389

390

391

392

393

394

395

396

397

398

388

-40 85

REVERSE CURRENT-LIMIT THRESHOLD

vs. TEMPERATURE

MAX5974A/B/C/D toc22

TEMPERATURE (°C)

REVERSE CURRENT-LIMIT THRESHOLD (mV)

603510-15

-106

-105

-104

-103

-102

-101

-100

-99

-98

-97

-107

-40 85

SLOPE COMPENSATION CURRENT

vs. TEMPERATURE

MAX5974A/B/C/D toc23

TEMPERATURE (°C)

SLOPE COMPENSATION CURRENT (mA)

6035-15 10

50.5

51.0

51.5

52.0

53.0

52.5

53.5

54.0

50.0

-40 85

NDRV MINIMUM ON-TIME

vs. TEMPERATURE

MAX5974A/B/C/D toc24

TEMPERATURE (°C)

NDRV MINIMUM ON-TIME (ns)

603510-15

145

150

155

160

165

170

140

-40 85

CURRENT-SENSE GAIN

vs. TEMPERATURE

MAX5974A/B/C/D toc25

TEMPERATURE (°C)

CURRENT-SENSE GAIN (V/V)

603510-15

3.31

3.32

3.33

3.34

3.35

3.36

3.37

3.38

3.39

3.40

3.30

-40 85

FEEDBACK VOLTAGE

vs. TEMPERATURE

MAX5974A/B/C/D toc26

TEMPERATURE (°C)

FEEDBACK VOLTAGE (V)

603510-15

1.211

1.212

1.213

1.214

1.215

1.216

1.217

1.218

1.219

1.220

1.210

-40 85

MAX5974C/MAX5974D

FEEDBACK VOLTAGE

vs. TEMPERATURE

MAX5974A/B/C/D toc27

TEMPERATURE (°C)

FEEDBACK VOLTAGE (V)

603510-15

1.517

1.518

1.519

1.520

1.521

1.522

1.516

-40 85

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Typical Operating Characteristics (continued)

(VIN = 12V (for MAX5974A/MAX5974C, bring VIN up to 17V for startup), VCS = VCSSC = VDITHER/SYNC = VFB = VFFB = VDCLMP =

VGND, VEN = 2V, NDRV = AUXDRV = SS = COMP = unconnected, RRT = 34.8kI, RDT = 25kI, unless otherwise noted.)

TRANSCONDUCTANCE

vs. TEMPERATURE

MAX5974A/B/C/D toc28

TEMPERATURE (°C)

TRANSCONDUCTANCE (mS)

603510-15

2.1

2.2

2.3

2.4

2.5

2.6

2.7

2.8

2.9

3.0

2.0

-40 85

MAX5974C/MAX5974D

MAX5974A/MAX5974B

TRANSCONDUCTANCE HISTOGRAM

(MAX5974A/MAX5974B)

MAX5974A/B/C/D toc29

TRANSCONDUCTANCE (mS)

N (%)

2.642.622.602.582.562.542.522.502.482.46

2.44

TRANSCONDUCTANCE HISTOGRAM

(MAX5974C/MAX5974D)

MAX5974A/B/C/D toc30

TRANSCONDUCTANCE (mS)

N (%)

2.762.742.722.702.682.662.642.622.602.58

2.56

ENABLE RESPONSE

MAX5974A/B/C/D toc31

VEN

5V/div

VNDRV

10V/div

VAUXDRV

10V/div

VOUT

5V/div

200µs/div

VSS RAMP RESPONSE

MAX5974A/B/C/D toc33

10µs/div

VSS

2V/div

VNDRV

10V/div

VAUXDRV

10V/div

SHUTDOWN RESPONSE

MAX5974A/B/C/D toc32

100µs/div

VEN

5V/div

VNDRV

10V/div

VAUXDRV

10V/div

VOUT

5V/div

VDCLMP RAMP RESPONSE

MAX5974A/B/C/D toc34

10µs/div

VDCLMP

2V/div

VNDRV

10V/div

VAUXDRV

10V/div

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Typical Operating Characteristics (continued)

(VIN = 12V (for MAX5974A/MAX5974C, bring VIN up to 17V for startup), VCS = VCSSC = VDITHER/SYNC = VFB = VFFB = VDCLMP =

VGND, VEN = 2V, NDRV = AUXDRV = SS = COMP = unconnected, RRT = 34.8kI, RDT = 25kI, unless otherwise noted.)

NDRV 10% TO 90% RISE TIME

MAX5974A/B/C/D toc35

10ns/div

VNDRV

2V/div

0ns

27.6ns

NDRV 90% TO 10% FALL TIME

MAX5974A/B/C/D toc36

10ns/div

VNDRV

2V/div

0ns

13.8ns

AUXDRV 10% TO 90% RISE TIME

MAX5974A/B/C/D toc37

10ns/div

VAUXDRV

2V/div

0ns

45.6ns

AUXDRV 90% TO 10% FALL TIME

MAX5974A/B/C/D toc38

10ns/div

VAUXDRV

2V/div

0ns

21ns

PEAK AUXDRV CURRENT

MAX5974A/B/C/D toc40

400ns/div

IAUXDRV

0.2A/div

PEAK SOURCE

CURRENT

PEAK SINK CURRENT

PEAK NDRV CURRENT

MAX5974A/B/C/D toc39

200ns/div

INDRV

0.5A/div

PEAK SOURCE CURRENT

PEAK SINK CURRENT

SHORT-CURRENT BEHAVIOR

MAX5974A/B/C/D toc41

VIN

5V/div

15V

VNDRV

10V/div

VCS

500mV/div

40ms/div

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Pin Configuration

Pin Description

FFB

PGND

AUXDRV

1 3

12 10 9

DCLMP

CSSC

GND

COMP

MAX5974A

MAX5974B

MAX5974C

MAX5974D

DITHER/

SYNC NDRV

THIN QFN

TOP VIEW

PIN NAME FUNCTION

1 DT

Dead-Time Programming Resistor Connection. Connect resistor RDT from DT to GND to set the

desired dead time between the NDRV and AUXDRV signals. See the Dead Time section to calculate

the resistor value for a particular dead time.

2DITHER/

SYNC

Frequency Dithering Programming or Synchronization Connection. For spread-spectrum frequency

operation, connect a capacitor from DITHER to GND and a resistor from DITHER to RT. To

synchronize the internal oscillator to the externally applied frequency, connect DITHER/SYNC to the

synchronization pulse.

3 RT

Switching Frequency Programming Resistor Connection. Connect resistor RRT from RT to GND to

set the PWM switching frequency. See the Oscillator/Switching Frequency section to calculate the

resistor value for the desired oscillator frequency.

4 FFB

Frequency Foldback Threshold Programming Input. Connect a resistor from FFB to GND to set the

output average current threshold below which the converter folds back the switching frequency to

1/2 of its original value. Connect to GND to disable frequency foldback.

5 COMP Transconductance Amplifier Output and PWM Comparator Input. COMP is level shifted down and

connected to the inverting input of the PWM comparator.

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Pin Description (continued)

PIN NAME FUNCTION

6 FB Transconductance Amplifier Inverting Input

7 GND Signal Ground

8 CSSC Current Sense with Slope Compensation Input. A resistor connected from CSSC to CS programs the

amount of slope compensation. See the Programmable Slope Compensation section.

9 CS Current-Sense Input. Current-sense connection for average current sense and cycle-by-cycle

current limit. Peak current-limit trip voltage is 400mV and reverse current-limit trip voltage is -100mV.

10 PGND Power Ground. PGND is the return path for gate-driver switching currents.

11 NDRV Main Switch Gate-Driver Output

12 AUXDRV pMOS Active Clamp Switch Gate-Driver Output. AUXDRV can also be used to drive a pulse

transformer for synchronous flyback application.

13 IN Converter Supply Input. IN has wide UVLO hysteresis, enabling the design of efficient power

supplies. See the Enable Input section to determine if an external zener diode is required at IN.

14 EN

Enable Input. The gate drivers are disabled and the device is in a low-power UVLO mode when the

voltage on EN is below VENF. When the voltage on EN is above VENR, the device checks for other

enable conditions. See the Enable Input section for more information about interfacing to EN.

15 DCLMP

Feed-Forward Maximum Duty-Cycle Clamp Programming Input. Connect a resistive divider between

the input supply voltage DCLMP and GND. The voltage at DCLMP sets the maximum duty cycle

(DMAX) of the converter inversely proportional to the input supply voltage, so that the MOSFET

remains protected during line transients.

16 SS

Soft-Start Programming Capacitor Connection. Connect a capacitor from SS to GND to program the

soft-start period. This capacitor also determines hiccup mode current-limit restart time. A resistor

from SS to GND can also be used to set the DMAX below 75%.

— EP Exposed Pad. Internally connected to GND. Connect to a large ground plane to maximize thermal

performance. Not intended as an electrical connection point.

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Block Diagrams

DRIVER

1A/-0.65A

DEAD-TIME

CONTROL

11NDRV

SS16

CS9

CSSC8

COMP5

FB6

IN13

EN14

PGND

DRIVER

0.5A/-0.3A

12AUXDRV

1DT

20% < DMAX < 80%

DCLMP

FFB

SYNC

30µA/

90µA

NDRV

BLANKING

PULSE

PGND

FFB COMP

VCSAVG

DEAD TIME

NDRV

AUXDRV

POK

OSCILLATOR

DRIVER LOGIC

REVERSE ILIM LIMIT TURNS

OFF AUX IMMEDIATELY

PEAK ILIM

COMP

400mV

DITHER/

SYNC

7GND

10PGND

50µA/

-50µA

2V/400mV

10X

QSET

QCLR

SCOUNT 8

EVENTS

HICCUP

LATCH

115ns

BLANKING

REGULATOR

ENABLE

1.215V

18V

POK

THERMAL

SHUTDOWN

UVLO

115ns

BLANKING

VSS < 150mV

REVERSE ILIM COMP

-100mV

2µA

10µA

2mA

POK

PWM

COMP

2 x R1

S/H

1.52V

VB SLOPE

COMPENSATION

LOW-POWER UVLO

VINUVR = 16V (MAX5974A)

VINUVR = 8.4V (MAX5974B)

VINUVF = 7V

MAX5974A

MAX5974B

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Block Diagrams (continued)

DRIVER

1A/-0.65A

DEAD-TIME

CONTROL

11NDRV

SS16

CS9

CSSC8

COMP5

FB6

IN13

EN14

PGND

DRIVER

0.5A/-0.3A

12AUXDRV

1DT

20% < DMAX < 80%

DCLMP

FFB

SYNC

30µA/

90µA

NDRV

BLANKING

PULSE

PGND

FFB COMP

VCSAVG

DEAD TIME

NDRV

AUXDRV

POK

OSCILLATOR

DRIVER LOGIC

REVERSE ILIM LIMIT TURNS

OFF AUX IMMEDIATELY

PEAK ILIM

COMP

400mV

DITHER/

SYNC

7GND

10PGND

50µA/

-50µA

2V/400mV

10X

QSET

QCLR

SCOUNT 8

EVENTS

HICCUP

LATCH

115ns

BLANKING

REGULATOR

ENABLE

1.215V

POK

THERMAL

SHUTDOWN

UVLO

115ns

BLANKING

VSS < 150mV

REVERSE ILIM COMP

-100mV

2µA

10µA

2mA

POK

PWM

COMP

2 x R1

1.215V

VB SLOPE

COMPENSATION

LOW-POWER UVLO

VINUVR = 16V (MAX5974C)

VINUVR = 8.4V (MAX5974D)

VINUVF = 7V

MAX5974C

MAX5974D

18V

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Detailed Description

The MAX5974A/MAX5974B/MAX5974C/MAX5974D are

optimized for controlling a 25W to 50W active-clamped,

self-driven synchronous rectification forward converter

in continuous-conduction mode. The main switch gate

driver (NDRV) and the active-clamped switch driver

(AUXDRV) are sized to optimize efficiency for 25W

design. The features-rich devices are ideal for PoE IEEE

802.3af/at-powered devices.

The MAX5974A/MAX5974C offer a 16V bootstrap UVLO

wake-up level with a 9V wide hysteresis. The low startup

and operating currents allow the use of a smaller storage

capacitor at the input without compromising startup and

hold times. The MAX5974A/MAX5974C are well-suited

for universal input (rectified 85V AC to 265V AC) or tele-

com (-36V DC to -72V DC) power supplies.

The MAX5974B/MAX5974D have a UVLO rising threshold

of 8.4V and can accommodate for low-input voltage (12V

DC to 24V DC) power sources such as wall adapters.

Power supplies designed with the MAX5974A/MAX5974C

use a high-value startup resistor, RIN, that charges a

reservoir capacitor, CIN (see the Typical Application

Circuits). During this initial period, while the voltage is

less than the internal bootstrap UVLO threshold, the

device typically consumes only 100FA of quiescent cur-

rent. This low startup current and the large bootstrap

UVLO hysteresis help to minimize the power dissipation

across RIN even at the high end of the universal AC input

voltage (265V AC).

Feed-forward maximum duty-cycle clamping detects chang-

es in line conditions and adjusts the maximum duty cycle

accordingly to eliminate the clamp voltage’s (i.e., the main

power FET’s drain voltage) dependence on the input voltage.

For EMI-sensitive applications, the programmable fre-

quency dithering feature allows up to Q10% variation in

the switching frequency. This spread-spectrum modula-

tion technique spreads the energy of switching harmon-

ics over a wider band while reducing their peaks, help-

ing to meet stringent EMI goals.

The devices include a cycle-by-cycle current limit

that turns off the main and AUX drivers whenever the

internally set threshold of 400mV is exceeded. Eight

consecutive occurrences of current-limit events trigger

hiccup mode, which protects external components by

halting switching for a period of time (tRSTRT) and allow-

ing the overload current to dissipate in the load and

body diode of the synchronous rectifier before soft-start

is reattempted.

The reverse current-limit feature of the devices turns

the AUX driver off for the remaining off period when

VCS exceeds the -100mV threshold. This protects the

transformer core from saturation due to excess reverse

current under some extreme transient conditions.

Current-Mode Control Loop

The advantages of current-mode control over voltage-

mode control are twofold. First, there is the feed-forward

characteristic brought on by the controller’s ability to adjust

for variations in the input voltage on a cycle-by-cycle basis.

Second, the stability requirements of the current-mode

controller are reduced to that of a single-pole system,

unlike the double pole in voltage-mode control.

The devices use a current-mode control loop where the

scaled output of the error amplifier (COMP) is compared

to a slope-compensated current-sense signal at CSSC.

Input Clamp

When the device is enabled, an internal 18V input clamp

is active. During an overvoltage condition, the clamp

prevents the voltage at the supply input IN from rising

above 18.5V (typ).

When the device is disabled, the input clamp circuitry is

also disabled.

Enable Input

The enable input is used to enable or disable the device.

Driving EN low disables the device. Note that the inter-

nal 18V input clamp is also disabled when EN is low.

Therefore, an external 18V zener diode is needed for

certain operating conditions as described below.

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

UVLO on Power Source

The enable input has an accurate threshold of 1.26V

(max). For applications that require a UVLO on the

power source, connect a resistive divider from the power

source to EN to GND as shown in Figure 1. A zener

diode between IN and PGND is required to prevent the

NDRV and AUXDRV gate-drive voltages from exceeding

20V, the maximum allowed gate voltage of power FETs.

The external zener diode should clamp in the following

range:

Z UVLO(MAX)

20V V V> >

where VZ is the zener voltage and VUVLO(MAX) is the

maximum wakeup level (16.5V or 8.85V depending on

the device version). An 18V zener diode is the best

choice.

Design the resistive divider by first selecting the value of

REN1 to be on the order of 100kω. Then calculate REN2

as follows:

EN(MAX)

EN2 EN1 S(UVLO) EN(MAX)

V R V V

= ×

where VEN(MAX) is the maximum enable threshold volt-

age and is equal to 1.26V and VS(UVLO) is the desired

UVLO threshold for the power source, below which the

device is disabled.

The digital output connected to EN should be capable

of withstanding more than the maximum supply voltage.

MCU Control of Enable Input

When using a microcontroller GPIO to control the enable

input, an 18V zener diode is required on IN as shown in

Figure 2.

High-Voltage Logic Control of Enable Input

In the case where EN is externally controlled by a high-

voltage open-drain/collector output (e.g., PGOOD indi-

cator of a powered device controller), connect IN to EN

through a resistor REN and connect EN to an open-drain

or open-collector output as shown in Figure 3. Select

REN such that the voltage at IN, when EN is low, is less

than 20V (i.e., the maximum gate voltage of the main and

AUX FETs):

S(MAX) EN IN

V 20V

R R

× <

where VS(MAX) is the maximum supply voltage. Obeying

this relationship eliminates the need for an external zener

diode.

The digital output connected to EN should be capable of

withstanding more than 20V.

Figure 1. Programmable UVLO for the Power Source Figure 2. MCU Control of the Enable Input

RIN

REN1

DIGITAL

CONTROL

18V

REN2

CIN

MAX5974

CIN

18V

RIN

MCU

I/O

MAX5974_

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Always-On Operation

For always-on operation, connect EN to IN as shown

in Figure 4. No external zener diode is needed for this

configuration.

Bootstrap Undervoltage Lockout

The devices have an internal bootstrap UVLO that is very

useful when designing high-voltage power supplies (see

the Block Diagrams). This allows the device to bootstrap

itself during initial power-up. The MAX5974A/MAX5974C

soft-start when VIN exceeds the bootstrap UVLO thresh-

old of VINUVR (16V typ).

Because the MAX5974B/MAX5974D are designed for

use with low-voltage power sources such as wall adapt-

ers outputting 12V to 24V, they have a lower UVLO

wake-up threshold of 8.4V.

Startup Operation

The device starts up when the voltage at IN exceeds 16V

(MAX5974A/MAX5974C) or 8.4V (MAX5974B/MAX5974D)

and the enable input voltage is greater than 1.26V.

During normal operation, the voltage at IN is nor-

mally derived from a tertiary winding of the transformer

(MAX5974C/MAX5974D). However, at startup there is

no energy being delivered through the transformer;

hence, a special bootstrap sequence is required. In the

Typical Application Circuits, CIN charges through the

startup resistor, RIN, to an intermediate voltage. Only

100FA of the current supplied through RIN is used by

the ICs, the remaining input current charges CIN until

VIN reaches the bootstrap UVLO wake-up level. Once

VIN exceeds this level, NDRV begins switching the

n-channel MOSFET and transfers energy to the second-

ary and tertiary outputs. If the voltage on the tertiary

output builds to higher than 7V (the bootstrap UVLO

shutdown level), then startup has been accomplished

and sustained operation commences. If VIN drops below

7V before startup is complete, the device goes back to

low-current UVLO. In this case, increase the value of CIN

in order to store enough energy to allow for the voltage

at the tertiary winding to build up.

While the MAX5974A/MAX5974B derive their input volt-

age from the coupled inductor output during normal

operation, the startup behavior is similar to that of the

MAX5974C/MAX5974D.

Soft-Start

A capacitor from SS to GND, CSS, programs the soft-

start time. VSS controls the oscillator duty cycle during

startup to provide a slow and smooth increase of the

duty cycle to its steady-state value. Calculate the value

of CSS as follows:

SS-CH SS

I t

C2V

where ISS-CH (10FA typ) is the current charging CSS dur-

ing soft-start and tSS is the programmed soft-start time.

A resistor can also be added from the SS pin to GND to

clamp VSS < 2V and, hence, program the maximum duty

cycle to be less than 80% (see the Duty-Cycle Clamping

section).

Figure 3. High-Voltage Logic Control of the Enable Input

Figure 4. Always-On Operation

RIN

CIN

MAX5974

DIGITAL

CONTROL

REN

CIN

RIN

MAX5974_

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

n-Channel MOSFET Gate Driver

The NDRV output drives an external n-channel MOSFET.

NDRV can source/sink in excess of 650mA/1000mA

peak current; therefore, select a MOSFET that yields

acceptable conduction and switching losses. The exter-

nal MOSFET used must be able to withstand the maxi-

mum clamp voltage.

p-Channel MOSFET Gate Driver

The AUXDRV output drives an external p-channel

MOSFET with the aid of a level shifter. The level shifter

consists of CAUX, RAUX, and D5 as shown in the Typical

Application Circuits. When AUXDRV is high, CAUX is

recharged through D5. When AUXDRV is low, the gate

of the p-channel MOSFET is pulled below the source by

the voltage stored on CAUX, turning on the pFET.

Add a zener diode between gate to source of the exter-

nal n-channel and p-channel MOSFETs after the gate

resistors to protect VGS from rising above its absolute

maximum rating during transient condition (see the

Typical Application Circuits).

Dead Time

Dead time between the main and AUX output edges allow

ZVS to occur, minimizing conduction losses and improv-

ing efficiency. The dead time (tDT) is applied to both

leading and trailing edges of the main and AUX outputs

as shown in Figure 5. Connect a resistor between DT and

GND to set tDT to any value between 40ns and 400ns:

DT DT

10k

R t

40ns

Ω

= ×

Oscillator/Switching Frequency

The ICs’ switching frequency is programmable between

100kHz and 600kHz with a resistor RRT connected

between RT and GND. Use the following formula to

determine the appropriate value of RRT needed to gen-

erate the desired output-switching frequency (fSW):

RT SW

8.7 10

where fSW is the desired switching frequency.

Figure 5. Dead Time Between AUXDRV and NDRV

BLANKING, tBLK

NDRV

AUXDRV

DEAD TIME, tDT

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Figure 6. Hiccup Mode Timing Diagram

Peak Current Limit

The current-sense resistor (RCS in the Typical

Application Circuits), connected between the source

of the n-channel MOSFET and PGND, sets the current

limit. The current-limit comparator has a voltage trip level

(VCS-PEAK) of 400mV. Use the following equation to cal-

culate the value of RCS:

CS PRI

400mV

where IPRI is the peak current in the primary side of

the transformer, which also flows through the MOSFET.

When the voltage produced by this current (through the

current-sense resistor) exceeds the current-limit com-

parator threshold, the MOSFET driver (NDRV) terminates

the current on-cycle, within 35ns (typ).

The devices implement 115ns of leading-edge blanking

to ignore leading-edge current spikes. These spikes

are caused by reflected secondary currents, current-

discharging capacitance at the FET’s drain, and gate-

charging current. Use a small RC network for additional

filtering of the leading-edge spike on the sense wave-

form when needed. Set the corner frequency between

10MHz and 20MHz.

After the leading-edge blanking time, the device moni-

tors VCS for any breaches of the peak current limit of

400mV. The duty cycle is terminated immediately when

VCS exceeds 400mV.

Reverse Current Limit

The devices protect the transformer against saturation

due to reverse current by monitoring the voltage across

RCS while the AUX output is low and the p-channel FET

is on.

Output Short-Circuit Protection

with Hiccup Mode

When the device detects eight consecutive peak current-

limit events, both NDRV and AUXDRV driver outputs are

turned off for a restart period, tRSTRT. After tRSTRT, the

device undergoes soft-start. The duration of the restart

period depends on the value of the capacitor at SS (CSS).

During this period, CSS is discharged with a pulldown cur-

rent of ISS-DH (2FA typ). Once its voltage reaches 0.15V,

the restart period ends and the device initiates a soft-start

sequence. An internal counter ensures that the minimum

restart period (tRSTRT-MIN) is 1024 clock cycles when the

time required for CSS to discharge to 0.15V is less than

1024 clock cycles. Figure 6 shows the behavior of the

device prior and during hiccup mode.

Frequency Foldback for High-Efficiency

Light-Load Operation

The frequency foldback threshold can be programmed

from 0 to 20% of the full load current using a resistor from

FFB to GND.

VCSBL

(BLANKED CS

VOLTAGE)

VCS-PEAK

(400mV)

HICCUP

DISCHARGE WITH ISS-DH

VSS-DTH

SOFT-START

VOLTAGE,

VSS

VSS-HI

tRSTRT

tSS

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

When VCSAVG falls below VFFB, the device folds back

the switching frequency to 1/2 the original value to

reduce switching losses and increase the converter effi-

ciency. Calculate the value of RFFB as follows:

LOAD(LIGHT) CS

FFB FFB

10 I R

× ×

where RFFB is the resistor between FFB and GND,

ILOAD(LIGHT) is the current at light-load conditions that

triggers frequency foldback, RCS is the value of the

sense resistor connected between CS and PGND, and

IFFB is the current sourced from FFB to RFFB (30FA typ).

Duty-Cycle Clamping

The maximum duty cycle is determined by the lowest

of three voltages: 2V, the voltage at SS (VSS), and the

voltage (2.43V - VDCLMP). The maximum duty cycle is

calculated as:

MIN

MAX

D2.43V

where VMIN = minimum (2V, VSS, 2.43V - VDCLMP).

By connecting a resistor between SS and ground, the

voltage at SS can be made to be lower than 2V. VSS is

calculated as follows:

SS SS SS-CH

V R I= ×

where RSS is the resistor connected between SS and

GND, and ISS-CH is the current sourced from SS to RSS

(10FA typ).

DCLMP

To set DMAX using supply voltage feed-forward, connect

a resistive divider between the supply voltage, DCLMP,

and GND as shown in the Typical Application Circuits.

This feed-forward duty-cycle clamp ensures that the

external n-channel MOSFET is not stressed during sup-

ply transients. VDCLMP is calculated as follows:

DCLMP2

DCLMP S

DCLMP1 DCLMP2

V V

R R

= ×

where RDCLMP1 and RDCLMP2 are the resistive divider

values shown in the Typical Application Circuits and VS

is the input supply voltage.

Oscillator Synchronization

The internal oscillator can be synchronized to an external

clock by applying the clock to DITHER/SYNC directly. The

external clock frequency can be set anywhere between

1.1x to 2x the internal clock frequency.

Using an external clock increases the maximum duty

cycle by a factor equal to fSYNC/fSW. This factor should

be accounted for in setting the maximum duty cycle

using any of the methods described in the Duty-Cycle

Clamping section. The formula below shows how the

maximum duty cycle is affected by the external clock

frequency:

SYNC

MIN

MAX SW

D2.43V f

= ×

where VMIN is described in the Duty-Cycle Clamping

section, fSW is the switching frequency as set by the

resistor connected between RT and GND, and fSYNC is

the external clock frequency.

Frequency Dithering for Spread-

Spectrum Applications (Low EMI)

The switching frequency of the converter can be dith-

ered in a range of Q10% by connecting a capaci-

tor from DITHER/SYNC to GND, and a resistor from

DITHER/SYNC to RT as shown in the Typical Application

Circuits. This results in lower EMI.

A current source at DITHER/SYNC charges the capaci-

tor CDITHER to 2V at 50FA. Upon reaching this trip point,

it discharges CDITHER to 0.4V at 50FA. The charging

and discharging of the capacitor generates a triangular

waveform on DITHER/SYNC with peak levels at 0.4V and

2V and a frequency that is equal to:

TRI DITHER

50 A

fC 3.2V

=×

Typically, fTRI should be set close to 1kHz. The resistor

RDITHER connected from DITHER/SYNC to RT deter-

mines the amount of dither as follows:

DITHER

%DITHER 3 R

= ×

where %DITHER is the amount of dither expressed as a

percentage of the switching frequency. Setting RDITHER

to 10 x RRT generates Q10% dither.

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Programmable Slope Compensation

The device generates a current ramp at CSSC such that

its peak is 50FA at 80% duty cycle of the oscillator. An

external resistor connected from CSSC to the CS then

converts this current ramp into programmable slope-

compensation amplitude, which is added to the current-

sense signal for stability of the peak current-mode

control loop. The ramp rate of the slope compensation

signal is given by:

CSSC SW

R 50 A f

m80%

× µ ×

where m is the ramp rate of the slope-compensation

signal, RCSSC is the value of the resistor connected

between CSSC and CS used to program the ramp rate,

and fSW is the switching frequency.

Error Amplifier

The MAX5974A/MAX5974B include an internal error

amplifier with a sample-and-hold input. The feedback

input of the MAX5974C/MAX5974D is continuously con-

nected. The noninverting input of the error amplifier is

connected to the internal reference and feedback is

provided at the inverting input. High open-loop gain and

unity-gain bandwidth allow good closed-loop bandwidth

and transient response. Calculate the power-supply out-

put voltage using the following equation:

FB1 FB2

OUT REF FB2

R R

V V R

= ×

where VREF = 1.52V for the MAX5974A/MAX5974B

and VREF = 1.215V for the MAX5974C/MAX5974D. The

amplifier’s noninverting input is internally connected to

a soft-start circuit that gradually increases the reference

voltage during startup. This forces the output voltage to

come up in an orderly and well-defined manner under

all load conditions.

Applications Information

Startup Time Considerations

The bypass capacitor at IN, CIN, supplies current

immediately after the devices wake up (see the Typical

Application Circuits). Large values of CIN increase

the startup time, but also supply gate charge for more

cycles during initial startup. If the value of CIN is too

small, VIN drops below 7V because NDRV does not have

enough time to switch and build up sufficient voltage

across the tertiary output (MAX5974C/MAX5974D) or

coupled inductor output (MAX5974A/MAX5974B), which

powers the device. The device goes back into UVLO

and does not start. Use a low-leakage capacitor for CIN.

Typically, offline power supplies keep startup times to

less than 500ms even in low-line conditions (85V AC

input for universal offline or 36V DC for telecom applica-

tions). Size the startup resistor, RIN, to supply both the

maximum startup bias of the device (150FA) and the

charging current for CIN. CIN must be charged to 16V

within the desired 500ms time period. CIN must store

enough charge to deliver current to the device for at

least the soft-start time (tSS) set by CSS. To calculate the

approximate amount of capacitance required, use the

following formula:

G GTOT SW

IN G SS

IN HYST

I Q f

(I I )(t )

where IIN is the ICs’ internal supply current (1.8mA)

after startup, QGTOT is the total gate charge for the

n-channel and p-channel FETs, fSW is the ICs’ switch-

ing frequency, VHYST is the bootstrap UVLO hysteresis

(9V typ), and tSS is the soft-start time. RIN is then cal-

culated as follows:

S(MIN) INUVR

IN START

V V

−

≅

where VS(MIN) is the minimum input supply voltage for

the application (36V for telecom), VINUVR is the boot-

strap UVLO wake-up level (16V), and ISTART is the IN

supply current at startup (150FA max).

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Choose a higher value for RIN than the one calculated

above if a longer startup time can be tolerated in order

to minimize power loss on this resistor.

Active Clamp Circuit

Traditional clamp circuits prevent transformer saturation

by channeling the magnetizing current (IM) of the trans-

former onto a dissipative RC network. To improve effi-

ciency, the active clamp circuit recycles IM between the

magnetizing inductance and clamp capacitor. VCLAMP

is given by:

CLAMP

V1 D

−

where VS is the voltage of the power source and D is

the duty cycle. To select n-channel and p-channel FETs

with adequate breakdown voltages, use the maximum

value of VCLAMP. VCLAMP(MAX) occurs when the input

voltage is at its minimum and the duty cycle is at its

maximum. VCLAMP(MAX-NORMAL) during normal opera-

tion is therefore:

S(MIN)

CLAMP(MAX-NORMAL) P O

S S(MIN)

VN V

1N V

−

=×

where VS(MIN) is the minimum voltage of the power

source, NP/NS is the primary to secondary turns ratio,

and VO is the output voltage. The clamp capacitor,

n-channel, and p-channel FETs must have breakdown

voltages exceeding this level.

If feed-forward maximum duty-cycle clamp is used then:

DCLMP

MIN

MAX-FF

S DCLMP2

DCLMP1 DCLMP2

D 1

2.43 2.43

V R

12.43 R R

 

= = −

 

 

 

= − ×

 

 

Therefore, VCLAMP(MAX-FF) during feed-forward maxi-

mum duty clamp is:

( )

CLAMP(MAX-FF) MAX FF

DCLMP1 DCLMP2

DCLMP2

V1 D

2.43 R R

−

=−

× +

The AUX driver controls the p-channel FET through a

level shifter. The level shifter consists of an RC network

(formed by CAUX and RAUX) and diode D5, as shown in

the Typical Application Circuits. Choose RAUX and CAUX

so that the time constant exceeds 100/fSW. Diode D5 is a

small-signal diode with a voltage rating exceeding 25V.

Additionally, CCLAMP should be chosen such that the

complex poles formed with magnetizing inductance

(LMAG) and CCLAMP are 2x to 4x away from the loop

bandwidth:

MAG CLAMP

1-D 3 f

2 L C > ×

π ×

Bias Circuit

Optocoupler Feedback (MAX5974C/MAX5974D)

An in-phase tertiary winding is needed to power the bias

circuit when using optocoupler feedback. The voltage

across the tertiary VT during the on-time is:

TOUT S

V V N

= ×

where VOUT is the output voltage and NT/NS is the turns

ratio from the tertiary to the secondary winding. Select the

turns ratio so that VT is above the UVLO shutdown level

(7.35V max) by a margin determined by the holdup time

needed to “ride through” a brownout.

Coupled-Inductor Feedback (MAX5974A/MAX5974B)

When using coupled-inductor feedback, the power for

the devices can be taken from the coupled inductor dur-

ing the off-time. The voltage across the coupled induc-

tor, VCOUPLED, during the off-time is:

COUPLED OUT O

V V N

= ×

where VOUT is the output voltage and NC/NO is the

turns ratio from the coupled output to the main output

winding. Select the turns ratio so that VCOUPLED is

above the UVLO shutdown level (7.5V max) by a margin

determined by the holdup time needed to “ride through”

a brownout.

This voltage appears at the input of the devices, less

a diode drop. An RC network consisting of RSNUB and

CSNUB is for damping the reverse recovery transients of

diode D6.

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

During on-time, the coupled output is:

S C

COUPLED-ON OUT

SP O

N N

V (V V )

N N

−= − ×

where VS is the input supply voltage.

Care must be taken to ensure that the voltage at FB

(equal to VCOUPLED-ON attenuated by the feedback

resistive divider) is not more than 5V:

( )

FB2

FB-ON COUPLED-ON FB1 FB2

V V 5V

R R

= × <

If this condition is not met, a signal diode should be

placed from GND (anode) to FB (cathode).

Layout Recommendations

Typically, there are two sources of noise emission in a

switching power supply: high di/dt loops and high dV/dt

surfaces. For example, traces that carry the drain current

often form high di/dt loops. Similarly, the heatsink of the

main MOSFET presents a dV/dt source; therefore, mini-

mize the surface area of the MOSFET heatsink as much

as possible. Keep all PCB traces carrying switching cur-

rents as short as possible to minimize current loops. Use

a ground plane for best results.

For universal AC input design, follow all applicable

safety regulations. Offline power supplies can require

UL, VDE, and other similar agency approvals.

Refer to the MAX5974A and MAX5974C Evaluation Kit

data sheets for recommended layout and component

values.

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Typical Application Circuits

DCLMP

DITHER/

SYNC

CIN

1µF

25V

330pF

CCOMP2

6.8pF

RIN

100kI

6.8µH

COUT1 COUT2 COUT3 COUT4

COUT5

0.1µF

CBULK

33µF

RDCLMP1

30.1kI

PGOOD

ROPTO2

1kI

RDCLMP2

750I 1%

RDT

16.9kI 1%

CSS

0.1µF

3.3mH

36V TO 57V

CDITHER

10nF

FFB

COMP

AUXDRV

CSSC

GND

RFFB

10.0kI 1%

RG1

121kI 1%

RG2

200kI 1%

RCSSC

4.02kI 1%

RAUX

10kI

RCS

0.2I

RGATE3

10I

RGATE4

10I

RGATE1

10I

RGATE2

10I

RFB1

7.5kI

5V, 5A

RFB2

2.49kI

FDS3692

5i412DP

FOD817CSD

IRF6217

NPNS

CAUX

47nF

CINT

0.1µF

TLV4314AIDBVT-1.24V

CCLAMP

47nF

CCOMP1

2.2nF

ROPTO1

825I

RBIAS

4.02kI

RRT

14.7kI 1%

499I 1%

NDRV

PGND

5i412DP

RCOMP2

2.00kI

ROPTO3

4.99kI

1% RCOMP2

499I

MAX5974C

MAX5974D

(OPTOCOUPLER

FEEDBACK)

REN

100kI

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Typical Application Circuits (continued)

DCLMP

DITHER/

SYNC

CIN

1µF

25V

RIN

100kI

COUT1 COUT2 COUT3 COUT4 COUT5

0.1µF

CBULK

33µF

63V

RDCLMP1

30.1kI

RDCLMP2

750I 1%

RDT

16.9kI 1%

CSS

0.1µF

TO FB

5V, 5A

4 x 47µF

6.3V

LCOUPLED

36V TO 57V

CDITHER

10nF

CINT

47nF

CCOMP

4.7nF

FFB

COMP

AUXDRV

CSSC

GND

RFFB

10kI 1%

2kI 1%

RCSSC

4.02kI 1%

RAUX

10kI

RCS

0.2I

RGATE3

10I

RGATE4

10I

RGATE1

10I

RGATE2

10I

RSNUB

69.8I 1%

RFB1

54.9kI 1%

RFB2

10kI 1%

CSNUB

10pF

FDS3692

5i412DP

IRF6217

NPNS

CAUX

47nF

CCLAMP

47nF

RRT

14.7kI 1%

NDRV

PGND

5i412DP

MAX5974A

MAX5974B

(COUPLED INDUCTOR

FEEDBACK)

330pF

499I 1%

PGOOD

REN

100kI

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Typical Application Circuits (continued)

Chip Information

PROCESS: BiCMOS

Package Information

For the latest package outline information and land patterns

(footprints), go to www.maxim-ic.com/packages. Note that a

“+”, “#”, or “-” in the package code indicates RoHS status only.

Package drawings may show a different suffix character, but

the drawing pertains to the package regardless of RoHS status.

PACKAGE

TYPE

PACKAGE

CODE

OUTLINE

NO.

LAND

PATTERN NO.

16 TQFN-EP T1633+4 21-0136 90-0031

DCLMP

DITHER/

SYNC

CIN

RIN

COUT1 COUT2 COUT3 COUT4

CBULK

RDCLMP1

RDCLMP2

RDT

RDITHER

CSS

CDITHER

FFB

COMP

CCOMP CHF

AUXDRV

CSSC

GND

RFFB

RCSSC

RAUX

RCS

RGATE3

RGATE4

RGATE1

RGATE2

CAUX

CCLAMP

RRT

NDRV

PGND

MAX5974C

MAX5974D

RFB1

RFB2

REN

100kI

PGOOD

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied.

Maxim reserves the right to change the circuitry and specifications without notice at any time.

Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 27

MAX5974A/MAX5974B/MAX5974C/MAX5974D

Active-Clamped, Spread-Spectrum,

Current-Mode PWM Controllers

Revision History

REVISION

NUMBER

REVISION

DATE DESCRIPTION PAGES

CHANGED

0 6/10 Initial release —

1 9/10

Introduced the MAX5974B/MAX5974D. Updated the Absolute Maximum

Ratings, Electrical Characteristics, Pin Description, the p-Channel MOSFET

Gate Driver, Frequency Foldback for High-Efficiency Light-Load Operation

sections, and Typical Application Circuits.

1, 2, 3, 12, 15, 17,

19, 21, 23, 24, 25

2 6/11 Added internal zener diode information 1–10, 12–17, 19–25