Sample & Buy Product Folder Support & Community Tools & Software Technical Documents LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 LM2734x and LM2734x-Q1 2-MHz, 1.5-A or 2-A, Wide Input Range, Step-Down, DC-DC Regulator With Frequency Synchronization 1 Features 3 Description * The LM2734x and LM2734x-Q1 regulators are monolithic, high-frequency, PWM step-down DC-DC converters in 10-pin WSON and 10-pin MSOPPowerPAD packages. They contain all the active functions to provide local DC-DC conversion with fast transient response and accurate regulation in the smallest possible PCB area. 1 * * * * * * * * * * * * 2 * * * * * * * Space-Saving, 3 mm x 3 mm 10-Pin WSON and MSOP-PowerPAD Packages Wide Input Voltage Range: 3 V to 20 V Wide Output Voltage Range: 1 V to 18 V LM27341 Delivers 1.5-A Maximum Output Current LM27342 Delivers 2-A Maximum Output Current High Switching Frequency: 2 MHz Frequency Synchronization: 1 MHz < fSW < 2.35 MHz 150-m NMOS Switch With Internal Bootstrap Supply 70-nA Shutdown Current Internal Voltage Reference Accuracy of 1% Peak Current-Mode, PWM Operation Thermal Shutdown LM27341-Q1 and LM27342-Q1 are AEC-Q100 Grade 1 Qualified and Manufactured on an Automotive Grade Flow With a minimum of external components, the LM2734x and LM2734x-Q1 are easy to use. The ability to drive 1.5-A or 2-A loads respectively, with an internal 150-m NMOS switch results in the best power density available. The world-class control circuitry allows for on-times as low as 65 ns, thus supporting exceptionally high frequency conversion. Switching frequency is internally set to 2 MHz and synchronizable from 1 to 2.35 MHz, which allows the use of extremely small surface mount inductors and chip capacitors. Even though the operating frequency is very high, efficiencies up to 90% are easy to achieve. External shutdown is included, which features an ultra-low shutdown current of 70 nA. The LM2734x and LM2734x-Q1 use peak current-mode control and internal compensation to provide highperformance regulation over a wide range of operating conditions. Additional features include internal soft-start circuitry to reduce inrush current, pulse-by-pulse current limit, thermal shutdown, and output overvoltage protection. Applications Local 12-V to Vcore Step-Down Converters Radio Power Supply Core Power in HDDs Set-Top Boxes Automotive USB Powered Devices DSL Modems Device Information(1) PART NUMBER PACKAGE BODY SIZE (NOM) MSOP-PowerPAD (10) 4.90 mm x 3.00 mm LM2734x LM2734x-Q1 WSON (10) 3.00 mm x 3.00 mm (1) For all available packages, see the orderable addendum at the end of the data sheet. Typical Application Circuit VIN PVIN BOOST C2 AVIN L1 C1 VOUT SW C3 D1 LM27341/2 ON OFF EN R1 CLK SYNC GND/DAP FB R2 Copyright (c) 2016, Texas Instruments Incorporated 1 An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications, intellectual property matters and other important disclaimers. PRODUCTION DATA. LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Table of Contents 1 2 3 4 5 6 7 Features .................................................................. Applications ........................................................... Description ............................................................. Revision History..................................................... Pin Configuration and Functions ......................... Specifications......................................................... 1 1 1 2 3 4 6.1 6.2 6.3 6.4 6.5 6.6 4 4 4 5 5 7 Absolute Maximum Ratings ...................................... ESD Ratings.............................................................. Recommended Operating Conditions....................... Thermal Information .................................................. Electrical Characteristics........................................... Typical Characteristics .............................................. Detailed Description ............................................ 12 7.1 7.2 7.3 7.4 Overview ................................................................. Functional Block Diagram ....................................... Feature Description................................................. Device Functional Modes........................................ 12 13 13 16 8 Application and Implementation ........................ 17 8.1 Application Information............................................ 17 8.2 Typical Applications ................................................ 28 9 Power Supply Recommendations...................... 41 10 Layout................................................................... 41 10.1 Layout Guidelines ................................................. 41 10.2 Layout Example ................................................... 42 11 Device and Documentation Support ................. 43 11.1 11.2 11.3 11.4 11.5 11.6 11.7 11.8 Device Support...................................................... Documentation Support ........................................ Related Links ........................................................ Receiving Notification of Documentation Updates Community Resources.......................................... Trademarks ........................................................... Electrostatic Discharge Caution ............................ Glossary ................................................................ 43 43 43 43 43 43 44 44 12 Mechanical, Packaging, and Orderable Information ........................................................... 44 4 Revision History NOTE: Page numbers for previous revisions may differ from page numbers in the current version. Changes from Revision E (April 2013) to Revision F Page * Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information section .................................................................................................. 1 * Changed values in the Thermal Information table to align with JEDEC standards................................................................ 5 Changes from Revision D (April 2013) to Revision E * 2 Page Changed layout of National Semiconductor Data Sheet to TI format .................................................................................... 1 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 5 Pin Configuration and Functions DSC Package 10-Pin WSON Top View DGQ Package 10-Pin MSOP-PowerPAD Top View SW 1 10 PVIN SW 1 10 PVIN SW 2 9 PVIN SW 2 9 PVIN BOOST 3 8 AVIN BOOST 3 8 AVIN EN 4 7 GND EN 4 7 GND SYNC 5 6 FB SYNC 5 6 FB DAP DAP Not to scale Not to scale Pin Functions PIN TYPE (1) DESCRIPTION NO. NAME 1, 2 SW O Output switch. Connects to the inductor, catch diode, and bootstrap capacitor. 3 BOOST I Boost voltage that drives the internal NMOS control switch. A bootstrap capacitor is connected between the BOOST and SW pins. 4 EN I Enable control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN + 0.3 V. 5 SYNC I Frequency synchronization input. Drive this pin with an external clock or pulse train. Ground it to use the internal clock. 6 FB I Feedback pin. Connect FB to the external resistor divider to set output voltage. 7 GND G Signal and power ground pin. Place the bottom resistor of the feedback network as close as possible to this pin for accurate regulation. 8 AVIN I Supply voltage for the control circuitry. 9, 10 PVIN I Supply voltage for output power stage. Connect a bypass capacitor to this pin. DAP DAP G Signal or power ground and thermal connection. Tie this directly to GND (pin 7). See Application Information regarding optimum thermal layout. (1) G = Ground, I = Input, O = Output Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 3 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com 6 Specifications 6.1 Absolute Maximum Ratings over operating free-air temperature range (unless otherwise noted) (1) (2) MIN MAX UNIT AVIN, PVIN -0.5 24 V SW voltage -0.5 24 V Boost voltage -0.5 28 V Boost to SW voltage -0.5 6 V FB voltage -0.5 3 V SYNC voltage -0.5 6 V EN voltage -0.5 VIN + 0.3 V Soldering, infrared reflow (5 s) 260 C Junction temperature, TJ 150 C 150 C Storage temperature, Tstg (1) (2) -65 Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and specifications. 6.2 ESD Ratings V(ESD) (1) (2) Electrostatic discharge Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1) (2) VALUE UNIT 2000 V JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process. Human body model, 1.5 k in series with 100 pF. 6.3 Recommended Operating Conditions over operating free-air temperature range (unless otherwise noted) (1) MIN MAX AVIN, PVIN 3 20 UNIT V SW voltage -0.5 20 V Boost voltage V -0.5 24 Boost to SW voltage 3 5.5 V Junction temperature -40 125 C (1) 4 Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or other conditions beyond those indicated in the recommended Operating Ratings is not implied. The recommended Operating Ratings indicate conditions at which the device is functional and should not be operated beyond such conditions. Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 6.4 Thermal Information LM2734x, LM2734x-Q1 THERMAL METRIC (1) DSC (WSON) DGQ (MSOP-PowerPAD) 10 PINS 10 PINS UNIT RJA Junction-to-ambient thermal resistance (2) 47.6 49.5 C/W RJC(top) Junction-to-case (top) thermal resistance 36.5 53.6 C/W RJB Junction-to-board thermal resistance 22.5 33.7 C/W JT Junction-to-top characterization parameter 0.4 3.9 C/W JB Junction-to-board characterization parameter 22.7 33.4 C/W RJC(bot) Junction-to-case (bottom) thermal resistance 4.7 3.5 C/W (1) (2) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application report. Thermal shutdown will occur if the junction temperature exceeds 165C. The maximum power dissipation is a function of TJ(MAX), RJA and TA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) - TA)/ RJA. All numbers apply for packages soldered directly onto a 3" x 3" PCB with 2 oz. copper on 4 layers in still air. 6.5 Electrical Characteristics TJ = 25C, VIN = 12 V, and VBOOST - VSW = 4.3 V (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX 0.99 1 1.01 0.984 1 1.014 UNIT SYSTEM PARAMETERS TJ = 0C to 85C VFB Feedback voltage VFB/VIN Feedback voltage line regulation IFB Feedback input bias current OVP Overvoltage protection TJ = -40C to 125C VIN = 3 V to 20 V TJ = 25C Undervoltage hysteresis SS 100 VFB at which PWM halts UVLO VIN falling from UVLO 1.13 TJ = 25C TJ = -40C to 125C TJ = -40C to 125C Quiescent current IBOOST Boost pin current 2.6 fSW= 2 MHz V 2.9 0.47 0.3 0.5 IQ = IQ_AVIN + IQ_PVIN nA 2.75 TJ = 25C Soft-start time IQ V 20 TJ = -40C to 125C VIN rising until VSW is switching Undervoltage lockout 0.003% V V 0.6 1 1.5 ms VFB = 1.1 (not switching) 2.4 mA VEN = 0 V (shutdown) 70 nA TJ = 25C 8.2 TJ = -40C to 125C 10 fSW= 1 MHz 4.4 mA 6 OSCILLATOR TJ = 25C 2 fSW Switching frequency SYNC = GND VFB_FOLD FB pin voltage SYNC input is overridden 0.53 fFOLD_MIN Frequency foldback minimum VFB = 0 V 220 TJ = -40C to 125C 1.75 2.3 MHz V 250 kHz 2.35 MHz LOGIC INPUTS (EN, SYNC) fSYNC SYNC frequency range 1 VIL EN, SYNC logic low threshold Logic falling edge VIH EN, SYNC logic high threshold Logic rising edge tSYNC_HIGH SYNC, time required above VIH to ensure a logical high 0.4 V Copyright (c) 2008-2016, Texas Instruments Incorporated 1.8 100 Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 ns 5 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Electrical Characteristics (continued) TJ = 25C, VIN = 12 V, and VBOOST - VSW = 4.3 V (unless otherwise noted) PARAMETER TEST CONDITIONS tSYNC_LOW SYNC, time required below VIL to ensure a logical low ISYNC SYNC pin current IEN MIN VSYNC < 5 V MAX UNIT 100 ns 20 VEN = 3 V Enable pin current TYP VIN = VEN = 20 V nA 6 15 50 100 A INTERNAL MOSFET TJ = 25C 150 RDS(ON) Switch ON-resistance ICL Switch current limit TJ = -40C to 125C DMAX Maximum duty cycle SYNC = GND tMIN Minimum ON-time 65 ns ISW Switch leakage current 40 nA 3.9 V TJ = -40C to 125C 320 LM27342 2.5 4 LM27341 2 3.7 TJ = 25C m A 93% TJ = -40C to 125C 85% BOOST LDO VLDO Boost LDO output voltage THERMAL TSHDN 6 Thermal shutdown temperature Junction temperature rising 165 Thermal shutdown hysteresis Junction temperature falling 15 C Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 6.6 Typical Characteristics TA = 25C, VIN = 12 V, and VBOOST - VSW = 4.3 V (unless otherwise noted) VOUT = 5 V fSW = 2 MHz VOUT = 5 V Figure 2. Load Transient See Figure 34 Figure 1. Efficiency vs Load Current See Figure 34 VOUT = 3.3 V fSW = 2 MHz VOUT = 3.3 V fSW = 2 MHz Figure 5. Efficiency vs Load Current See Figure 49 Copyright (c) 2008-2016, Texas Instruments Incorporated IOUT = 100 mA - 2 A at Slewrate = 2 A / s Figure 4. Load Transient See Figure 40 Figure 3. Efficiency vs Load Current See Figure 40 VOUT = 1.8 V IOUT = 100 mA - 2 A at Slewrate = 2 A / s VOUT = 1.8 V IOUT = 100 mA - 2 A at Slewrate = 2 A / s Figure 6. Load Transient See Figure 49 Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 7 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Typical Characteristics (continued) TA = 25C, VIN = 12 V, and VBOOST - VSW = 4.3 V (unless otherwise noted) VIN = 10 V to 15 V 8 VOUT = 3.3 V No CFF VIN = 10 V to 15 V VOUT = 3.3 V Figure 7. Line Transient See Figure 43 Figure 8. Line Transient See Figure 40 Figure 9. Short Circuit Figure 10. Short-Circuit Release Figure 11. Soft Start With EN Tied to VIN Figure 12. Soft Start With EN Tied to VIN Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 Typical Characteristics (continued) TA = 25C, VIN = 12 V, and VBOOST - VSW = 4.3 V (unless otherwise noted) VIN = 12 V VOUT = 5 V L = 2.2 H COUT = 44 F Iout =1 A VIN = 12 V VOUT = 3.3 V Figure 13. Bode Plot See Figure 34 VIN = 5 V VOUT = 1.8 V L = 1.0 H COUT = 44 F L = 1.5 H COUT = 44 F Figure 14. Bode Plot See Figure 40 Iout =1 A VIN = 5 V VOUT = 1.2 V L = 0.56 H COUT = 68 F Figure 15. Bode Plot See Figure 49 Figure 16. Bode Plot See Figure 55 Figure 17. Sync Functionality Figure 18. Loss of Synchronization Copyright (c) 2008-2016, Texas Instruments Incorporated Iout =1 A Iout =1 A Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 9 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Typical Characteristics (continued) TA = 25C, VIN = 12 V, and VBOOST - VSW = 4.3 V (unless otherwise noted) VSYNC = GND fSW = 2 MHz Figure 19. Oscillator Frequency vs Temperature Figure 20. Oscillator Frequency vs VFB Figure 21. VFB vs Temperature Figure 22. VFB vs VIN VIN = 12 V Figure 23. Current Limit vs Temperature 10 Submit Documentation Feedback Figure 24. RDSON vs Temperature Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 Typical Characteristics (continued) TA = 25C, VIN = 12 V, and VBOOST - VSW = 4.3 V (unless otherwise noted) IQ = IAVIN + IPVIN Figure 25. IQ (Shutdown) vs Temperature Copyright (c) 2008-2016, Texas Instruments Incorporated Figure 26. IEN vs VEN Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 11 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com 7 Detailed Description 7.1 Overview The LM2734x and LM2734x-Q1 are a constant-frequency, peak current-mode PWM buck regulator IC that delivers a 1.5-A or 2-A load current. The regulator has a preset switching frequency of 2 MHz. This high frequency allows the LM2734x and LM2734x-Q1 to operate with small surface-mount capacitors and inductors, resulting in a DC-DC converter that requires a minimum amount of board space. The LM2734x and LM2734x-Q1 are internally compensated, which reduces design time, and requires few external components. The following operating description of the LM2734x and LM2734x-Q1 refers to Functional Block Diagram and to the waveforms in Figure 27. The LM2734x and LM2734x-Q1 supply a regulated output voltage by switching the internal NMOS switch at a constant frequency and varying the duty cycle. A switching cycle begins at the falling edge of the reset pulse generated by the internal oscillator. When this pulse goes low, the output control logic turns on the internal NMOS switch. During this on-time, the SW pin voltage (VSW) swings up to approximately VIN, and the inductor current (iL) increases with a linear slope. The current-sense amplifier measures iL, which generates an output proportional to the switch current typically called the sense signal. The sense signal is summed with the regulator's corrective ramp and compared to the error amplifier's output, which is proportional to the difference between the feedback voltage (VFB) and VREF. When the output of the PWM comparator goes high, the switch turns off until the next switching cycle begins. During the switch off-time (tOFF), inductor current discharges through the catch diode D1, which forces the SW pin (VSW) to swing below ground by the forward voltage (VD1) of the catch diode. The regulator loop adjusts the duty cycle (D) to maintain a constant output voltage. VSW D = tON/TSW SW Voltage VIN tOFF tON 0 -VD1 Inductor Current iL t TSW ILPK IOUT 0 'iL t Figure 27. LM2734x Waveforms of SW Pin Voltage and Inductor Current 12 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 7.2 Functional Block Diagram BOOST D2 LDO PVIN C2 Switch 0.15 Y RSENSE L SW VOUT iL + AVIN D1 Driver Current Sense Amplifier C3 EN Under Voltage Lockout PWM Logic PWM Comparator Reset Pulse Current Limit Thermal Shutdown + + Error Signal ISENSE + OVP Comparator + Corrective Ramp Soft Start SYNC Oscillator 1.13V +FB Internal Compensation Error Amplifier + + R2 GND + Freq. Foldback Amplifier VREF +1.0V R1 + 0.53V - Copyright (c) 2016, Texas Instruments Incorporated 7.3 Feature Description 7.3.1 Boost Function Capacitor C2 in Functional Block Diagram, commonly referred to as CBOOST, is used to store a voltage VBOOST. When LM2734x and LM2734x-Q1 start up, an internal LDO charges CBOOST through an internal diode, to a voltage sufficient to turn the internal NMOS switch on. The gate drive voltage supplied to the internal NMOS switch is VBOOST - VSW. During a normal switching cycle, when the internal NMOS control switch is off (tOFF) (see Figure 27), VBOOST equals VLDO minus the forward voltage of the internal diode (VD2). At the same time the inductor current (iL) forward biases the catch diode D1 forcing the SW pin to swing below ground by the forward voltage drop of the catch diode (VD1). Therefore, the voltage stored across CBOOST is calculated with Equation 1 and Equation 2. VBOOST - VSW = VLDO - VD2 + VD1 VBOOST = VSW + VLDO - VD2 + VD1 (1) (2) When the NMOS switch turns on (tON), the switch pin rises to Equation 3. VSW = VIN - (RDSON x IL) (3) Then the D1 undergoes reverse biasing, and forces VBOOST to rise. The voltage at VBOOST is then calculated with Equation 4. VBOOST = VIN - (RDSON x IL) + VLDO - VD2 + VD1 (4) Which is approximately calculated with Equation 5. VIN + VLDO - 0.4 V (5) VBOOST has pulled itself up by its bootstraps, or boosted to a higher voltage. Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 13 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Feature Description (continued) 7.3.2 Low Input Voltage Considerations When the input voltage is below 5 V and the duty cycle is greater than 75%, the gate drive voltage developed across CBOOST might not be sufficient for proper operation of the NMOS switch. In this case, CBOOST must be charged through an external Schottky diode attached to a 5-V voltage rail (see Figure 28). This ensures that the gate drive voltage is high enough for proper operation of the NMOS switch in the triode region. Maintain VBOOST - VSW less than the 6-V absolute maximum rating. D2 VIN PVIN 5V BOOST C2 AVIN L1 C1 VOUT SW C3 D1 LM27342 ON EN OFF R1 SYNC CLK GND/DAP FB R2 Copyright (c) 2016, Texas Instruments Incorporated Figure 28. External Diode Charges CBOOST 7.3.3 High Output Voltage Considerations When the output voltage is greater than 3.3 V, a minimum load current is required to charge CBOOST (see Figure 29). The minimum load current forward biases the catch diode D1 forcing the SW pin to swing below ground. This allows CBOOST to charge, ensuring that the gate drive voltage is high enough for proper operation. The minimum load current depends on many factors including the inductor value. Figure 29. Minimum Load Current for L = 1.5 H 14 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 Feature Description (continued) 7.3.4 Frequency Synchronization The LM2734x and LM2734x-Q1 switching frequency can be synchronized to an external clock, between 1 MHz and 2.35 MHz, applied at the SYNC pin. At the first rising edge applied to the SYNC pin, the internal oscillator is overridden and subsequent positive edges initiate switching cycles. If the external SYNC signal is lost during operation, the LM2734x and LM2734x-Q1 revert to its internal 2-MHz oscillator within 1.5 s. To disable frequency synchronization and use the internal 2-MHz oscillator, connect the SYNC pin to GND. The SYNC pin gives the designer the flexibility to optimize their design. A lower switching frequency can be chosen for higher efficiency. A higher switching frequency can be chosen to keep EMI out of sensitive ranges such as the AM radio band. Synchronization can also be used to eliminate beat frequencies generated by the interaction of multiple switching power converters. Synchronizing multiple switching power converters result in cleaner power rails. The selected switching frequency (fSYNC) and the minimum on-time (tMIN) limit the minimum duty cycle (DMIN) of the device as calculated with Equation 6. DMIN = tMIN x fSYNC (6) Operation below DMIN is not recommended. The LM2734x and LM2734x-Q1 skip pulses to keep the output voltage in regulation, and the current limit is not ensured. The switching is in phase but no longer at the same switching frequency as the SYNC signal. 7.3.5 Current Limit The LM2734x and LM2734x-Q1 use cycle-by-cycle current limiting to protect the output switch. During each switching cycle, a current limit comparator detects if the output switch current exceeds 2 A minimum (LM27341) or 2.5 A minimum (LM27342), and turns off the switch until the next switching cycle begins. 7.3.6 Frequency Foldback The LM2734x and LM2734x-Q1 employ frequency foldback to protect the device from current run-away during output short-circuit. Once the FB pin voltage falls below regulation, the switch frequency smoothly reduce with the falling FB voltage until the switch frequency reaches 220 kHz (typical). If the device is synchronized to an external clock, synchronization is disabled until the FB pin voltage exceeds 0.53 V. 7.3.7 Output Overvoltage Protection The overvoltage comparator turns off the internal power NFET when the FB pin voltage exceeds the internal reference voltage by 13% (VFB > 1.13 x VREF). With the power NFET turned off the output voltage decreases toward the regulation level. 7.3.8 Undervoltage Lockout Undervoltage lockout (UVLO) prevents the LM2734x and LM2734x-Q1 from operating until the input voltage exceeds 2.75 V (typical). The UVLO threshold has approximately 470 mV of hysteresis, so the part operates until VIN drops below 2.28 V (typical). Hysteresis prevents the part from turning off during power up if VIN has finite impedance. 7.3.9 Thermal Shutdown Thermal shutdown limits total power dissipation by turning off the internal NMOS switch when the IC junction temperature exceeds 165C (typical). After thermal shutdown occurs, hysteresis prevents the internal NMOS switch from turning on until the junction temperature drops to approximately 150C. Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 15 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com 7.4 Device Functional Modes 7.4.1 Enable Pin and Shutdown Mode Connect the EN pin to a voltage source greater than 1.8 V to enable operation of the LM2734x and LM2734x-Q1. Apply a voltage less than 0.4 V to put the part into shutdown mode. In shutdown mode, the quiescent current drops to typically 70 nA. Switch leakage adds another 40 nA from the input supply. For proper operation, the LM2734x and LM2734x-Q1 EN pin must never be left floating, and the voltage must never exceed VIN + 0.3 V. The simplest way to enable the operation of LM2734x and LM2734x-Q1 is to connect the EN pin to AVIN which allows self start-up of the LM2734x and LM2734x-Q1 when the input voltage is applied. When the rise time of VIN is longer than the soft-start time of the LM2734x and LM2734x-Q1, this method may result in an overshoot in output voltage. In such applications, the EN pin voltage can be controlled by a separate logic signal, or tied to a resistor divider, which reaches 1.8 V after VIN is fully established (see Figure 30). This minimizes the potential for output voltage overshoot during a slow VIN ramp condition. Use the lowest value of VIN, seen in your application when calculating the resistor network using Equation 7, to ensure that the 1.8 V minimum EN threshold is reached. VIN PVIN BOOST C2 AVIN L1 C1 VOUT SW R3 C3 D1 LM27342 EN R4 R1 SYNC CLK GND/DAP FB R2 Copyright (c) 2016, Texas Instruments Incorporated Figure 30. Resistor Divider on EN R3 = VIN - 1 x R4 1.8 (7) 7.4.2 Soft-Start Mode The LM2734x and LM2734x-Q1 have a fixed internal soft-start of 1 ms (typical). During soft start, the error amplifier's reference voltage ramps from 0 V to its nominal value of 1 V in approximately 1 ms. This forces the regulator output to ramp in a controlled fashion, which helps reduce inrush current. Upon soft start, the part is initially in frequency foldback and the frequency rises as FB rises. The regulator rises gradually to 2 MHz. The LM2734x and LM2734x-Q1 allows synchronization to an external clock at FB > 0.53 V. 16 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 8 Application and Implementation NOTE Information in the following applications sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI's customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality. 8.1 Application Information 8.1.1 Inductor Selection Inductor selection is critical to the performance of the LM2734x and LM2734x-Q1. The selection of the inductor affects stability, transient response and efficiency. A key factor in inductor selection is determining the ripple current (iL) (see Figure 27). The ripple current (iL) is important in many ways. First, by allowing more ripple current, lower inductance values can be used with a corresponding decrease in physical dimensions and improved transient response. On the other hand, allowing less ripple current increases the maximum achievable load current and reduce the output voltage ripple (see Output Capacitor for more details on calculating output voltage ripple). Increasing the maximum load current is achieved by ensuring that the peak inductor current (ILPK) never exceeds the minimum current limit of 2 A for the LM27341 or 2.5 A for the LM27342 in Equation 8. ILPK = IOUT + iL / 2 (8) Secondly, the slope of the ripple current affects the current control loop. The LM2734x and LM2734x-Q1 have a fixed slope corrective ramp. When the slope of the current ripple becomes significantly less than the converter's corrective ramp, the inductor pole moves from high frequencies to lower frequencies. This negates one advantage that peak current-mode control has overvoltage-mode control, which is, a single low-frequency pole in the power stage of the converter. This can reduce the phase margin, crossover frequency and potentially cause instability in the converter. Contrarily, when the slope of the ripple current becomes significantly greater than the converter's corrective ramp, resonant peaking can occur in the control loop. This can also cause instability (subharmonic oscillation) in the converter. For the power supply designer, this means that for lower switching frequencies the current ripple must be increased to keep the inductor pole well above crossover. It also means that for higher switching frequencies the current ripple must be decreased to avoid resonant peaking. With all these factors, the desired ripple current is selected with Equation 9. The ripple ratio (r) is defined as the ratio of inductor ripple current (iL) to output current (IOUT), evaluated at maximum load. r 'iL IOUT (9) A good compromise between physical size, transient response and efficiency is achieved when we set the ripple ratio between 0.2 and 0.4. The recommended ripple ratio versus duty cycle shown in Figure 31 is based upon this compromise and control loop optimizations. Note that this is just a guideline. See AN-1197 Selecting Inductors for Buck Converters for further considerations. Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 17 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Application Information (continued) Figure 31. Recommended Ripple Ratio vs Duty Cycle The duty cycle (D) can be approximated quickly using the ratio of output voltage (VOUT) to input voltage (VIN) in Equation 10. VOUT VIN D (10) The application's lowest input voltage must be used to calculate the ripple ratio. The catch diode forward voltage drop (VD1) and the voltage drop across the internal NFET (VDS) must be included to calculate a more accurate duty cycle. Calculate D by using Equation 11. VOUT VD1 VIN VD1 VDS D (11) VDS can be approximated with Equation 12. VDS = IOUT x RDS(ON) (12) The diode forward drop (VD1) can range from 0.3 V to 0.5 V depending on the quality of the diode. The lower VD1 is, the higher the operating efficiency of the converter. Now that the ripple current or ripple ratio is determined, the required inductance is calculated with Equation 13. L VOUT VD1 u (1 DMIN) IOUT u r u fsw where * * * DMIN is the duty cycle calculated with the maximum input voltage fsw is the switching frequency IOUT is the maximum output current of 2 A (13) Using IOUT = 2 A minimizes the inductor's physical size. 18 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 Application Information (continued) 8.1.1.1 Inductor Calculation Example Operating conditions for the LM27342 are listed in Table 1. Table 1. Operating Conditions for Inductor Example OPERATING PARAMETERS VIN = 7 - 16 V VOUT = 3.3 V fSW = 2 MHz VD1 = 0.5 V IOUT = 2 A First the maximum duty cycle is calculated with Equation 14. DMAX = (VOUT + VD1) / (VIN + VD1 - VDS) = (3.3 V + 0.5 V) / (7 V + 0.5 V - 0.3 V) = 0.528 (14) Using Figure 31 gives us a recommended ripple ratio = 0.4. Now the minimum duty cycle is calculated with Equation 15. DMIN = (VOUT + VD1) / (VIN + VD1 - VDS) = (3.3 V + 0.5 V) / (16 V + 0.5 V - 0.3 V) = 0.235 (15) The inductance can now be calculated with Equation 16. L = (1 - DMIN) x (VOUT + VD1) / (IOUT x r x fsw) = (1 - 0.235) x (3.3 V + .5 V) / (2 A x 0.4 x 2 MHz) = 1.817 H (16) This is close to the standard inductance value of 1.8 H. This leads to a 1% deviation from the recommended ripple ratio, which is now 0.4038. Finally, we check that the peak current does not reach the minimum current limit of 2.5 A with Equation 17. ILPK = IOUT x (1 + r / 2) = 2 A x (1 + .4038 / 2) = 2.404 A (17) The peak current is less than 2.5 A, so the DC load specification can be met with this ripple ratio. To design for the LM27341 simply replace IOUT = 1.5 A in the equations for ILPK and see that ILPK does not exceed the LM27341's current limit of 2 A (minimum). 8.1.2 Inductor Material Selection When selecting an inductor, make sure that it is capable of supporting the peak output current without saturating. Inductor saturation results in a sudden reduction in inductance and prevent the regulator from operating correctly. To prevent the inductor from saturating over the entire -40 C to 125 C range, pick an inductor with a saturation current higher than the upper limit of ICL listed in Electrical Characteristics. Ferrite core inductors are recommended to reduce AC loss and fringing magnetic flux. The drawback of ferrite core inductors is their quick saturation characteristic. The current limit circuit has a propagation delay and so is oftentimes not fast enough to stop a saturated inductor from going above the current limit. This has the potential to damage the internal switch. To prevent a ferrite core inductor from getting into saturation, the inductor saturation current rating must be higher than the switch current limit ICL. The LM2734x and LM2734x-Q1 are quite robust in handling short pulses of current that are a few amps above the current limit. Saturation protection is provided by a second current limit which is 30% higher than the cycle by cycle current limit. When the saturation protection is triggered the part turns off the output switch and attempt to soft start. When a compromise must be made, pick an inductor with a saturation current just above the lower limit of the ICL. Be sure to validate the short-circuit protection over the intended temperature range. An inductor's saturation current is usually lower when hot. So consult the inductor vendor if the saturation current rating is only specified at room temperature. Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 19 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Soft saturation inductors such as the iron powder types can also be used. Such inductors do not saturate suddenly and therefore are safer when there is a severe overload or even shorted output. Their physical sizes are usually smaller than the Ferrite core inductors. The downside is their fringing flux and higher power dissipation due to relatively high AC loss, especially at high frequencies. 8.1.3 Input Capacitor An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The primary specifications of the input capacitor are capacitance, voltage, RMS current rating, and Equivalent Series Inductance (ESL). The recommended input capacitance is 10 F, although 4.7 F works well for input voltages below 6 V. The input voltage rating is specifically stated by the capacitor manufacturer. Make sure to check any recommended deratings and also verify if there is any significant change in capacitance at the operating input voltage and the operating temperature. The input capacitor maximum RMS input current rating (IRMS-IN) must be greater than Equation 18. IRMS IN r2 IOUT u D u (1 D ) 12 where * * * r is the ripple ratio defined earlier IOUT is the output current D is the duty cycle (18) Equation 18 shows that maximum RMS capacitor current occurs when D = 0.5. Always calculate the RMS at the point where the duty cycle, D, is closest to 0.5. The ESL of an input capacitor is usually determined by the effective cross sectional area of the current path. A large leaded capacitor has high ESL and a 0805 ceramic chip capacitor has very low ESL. At the operating frequencies of the LM2734x and LM2734x-Q1, certain capacitors may have an ESL so large that the resulting impedance (2fL) is higher than that required to provide stable operation. As a result, TI strongly recommends surface-mount capacitors. Sanyo POSCAP, Tantalum or Niobium, Panasonic SP or Cornell Dubilier Low ESR are all good choices for input capacitors and have acceptable ESL. Multilayer ceramic capacitors (MLCC) have very low ESL. For MLCCs, TI recommends using X7R or X5R dielectrics. Consult the capacitor manufacturer's data sheet to see how rated capacitance varies over operating conditions. 8.1.4 Output Capacitor The output capacitor is selected based upon the desired output ripple and transient response. The LM2734x and LM2734x-Q1 loop compensation is designed for ceramic capacitors. A minimum of 22 F is required at 2 MHz (33 F at 1 MHz) while 47 F to 100 F is recommended for improved transient response and higher phase margin. The output voltage ripple of the converter is calculated with Equation 19. 'VOUT 'iL u (RESR 1 ) 8 u fSW u COUT (19) When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the output ripple is approximately sinusoidal and 90 phase shifted from the switching action. Another benefit of ceramic capacitors is their ability to bypass high-frequency noise. A certain amount of switching edge noise couples through parasitic capacitances in the inductor to the output. A ceramic capacitor bypasses this noise while a tantalum does not. The transient response is determined by the speed of the control loop and the ability of the output capacitor to provide the initial current of a load transient. Capacitance can be increased significantly with little detriment to the regulator stability. However, increasing the capacitance provides dimininshing improvement over 100 F in most applications, because the bandwidth of the control loop decreases as output capacitance increases. If improved transient performance is required, add a feedforward capacitor. This becomes especially important for higher output voltages where the bandwidth of the LM2734x and LM2734x-Q1 is lower (see Feedforward Capacitor (Optional) and Frequency Synchronization for more information). 20 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 Check the RMS current rating of the capacitor. The RMS current rating of the capacitor chosen must also meet the following condition with Equation 20. IRMS OUT IOUT u r 12 where * * IOUT is the output current r is the ripple ratio (20) 8.1.5 Catch Diode The catch diode (D1) conducts during the switch off-time. A Schottky diode is recommended for its fast switching times and low forward voltage drop. The catch diode must be chosen so that its current rating is greater than Equation 21. ID1 = IOUT x (1 - D) (21) The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin. To improve efficiency, choose a Schottky diode with a low forward voltage drop. 8.1.6 Boost Diode (Optional) For circuits with input voltages VIN < 5 V and duty cycles (D) > 0.75 V, TI recommends a small-signal Schottky diode. A good choice is the BAT54 small signal diode. The cathode of the diode is connected to the BOOST pin and the anode to a 5-V voltage rail. 8.1.7 Boost Capacitor A ceramic 0.1-F capacitor with a voltage rating of at least 6.3 V is sufficient. The X7R and X5R MLCCs provide the best performance. 8.1.8 Output Voltage The output voltage is set using the following equation where R2 is connected between the FB pin and GND, and R1 is connected between VOUT and the FB pin in Equation 22. A good starting value for R2 is 1 k. VOUT * R1 1 u R2 (c) VREF (22) 8.1.9 Feedforward Capacitor (Optional) A feedforward capacitor (CFF) can improve the transient response of the converter. Place CFF in parallel with R1. The value of CFF must place a zero in the loop response at, or above, the pole of the output capacitor and RLOAD as calculated in Equation 23. The CFF capacitor increases the crossover frequency of the design, thus a larger minimum output capacitance is required for designs using CFF. CFF must only be used with an output capacitance greater than or equal to 44 F. CFF VOUT u COUT IOUT u R1 (23) 8.1.10 Calculating Efficiency and Junction Temperature The complete LM2734x and LM2734x-Q1 DC-DC converter efficiency can be calculated with Equation 24 or Equation 25. K POUT PIN (24) Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 21 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 K www.ti.com POUT POUT PLOSS (25) To determine the most significant power losses, see the following equations. Other losses totaling less than 2% are not discussed. Power loss (PLOSS) is the sum of two basic types of losses in the converter, switching and conduction. Conduction losses usually dominate at higher output loads, where as switching losses remain relatively fixed and dominate at lower output loads. The first step in determining the losses is to calculate the duty cycle (D) with Equation 26. D VOUT VD1 VIN VD1 VDS (26) VDS is the voltage drop across the internal NFET when it is on, and is equal to Equation 27. VDS = IOUT x RDSON (27) VD is the forward voltage drop across the Schottky diode. It can be obtained from the Electrical Characteristics section of the Schottky diode data sheet. If the voltage drop across the inductor (VDCR) is accounted for, the equation changes to Equation 28. D VOUT VD1 VDCR VIN VD1 VDS (28) VDCR usually gives only a minor duty cycle change, and has been omitted in the examples for simplicity. 8.1.10.1 Schottky Diode Conduction Losses The conduction losses in the free-wheeling Schottky diode are calculated with Equation 29. PDIODE = VD1 x IOUT (1 - D) (29) Often this is the single most significant power loss in the circuit. Care must be taken to choose a Schottky diode that has a low forward voltage drop. 8.1.10.2 Inductor Conduction Losses Another significant external power loss is the conduction loss in the output inductor. The equation can be simplified to Equation 30. PIND = IOUT2 x RDCR (30) 8.1.10.3 MOSFET Conduction Losses The LM2734x and LM2734x-Q1 conduction loss is mainly associated with the internal NFET calculated with Equation 31. PCOND = IOUT2 x RDSON x D (31) 8.1.10.4 MOSFET Switching Losses Switching losses are also associated with the internal NFET. They occur during the switch on and off transition periods, where voltages and currents overlap resulting in power loss. The simplest means to determine this loss is to empirically measuring the rise and fall times (10% to 90%) of the switch at the switch node with Equation 32, Equation 33, and Equation 34. Typical values are listed in Table 2. PSWF = 1 / 2 (VIN x IOUT x fSW x tFALL) PSWR = 1 / 2 (VIN x IOUT x fSW x tRISE) PSW = PSWF + PSWR 22 Submit Documentation Feedback (32) (33) (34) Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 Table 2. Typical Rise and Fall Times vs Input Voltage VIN tRISE tFALL 5V 8 ns 8 ns 10 V 9 ns 9 ns 15 V 10 ns 10 ns 8.1.10.5 IC Quiescent Losses Another loss is the power required for operation of the internal circuitry calculated with Equation 35. PQ = IQ x VIN (35) IQ is the quiescent operating current, and is typically around 2.4 mA. 8.1.10.6 MOSFET Driver Losses The other operating power that needs calculation is that required to drive the internal NFET with Equation 36. PBOOST = IBOOST x VBOOST (36) VBOOST is normally between 3 VDC and 5 VDC. The IBOOST rms current is dependant on switching frequency fSW. IBOOST is approximately 8.2 mA at 2 MHz and 4.4 mA at 1 MHz. 8.1.10.7 Total Power Losses Total power losses is calculated with Equation 37. PLOSS = PCOND + PSWR + PSWF + PQ + PBOOST + PDIODE + PIND (37) Losses internal to the LM2734x and LM2734x-Q1 is calculated with Equation 38. PINTERNAL = PCOND + PSWR + PSWF + PQ + PBOOST (38) 8.1.10.8 Efficiency Calculation Example Operating conditions are listed in Table 3. Table 3. Operating Conditions for Efficiency Calculation OPERATING PARAMETERS VIN = 12 V VOUT = 3.3 V IOUT = 2 A fSW = 2 MHZ VD1 = 0.5 V RDCR = 20 m Internal power losses are calculated with Equation 39 through Equation 43. PCOND = IOUT2 x RDSON x D = 22 x 0.15 x 0.314 = 188 mW PSW = (VIN x IOUT x fSW x tFALL) = (12 V x 2A x 2 MHz x 10 ns) = 480 mW PQ = IQ x VIN = 2.4 mA x 12 V = 29 mW PBOOST = IBOOST x VBOOST = 8.2 mA x 4.5 V = 37 mW PINTERNAL = PCOND + PSW + PQ + PBOOST= 733 mW (39) (40) (41) (42) (43) Total power losses are calculated with Equation 44 through Equation 46. PDIODE = VD1 x IOUT (1 - D) = 0.5 V x 2 x (1 - 0.314) = 686 mW PIND = IOUT2 x RDCR Copyright (c) 2008-2016, Texas Instruments Incorporated (44) Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 23 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com = 22 x 20 m = 80 mW PLOSS = PINTERNAL + PDIODE + PIND = 1.499 W (45) (46) The efficiency can now be estimated with Equation 47. POUT POUT PLOSS K 6.6W 6.6W 1.499W 81% (47) With this information, we can estimate the junction temperature of the LM2734x and LM2734x-Q1. 8.1.10.9 Calculating Junction Temperature The thermal definitions are: * TJ = IC junction temperature * TA = Ambient temperature * RJC = Thermal resistance from IC junction to device case * RJA = Thermal resistance from IC junction to ambient air Figure 32. Cross-Sectional View of Integrated Circuit Mounted on a Printed Circuit Board Heat in the LM2734x and LM2734x-Q1 due to internal power dissipation is removed through conduction and/or convection. 8.1.10.9.1 Conduction Heat transfer occurs through cross sectional areas of material. Depending on the material, the transfer of heat can be considered to have poor-to-good thermal conductivity properties (insulator versus conductor). Heat transfer goes from Silicon Lead Frame PCB. 8.1.10.9.2 Convection Heat transfer is by means of airflow. This could be from a fan or natural convection. Natural convection occurs when air currents rise from the hot device to cooler air. Thermal impedance is defined with Equation 48. RT 'T Power (48) Thermal impedance from the silicon junction to the ambient air is defined with Equation 49. RTJA 24 TJ TA Power (49) Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 This impedance can vary depending on the thermal properties of the PCB. This includes PCB size, weight of copper used to route traces , the ground plane, and the number of layers within the PCB. The type and number of thermal vias can also make a large difference in the thermal impedance. Thermal vias are necessary in most applications. They conduct heat from the surface of the PCB to the ground plane. Six to nine thermal vias must be placed under the exposed pad to the ground plane. Placing more than nine thermal vias results in only a small reduction to RJA for the same copper area. These vias must have 8-mil holes to avoid wicking solder away from the DAP. See AN-1187 Leadless Leadframe Package (SNOA401) and AN-1520 A Guide to Board Layout for Best Thermal Resistance for Exposed Packages (SNVA183) for more information on package thermal performance. If a compromise for cost needs to be made, the thermal vias for the MSOP-PowerPAD package can range from 8 mils to 14 mils, increasing the possibility of solder wicking. To predict the silicon junction temperature for a given application, three methods can be used. The first is useful before prototyping and the other two can more accurately predict the junction temperature within the application. 8.1.10.9.3 Method 1 The first method predicts the junction temperature by extrapolating a best guess RJA from the table or graph. The tables and graph are for natural convection. The internal dissipation can be calculated using the efficiency calculations. This allows the user to make a rough prediction of the junction temperature in their application. Methods two and three can later be used to determine the junction temperature more accurately. Table 4 and Table 5 have values of RJA for the WSON and the MSOP-PowerPAD packages. Table 4. RJA Values for the MSOP-PowerPAD at 1-W Dissipation NUMBER OF BOARD LAYERS SIZE OF BOTTOM LAYER COPPER CONNECTED TO DAP SIZE OF TOP LAYER COPPER CONNECTED TO DAP NUMBER OF 10 MIL THERMAL VIAS RJA 2 0.25 in2 0.05 in2 8 80.6C/W 2 0.5625 in 0.05 in2 8 70.9C/W 2 1 in2 0.05 in2 8 62.1C/W 2 2 2 2 1.3225 in 0.05 in 8 54.6C/W 4 (eval board) 3.25 in2 2.25 in2 14 35.3C/W Table 5. RJA Values for the WSON at 1-W Dissipation NUMBER OF BOARD LAYERS SIZE OF BOTTOM LAYER COPPER CONNECTED TO DAP SIZE OF TOP LAYER COPPER CONNECTED TO DAP NUMBER OF 8 MIL THERMAL VIAS RJA 2 0.25 in2 0.05 in2 8 78C/W 2 2 2 0.5625 in 0.05 in 8 65.6C/W 2 1 in2 0.05 in2 8 58.6C/W 2 1.3225 in2 0.05 in2 8 50C/W 15 30.7C/W 4 (eval board) 2 3.25 in Copyright (c) 2008-2016, Texas Instruments Incorporated 2 2.25 in Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 25 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Eight thermal vias and natural convection Figure 33. Estimate of Thermal Resistance vs Ground Copper Area 8.1.10.9.4 Method 2 The second method requires the user to know the thermal impedance of the silicon junction to case. RJC is approximately 9.5C/W for the MSOP-PowerPAD package or 9.1C/W for the WSON. The case temperature must be measured on the bottom of the PCB at a thermal through directly under the DAP of the LM2734x and LM2734x-Q1. The solder resist must be removed from this area for temperature testing. The reading is more accurate if it is taken midway between pins 2 and 9, where the NMOS switch is placed. Knowing the internal dissipation from Method 1, calculate the case temperature (TC) with Equation 50 and Equation 51. RTJC TJ TC Power (50) (51) TJ = (RJC x PLOSS) + TC 8.1.10.9.4.1 Method 2 Example The operating conditions are the same as the previous efficiency calculation listed in Table 6. Table 6. Operating Conditions for Efficiency Calculation OPERATING PARAMETERS VIN = 12 V VOUT = 3.3 V IOUT = 2 A fSW = 2 MHz VD1 = 0.5 V RDCR = 20 m Internal power losses are calculated with Equation 52 through Equation 56. PCOND = IOUT2 x RDSON x D = 22 x 0.15 x 0.314 = 188 mW PSW = (VIN x IOUT x fSW x tFALL) = (12 V x 2 A x 2 MHz x 10 ns) = 480 mW PQ = IQ x VIN = 1.5 mA x 12 V = 29 mW PBOOST = IBOOST x VBOOST = 7 mA x 4.5 V = 37 mW PINTERNAL = PCOND + PSW + PQ + PBOOST = 733 mW 26 Submit Documentation Feedback (52) (53) (54) (55) (56) Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 The junction temperature can now be estimated with Equation 57. TJ = (RJC x PINTERNAL) + TC (57) A TI MSOP-PowerPAD evaluation board was used to determine the TJ of the LM2734x and LM2734x-Q1. The four layer PCB is constructed using FR4 with 2-oz copper traces. There is a ground plane on the internal layer directly beneath the device, and a ground plane on the bottom layer. The ground plane is accessed by fourteen 10-mil vias. The board measures 2 in x 2 in (50.8 mm x 50.8 mm). It was placed in a container with no airflow. The case temperature measured on this LM27342MY Demo Board was 48.7C. Therefore, TJ is calculated with Equation 58 and Equation 59. TJ = (9.5C/W x 733 mW) + 48.7C TJ = 55.66C (58) (59) To keep the junction temperature below 125C for this layout, the ambient temperature must stay below 94.33C as in Equation 60, Equation 61, and Equation 62. TA_MAX = TJ_MAX - TJ +TA TA_MAX = 125C - 55.66C + 25C TA_MAX = 94.33C (60) (61) (62) 8.1.10.9.5 Method 3 The third method can also give a very accurate estimate of silicon junction temperature. The first step is to determine RJA of the application. The LM2734x and LM2734x-Q1 has overtemperature protection circuitry. When the silicon temperature reaches 165C, the device stops switching. The protection circuitry has a hysteresis of 15C. Once the silicon temperature has decreased to approximately 150C, the device starts switching again. Knowing this, the RJA for any PCB can be characterized during the early stages of the design by raising the ambient temperature in the given application until the circuit enters thermal shutdown. If the SWpin is monitored, it is obvious when the internal NFET stops switching indicating a junction temperature of 165C. We can calculate the internal power dissipation from the above methods. All that is required for calculation is the estimate of RDSON at 165C. The value is approximately 0.267 . With this, the junction temperature, and the ambient temperature, RJA, can be determined with Equation 63. RTJA 165qC TA PINTERNAL (63) Once this is determined, the maximum ambient temperature allowed for a desired junction temperature can be found. 8.1.10.9.5.1 Method 3 Example The operating conditions are the same as the previous efficiency calculation listed in Table 7. Table 7. Operating Conditions for Efficiency Calculation OPERATING PARAMETERS VIN = 12 V VOUT = 3.3 V IOUT = 2A fSW = 2 MHz VD1 = 0.5 V RDCR = 20 m Internal power losses are calculated with Equation 64 through Equation 68. PCOND = IOUT2 x RDSON x D = 22 x 0.267 x .314 = 335 mW PSW = (VIN x IOUT x fSW x tFALL) = (12 V x 2 A x 2 MHz x 10 nS) = 480 mW PQ = IQ x VIN = 1.5 mA x 12 V = 29 mW PBOOST = IBOOST x VBOOST = 7 mA x 4.5 V Copyright (c) 2008-2016, Texas Instruments Incorporated (64) (65) (66) Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 27 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com = 37 mW PINTERNAL = PCOND + PSW + PQ + PBOOST = 881 mW (67) (68) A TI MSOP-PowerPAD evaluation board was used to determine the RJA of the board. The four-layer PCB is constructed using FR4 with 2oz copper traces. There is a ground plane on the internal layer directly beneath the device, and a ground plane on the bottom layer. The ground plane is accessed by fourteen 10-mil vias. The board measures 2 in x 2 in (50.8 mm x 50.8 mm). It was placed in an oven with no forced airflow. The ambient temperature was raised to 132C, and at that temperature, the device went into thermal shutdown. RJA can be calculated with Equation 69. RTJA 165qC 132qC 0.881 W 37.46 qC / W (69) To keep the junction temperature below 125C for this layout, the ambient temperature must stay below 92C as in Equation 70, Equation 71, and Equation 72. TA_MAX = TJ_MAX - (RJA x PINTERNAL) TA_MAX = 125C - (37.46C/W x 0.881 W) TA_MAX = 92C (70) (71) (72) This calculation of the maximum ambient temperature is only 2.3C different from the calculation using method 2. The methods described above to find the junction temperature in the MSOP-PowerPAD package can also be used to calculate the junction temperature in the WSON package. The 10-pin WSON package has a RJC = 9.1C/W, while RJA can vary depending on the layout. RJA can be calculated in the same manner as described in method 3. 8.2 Typical Applications 8.2.1 LM2734x Configuration From VIN = 7 V to 16 V, VOUT = 5 V For Full Load at 2 MHz V IN PVIN BOOST C2 AVIN L1 C1 SW VOUT C3 D1 LM27341/2 C4 ON EN OFF R1 FB SYNC CLK 2 MHz GND/DAP R2 C5 Copyright (c) 2016, Texas Instruments Incorporated VIN = 7 V to 16 V, VOUT = 5 V fSW = 2 MHz IOUT = Full load Figure 34. LM2734x Configuration From VIN = 7 V to 16 V, VOUT = 5 V For Full Load at 2-MHz Schematic 28 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 Typical Applications (continued) 8.2.1.1 Design Requirements Create 5-V output at full-rated load for VIN range of 7 V to 16 V with switching frequency FSW = 2 MHz using external synchronization. 8.2.1.2 Detailed Design Procedure The device must be able to operate at any voltage within the recommended operating range. The load current must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle the full expected load current as well as the peak current generated during load transients and start-up. The inrush current at start-up depends on the output capacitor selection. Table 8 lists the bill of materials for VIN = 7 V to 16 V, VOUT = 5 V for full load at 2 MHz. See Figure 34. Table 8. Bill of Materials PART NAME PART ID PART VALUE PART NUMBER MANUFACTURER Buck regulator U1 1.5-A or 2-A Buck regulator LM2734x and LM2734x-Q1 TI CPVIN C1 10 F GRM32DR71E106KA12L Murata CBOOST C2 0.1 F GRM188R71C104KA01D Murata COUT C3 22 F C3225X7R1C226K TDK COUT C4 22 F C3225X7R1C226K TDK CFF C5 0.18 F 0603ZC184KAT2A AVX Catch diode D1 Schottky diode, Vf = 0.32 V CMS06 Toshiba Inductor L1 2.2 H CDRHD5D28RHPNP Sumida Feedback resistor R1 560 CRCW0603560RFKEA Vishay Feedback resistor R2 140 CRCW0603140RFKEA Vishay 8.2.1.3 Application Curves Figure 35. LM27342 Efficiency vs Load Current Copyright (c) 2008-2016, Texas Instruments Incorporated IOUT = 100 mA - 2 A at Slewrate = 2 A / s Figure 36. Transient Response Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 29 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com 8.2.2 LM2734x Configuration From VIN = 7 V to 16 V, VOUT = 5 V For Full Load at 1 MHz VIN PVIN BOOST C2 AVIN L1 C1 VOUT SW D1 LM27341/2 C3 C4 ON OFF EN R1 CLK 1 MHz SYNC FB GND / DAP R2 C5 Copyright (c) 2016, Texas Instruments Incorporated VIN = 7 V to 16 V, VOUT = 5 V fSW = 1 MHz IOUT = Full load Figure 37. LM2734x Configuration From VIN = 7 V to 16 V, VOUT = 5 V For Full Load at 1-MHz Schematic 8.2.2.1 Design Requirements Create 5-V output at full-rated load for VIN range of 7 V to 16 V with switching frequency FSW = 1 MHz using external synchronization. 8.2.2.2 Detailed Design Procedure The device must be able to operate at any voltage within the recommended operating range. The load current must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle the full expected load current as well as the peak current generated during load transients and start-up. The inrush current at start-up depends on the output capacitor selection. Table 9 lists the bill of materials for VIN = 7 V to 16 V, VOUT = 5 V for full load at 1 MHz. See Figure 37. Table 9. Bill of Materials PART NAME PART ID PART VALUE PART NUMBER MANUFACTURER Buck regulator U1 1.5-A or 2-A Buck regulator LM2734x and LM2734x-Q1 TI CPVIN C1 10 F GRM32DR71E106KA12L Murata CBOOST C2 0.1 F GRM188R71C104KA01D Murata COUT C3 47 F GRM32ER61A476KE20L Murata COUT C4 22 F C3225X7R1C226K TDK CFF C5 0.27 F C0603C274K4RACTU Kemet Catch diode D1 Schottky diode, Vf = 0.32 V CMS06 Toshiba Inductor L1 3.3 H CDRH6D26HPNP Sumida Feedback resistor R1 560 CRCW0603560RFKEA Vishay Feedback resistor R2 140 CRCW0603140RFKEA Vishay 30 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 8.2.2.3 Application Curves Figure 38. LM27342 Efficiency vs Load Current 8.2.3 IOUT = 100 mA - 2 A at Slewrate = 2 A / s Figure 39. Transient Response LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V For Full Load at 2 MHz V IN PVIN BOOST C2 AVIN L1 C1 SW VOUT D1 LM27341/2 C3 C4 ON EN OFF R1 FB SYNC CLK 2 MHz GND/DAP R2 C5 Copyright (c) 2016, Texas Instruments Incorporated VIN = 5 V to 16 V, VOUT = 3.3 V fSW = 2 MHz IOUT = Full load Figure 40. LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V For Full Load at 2-MHz Schematic 8.2.3.1 Design Requirements Create 3.3-V output at full-rated load for VIN range of 5 V to 16 V with switching frequency FSW = 2 MHz using external synchronization. Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 31 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com 8.2.3.2 Detailed Design Procedure The device must be able to operate at any voltage within the recommended operating range. The load current must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle the full expected load current as well as the peak current generated during load transients and start-up. The inrush current at start-up depends on the output capacitor selection. Table 10 lists the bill of materials for VIN = 5 V to 16 V, VOUT = 3.3 V for full load at 2 MHz. See Figure 40. Table 10. Bill of Materials PART NAME PART ID PART VALUE PART NUMBER MANUFACTURER Buck regulator U1 1.5-A or 2-A Buck regulator LM2734x and LM2734x-Q1 TI CPVIN C1 10 F GRM32DR71E106KA12L Murata CBOOST C2 0.1 F GRM188R71C104KA01D Murata COUT C3 22 F C3225X7R1C226K TDK COUT C4 22 F C3225X7R1C226K TDK CFF C5 0.18 F 0603ZC184KAT2A AVX Catch diode D1 Schottky diode, Vf = 0.32 V CMS06 Toshiba Inductor L1 1.5 H CDRH5D18BHPNP Sumida Feedback resistor R1 430 CRCW0603430RFKEA Vishay Feedback resistor R2 187 CRCW0603187RFKEA Vishay 8.2.3.3 Application Curves Figure 41. LM27342 Efficiency vs Load Current 32 Submit Documentation Feedback IOUT = 100 mA - 2 A at Slewrate = 2 A / s Figure 42. Transient Response Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 8.2.4 LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V For Full Load at 2 MHz With SYNC = GND BOOST PVIN VIN C2 AVIN L1 C1 SW EN VOUT D1 LM27341/2 C3 C4 R1 SYNC FB GND/DAP R2 Copyright (c) 2016, Texas Instruments Incorporated VIN = 5 V to 16 V, VOUT = 3.3 V fSW = 2 MHz IOUT = Full load Figure 43. LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V For Full Load at 2 MHz With SYNC = GND Schematic 8.2.4.1 Design Requirements Create 3.3-V output at full-rated load for VIN range of 5 V to 16 V with switching frequency FSW = 2 MHz using internal oscillator. 8.2.4.2 Detailed Design Procedure The device must be able to operate at any voltage within the recommended operating range. The load current must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle the full expected load current as well as the peak current generated during load transients and start-up. The inrush current at start-up depends on the output capacitor selection. Table 11 lists the bill of materials for VIN = 5 V to 16 V, VOUT = 3.3 V for full load at 2 MHz with SYNC = GND. See Figure 43. Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 33 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Table 11. Bill of Materials PART NAME PART ID PART VALUE PART NUMBER MANUFACTURER Buck regulator U1 1.5-A or 2-A Buck regulator LM2734x and LM2734x-Q1 TI CPVIN C1 10 F GRM32DR71E106KA12L Murata CBOOST C2 0.1 F GRM188R71C104KA01D Murata COUT C3 22 F C3225X7R1C226K TDK COUT C4 22 F C3225X7R1C226K TDK Catch diode D1 Schottky diode, Vf = 0.32 V CMS06 Toshiba Inductor L1 1.5 H CDRH5D18BHPNP Sumida Feedback resistor R1 430 CRCW0603430RFKEA Vishay Feedback resistor R2 187 CRCW0603187RFKEA Vishay 8.2.4.3 Application Curves IOUT = 100 mA - 2 A at Slewrate = 2 A / s Figure 45. Transient Response Figure 44. LM27342 Efficiency vs Load Current 8.2.5 LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V For Full Load at 2 MHz With SYNC = 1 MHz VIN PVIN BOOST C2 AVIN L1 C1 VOUT SW D1 LM27341/2 C3 C4 ON OFF EN R1 CLK 1 MHz SYNC FB GND / DAP R2 C5 Copyright (c) 2016, Texas Instruments Incorporated VIN = 5 V to 16 V, VOUT = 3.3 V fSW = 1 MHz IOUT = Full load Figure 46. LM2734x Configuration From VIN = 5 V to 16 V, VOUT = 3.3 V For Full Load at 2 MHz With SYNC = 1-MHz Schematic 34 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 8.2.5.1 Design Requirements Create 1.8-V output at full-rated load for VIN range of 5 V to 16 V with switching frequency FSW = 1 MHz using external synchronization. 8.2.5.2 Detailed Design Procedure The device must be able to operate at any voltage within the recommended operating range. The load current must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle the full expected load current as well as the peak current generated during load transients and start-up. The inrush current at start-up depends on the output capacitor selection. Table 12 lists the bill of materials for VIN = 5 V to 16 V, VOUT = 3.3 V for full load at 2 MHz with SYNC = 1 MHz. See Figure 46. Table 12. Bill of Materials PART NAME PART ID PART VALUE PART NUMBER MANUFACTURER Buck regulator U1 1.5-A or 2-A Buck regulator LM2734x and LM2734x-Q1 TI CPVIN C1 10 F GRM32DR71E106KA12L Murata CBOOST C2 0.1 F GRM188R71C104KA01D Murata COUT C3 47 F GRM32ER61A476KE20L Murata COUT C4 22 F C3225X7R1C226K TDK CFF C5 0.27 F C0603C274K4RACTU Kemet Catch diode D1 Schottky diode, Vf = 0.32 V CMS06 Toshiba Inductor L1 2.7 H CDRH5D18BHPNP Sumida Feedback resistor R1 430 CRCW0603430RFKEA Vishay Feedback resistor R2 187 CRCW0603187RFKEA Vishay 8.2.5.3 Application Curves Figure 47. LM27342 Efficiency vs Load Current Copyright (c) 2008-2016, Texas Instruments Incorporated IOUT = 100 mA - 2 A at Slewrate = 2 A / s Figure 48. Transient Response Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 35 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com 8.2.6 LM2734x Configuration From VIN = 3.3 V to 16 V, VOUT = 1.8 V For Full Load at 2 MHz With SYNC = 1 GND BOOST PVIN VIN C2 AVIN L1 C1 SW EN VOUT C3 D1 LM27341/2 C4 R1 SYNC FB GND/DAP R2 Copyright (c) 2016, Texas Instruments Incorporated VIN = 3.3 V to 16 V, VOUT = 1.8 V fSW = 2 MHz IOUT = Full load Figure 49. LM2734x Configuration From VIN = 3.3 V to 16 V, VOUT = 1.8 V For Full Load at 2 MHz With SYNC = GND Schematic 8.2.6.1 Design Requirements Create 1.8-V output at full-rated load for VIN range of 3.3 V to 16 V with switching frequency FSW = 2 MHz using internal oscillator. 8.2.6.2 Detailed Design Procedure The device must be able to operate at any voltage within the recommended operating range. The load current must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle the full expected load current as well as the peak current generated during load transients and start-up. The inrush current at start-up depends on the output capacitor selection. Table 13 lists the bill of materials for VIN = 3.3 V to 16 V, VOUT = 1.8 V for full load at 2 MHz with SYNC = GND. See Figure 49. Table 13. Bill of Materials PART NAME PART ID PART VALUE PART NUMBER MANUFACTURER Buck regulator U1 1.5-A or 2-A Buck regulator LM2734x and LM2734x-Q1 TI CPVIN C1 10 F GRM32DR71E106KA12L Murata CBOOST C2 0.1 F GRM188R71C104KA01D Murata COUT C3 22 F C3225X7R1C226K TDK COUT C4 22 F C3225X7R1C226K TDK Catch diode D1 Schottky diode, Vf = 0.32 V CMS06 Toshiba Inductor L1 1 H CDRH5D18BHPNP Sumida Feedback resistor R1 12 k CRCW060312K0FKEA Vishay Feedback resistor R2 15 k CRCW060315K0FKEA Vishay 36 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 8.2.6.3 Application Curves IOUT = 100 mA - 2 A at Slewrate = 2 A / s Figure 51. Transient Response Figure 50. LM27342 Efficiency vs Load Current 8.2.7 LM2734x Configuration From VIN = 3.3 V to 16 V, VOUT = 1.8 V For Full Load at 2 MHz With SYNC = 1 MHz VIN PVIN BOOST C2 AVIN L1 C1 VOUT SW D1 LM27341/2 C3 C4 ON OFF EN R1 CLK 1 MHz SYNC FB GND / DAP R2 C5 Copyright (c) 2016, Texas Instruments Incorporated VIN = 3.3 V to 16 V, VOUT = 1.8 V fSW = 1 MHz IOUT = Full load Figure 52. LM2734x Configuration From VIN = 3.3 V to 16 V, VOUT = 1.8 V For Full Load at 2 MHz With SYNC = 1-MHz Schematic 8.2.7.1 Design Requirements Create 1.8-V output at full-rated load for VIN range of 3.3 V to 16 V with switching frequency FSW = 1 MHz using external synchronization. 8.2.7.2 Detailed Design Procedure The device must be able to operate at any voltage within the recommended operating range. The load current must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle the full expected load current as well as the peak current generated during load transients and start-up. The inrush current at start-up depends on the output capacitor selection. Table 14 lists the bill of materials for VIN = 3.3 V to 16 V, VOUT = 1.8 V for full load at 2 MHz with SYNC = 1 MHz. See Figure 52. Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 37 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Table 14. Bill of Materials PART NAME PART ID PART VALUE PART NUMBER MANUFACTURER Buck regulator U1 1.5-A or 2-A Buck regulator LM2734x and LM2734x-Q1 TI CPVIN C1 10 F GRM32DR71E106KA12L Murata CBOOST C2 0.1 F GRM188R71C104KA01D Murata COUT C3 22 F C3225X7R1C226K TDK COUT C4 22 F C3225X7R1C226K TDK CFF C5 3.9 nF GRM188R71H392KA01D Murata Catch diode D1 Schottky diode, Vf = 0.32 V CMS06 Toshiba Inductor L1 1.8 H CDRH5D18BHPNP Sumida Feedback resistor R1 12 k CRCW060312K0FKEA Vishay Feedback resistor R2 15 k CRCW060315K0FKEA Vishay 8.2.7.3 Application Curves Figure 53. LM27342 Efficiency vs Load Current 38 Submit Documentation Feedback IOUT = 100 mA - 2 A at Slewrate = 2 A / s Figure 54. Transient Response Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 8.2.8 LM2734x Configuration From VIN = 3.3 V to 9 V, VOUT = 1.2 V For Full Load at 2 MHz With SYNC = 2 MHz V IN PVIN BOOST C2 AVIN L1 C1 SW VOUT C3 D1 LM27341/2 C4 ON EN OFF R1 FB SYNC CLK 2 MHz GND/DAP R2 C5 Copyright (c) 2016, Texas Instruments Incorporated VIN = 3.3 V to 9 V, VOUT = 1.2 V fSW = 2 MHz IOUT = Full load Figure 55. LM2734x Configuration From VIN = 3.3 V to 9 V, VOUT = 1.2 V For Full Load at 2 MHz With SYNC = 2-MHz Schematic 8.2.8.1 Design Requirements Create 1.2-V output at full-rated load for VIN range of 3.3 V to 9 V with switching frequency FSW = 2 MHz using external synchronization. 8.2.8.2 Detailed Design Procedure The device must be able to operate at any voltage within the recommended operating range. The load current must be defined to properly size the inductor, input, and output capacitors. The inductor must be able to handle the full expected load current as well as the peak current generated during load transients and start-up. The inrush current at start-up depends on the output capacitor selection. Table 15 lists the bill of materials for VIN = 3.3 V to 9 V, VOUT = 1.2 V for full load at 2 MHz with SYNC = 2 MHz. See Figure 55. Table 15. Bill of Materials PART NAME PART ID PART VALUE PART NUMBER MANUFACTURER Buck regulator U1 1.5-A or 2-A Buck regulator LM2734x and LM2734x-Q1 TI CPVIN C1 10 F GRM32DR71E106KA12L Murata CBOOST C2 0.1 F GRM188R71C104KA01D Murata COUT C3 47 F GRM32ER61A476KE20L Murata COUT C4 22 F C3225X7R1C226K TDK CFF C5 Not mounted Catch diode D1 Schottky diode, Vf = 0.32 V CMS06 Toshiba Inductor L1 0.56 H CDRH2D18/HPNP Sumida Feedback resistor R1 1.02 k CRCW06031K02FKEA Vishay Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 39 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Table 15. Bill of Materials (continued) PART NAME Feedback resistor PART ID R2 PART VALUE 5.1 k PART NUMBER CRCW06035K10FKEA MANUFACTURER Vishay 8.2.8.3 Application Curves Figure 56. LM27342 Efficiency vs Load Current 40 Submit Documentation Feedback IOUT = 100 mA - 2 A at Slewrate = 2 A / s Figure 57. Transient Response Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 9 Power Supply Recommendations The input voltage is rated as 3 V to 18 V; however, take care in certain circuit configurations (for example, VBOOST derived from VIN where the requirement that VBOOST - VSW < 5.5 V must be observed). Also, for the best efficiency, VBOOST must be at least 2.5 V above VSW. The voltage on the enable pin must not exceed VIN by more than 0.3 V. 10 Layout 10.1 Layout Guidelines 10.1.1 Compact Layout The performance of any switching converter depends as much upon the layout of the PCB as the component selection. The following guidelines help the user design a circuit with maximum rejection of outside EMI and minimum generation of unwanted EMI. Parasitic inductance can be reduced by keeping the power path components close together and keeping the area of the loops small, on which high currents travel. Short, thick traces or copper pours (shapes) are best. In particular, the switch node (where L1, D1, and the SW pin connect) must be just large enough to connect all three components without excessive heating from the current it carries. The LM2734x and LM2734x-Q1 operate in two distinct cycles (see Figure 27) whose high current paths are shown in Figure 58. + - Figure 58. Buck Converter Current Loops The dark grey, inner loop represents the high current path during the MOSFET on-time. The light grey, outer loop represents the high current path during the off-time. 10.1.2 Ground Plane and Shape Routing The diagram of Figure 58 is also useful for analyzing the flow of continuous current versus the flow of pulsating currents. The circuit paths with current flow during both the on-time and off-time are considered to be continuous current, while those that carry current during the on-time or off-time only are pulsating currents. Preference in routing must be given to the pulsating current paths, as these are the portions of the circuit most likely to emit EMI. The ground plane of a PCB is a conductor and return path, and it is susceptible to noise injection just like any other circuit path. The path between the input source and the input capacitor and the path between the catch diode and the load are examples of continuous current paths. In contrast, the path between the catch diode and the input capacitor carries a large pulsating current. This path must be routed with a short, thick shape, preferably on the component side of the PCB. Multiple vias in parallel must be used right at the pad of the input capacitor to connect the component side shapes to the ground plane. A second pulsating current loop that is often ignored is the gate drive loop formed by the SW and BOOST pins and boost capacitor CBOOST. To minimize this loop and the EMI it generates, keep CBOOST close to the SW and BOOST pins. 10.1.3 FB Loop The FB pin is a high-impedance input, and the loop created by R2, the FB pin and ground must be made as small as possible to maximize noise rejection. R2 must therefore be placed as close as possible to the FB and GND pins of the IC. Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 41 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com Layout Guidelines (continued) 10.1.4 PCB Summary 1. Minimize the parasitic inductance by keeping the power path components close together and keeping the area of the high-current loops small. 2. The most important consideration when completing the layout is the close coupling of the GND connections of the CIN capacitor and the catch diode D1. These ground connections must be immediately adjacent, with multiple vias in parallel at the pad of the input capacitor connected to GND. Place CIN and D1 as close to the IC as possible. 3. Next in importance is the location of the GND connection of the COUT capacitor, which must be near the GND connections of CIN and D1. 4. There must be a continuous ground plane on the copper layer directly beneath the converter. This reduces parasitic inductance and EMI. 5. The FB pin is a high impedance node and care must be taken to make the FB trace short to avoid noise pickup and inaccurate regulation. The feedback resistors must be placed as close as possible to the IC, with the GND of R2 placed as close as possible to the GND of the IC. The VOUT trace to R1 must be routed away from the inductor and any other traces that are switching. 6. High AC currents flow through the VIN, SW and VOUT traces, so they must be as short and wide as possible. However, making the traces wide increases radiated noise, so the layout designer must make this trade-off. Radiated noise can be decreased by choosing a shielded inductor. The remaining components must also be placed as close as possible to the IC. See AN-1229 SIMPLE SWITCHER(R) PCB Layout Guidelines (SNVA054) for further considerations and the LM27342 demo board as an example of a four-layer layout. 10.2 Layout Example Figure 59. Top Layer and Overlay 42 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 LM27341, LM27342, LM27341-Q1, LM27342-Q1 www.ti.com SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 11 Device and Documentation Support 11.1 Device Support 11.1.1 Third-Party Products Disclaimer TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE. 11.2 Documentation Support 11.2.1 Related Documentation For related documentation see the following: * AN-1197 Selecting Inductors for Buck Converters (SNVA038) * AN-1187 Leadless Leadframe Package (SNOA401) * AN-1520 A Guide to Board Layout for Best Thermal Resistance for Exposed Packages (SNVA183) * AN-1229 SIMPLE SWITCHER(R) PCB Layout Guidelines (SNVA054) 11.3 Related Links The table below lists quick access links. Categories include technical documents, support and community resources, tools and software, and quick access to sample or buy. Table 16. Related Links PARTS PRODUCT FOLDER SAMPLE & BUY TECHNICAL DOCUMENTS TOOLS & SOFTWARE SUPPORT & COMMUNITY LM27341 Click here Click here Click here Click here Click here LM27342 Click here Click here Click here Click here Click here LM27341-Q1 Click here Click here Click here Click here Click here LM27342-Q1 Click here Click here Click here Click here Click here 11.4 Receiving Notification of Documentation Updates To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper right corner, click on Alert me to register and receive a weekly digest of any product information that has changed. For change details, review the revision history included in any revised document. 11.5 Community Resources The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of Use. TI E2ETM Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help solve problems with fellow engineers. Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and contact information for technical support. 11.6 Trademarks E2E is a trademark of Texas Instruments. All other trademarks are the property of their respective owners. Copyright (c) 2008-2016, Texas Instruments Incorporated Submit Documentation Feedback Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 43 LM27341, LM27342, LM27341-Q1, LM27342-Q1 SNVS497F - NOVEMBER 2008 - REVISED SEPTEMBER 2016 www.ti.com 11.7 Electrostatic Discharge Caution These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 11.8 Glossary SLYZ022 -- TI Glossary. This glossary lists and explains terms, acronyms, and definitions. 12 Mechanical, Packaging, and Orderable Information The following pages include mechanical, packaging, and orderable information. This information is the most current data available for the designated devices. This data is subject to change without notice and revision of this document. For browser-based versions of this data sheet, refer to the left-hand navigation. 44 Submit Documentation Feedback Copyright (c) 2008-2016, Texas Instruments Incorporated Product Folder Links: LM27341 LM27342 LM27341-Q1 LM27342-Q1 PACKAGE OPTION ADDENDUM www.ti.com 4-Feb-2016 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan Lead/Ball Finish MSL Peak Temp (2) (6) (3) Op Temp (C) Device Marking (4/5) LM27341MY/NOPB ACTIVE MSOPPowerPAD DGQ 10 1000 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 SSCB LM27341QMY/NOPB ACTIVE MSOPPowerPAD DGQ 10 1000 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 SSJB LM27341QMYX/NOPB ACTIVE MSOPPowerPAD DGQ 10 3500 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 SSJB LM27341SD/NOPB ACTIVE WSON DSC 10 1000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 L231B LM27342MY/NOPB ACTIVE MSOPPowerPAD DGQ 10 1000 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 SSCA LM27342MYX/NOPB ACTIVE MSOPPowerPAD DGQ 10 3500 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 SSCA LM27342QMY/NOPB ACTIVE MSOPPowerPAD DGQ 10 1000 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 SSJA LM27342QMYX/NOPB ACTIVE MSOPPowerPAD DGQ 10 3500 Green (RoHS & no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 SSJA LM27342SD/NOPB ACTIVE WSON DSC 10 1000 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 L231A LM27342SDX/NOPB ACTIVE WSON DSC 10 4500 Green (RoHS & no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 L231A (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) Addendum-Page 1 Samples PACKAGE OPTION ADDENDUM www.ti.com 4-Feb-2016 (3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. (4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device. (5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation of the previous line and the two combined represent the entire Device Marking for that device. (6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish value exceeds the maximum column width. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. OTHER QUALIFIED VERSIONS OF LM27341, LM27341-Q1, LM27342, LM27342-Q1 : * Catalog: LM27341, LM27342 * Automotive: LM27341-Q1, LM27342-Q1 NOTE: Qualified Version Definitions: * Catalog - TI's standard catalog product * Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects Addendum-Page 2 PACKAGE MATERIALS INFORMATION www.ti.com 20-Sep-2016 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant LM27341MY/NOPB MSOPPower PAD DGQ 10 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 LM27341QMY/NOPB MSOPPower PAD DGQ 10 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 LM27341QMYX/NOPB MSOPPower PAD DGQ 10 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 LM27341SD/NOPB WSON DSC 10 1000 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1 LM27342MY/NOPB MSOPPower PAD DGQ 10 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 LM27342MYX/NOPB MSOPPower PAD DGQ 10 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 LM27342QMY/NOPB MSOPPower PAD DGQ 10 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 LM27342QMYX/NOPB MSOPPower PAD DGQ 10 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 20-Sep-2016 Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant LM27342SD/NOPB WSON DSC 10 1000 178.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1 LM27342SDX/NOPB WSON DSC 10 4500 330.0 12.4 3.3 3.3 1.0 8.0 12.0 Q1 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) LM27341MY/NOPB MSOP-PowerPAD DGQ 10 1000 210.0 185.0 35.0 LM27341QMY/NOPB MSOP-PowerPAD DGQ 10 1000 210.0 185.0 35.0 LM27341QMYX/NOPB MSOP-PowerPAD DGQ 10 3500 367.0 367.0 35.0 LM27341SD/NOPB WSON DSC 10 1000 210.0 185.0 35.0 LM27342MY/NOPB MSOP-PowerPAD DGQ 10 1000 210.0 185.0 35.0 LM27342MYX/NOPB MSOP-PowerPAD DGQ 10 3500 367.0 367.0 35.0 LM27342QMY/NOPB MSOP-PowerPAD DGQ 10 1000 210.0 185.0 35.0 LM27342QMYX/NOPB MSOP-PowerPAD DGQ 10 3500 367.0 367.0 35.0 LM27342SD/NOPB WSON DSC 10 1000 210.0 185.0 35.0 LM27342SDX/NOPB WSON DSC 10 4500 367.0 367.0 35.0 Pack Materials-Page 2 MECHANICAL DATA DGQ0010A MUC10A (Rev A) BOTTOM VIEW www.ti.com PACKAGE OUTLINE DSC0010B WSON - 0.8 mm max height SCALE 4.000 PLASTIC SMALL OUTLINE - NO LEAD 3.1 2.9 B A PIN 1 INDEX AREA 3.1 2.9 C 0.8 MAX 0.08 SEATING PLANE 0.05 0.00 1.20.1 (0.2) TYP 6 5 8X 0.5 2X 2 20.1 1 10 10X PIN 1 ID (OPTIONAL) 10X 0.5 0.4 0.3 0.2 0.1 0.05 C A C B 4214926/A 07/2014 NOTES: 1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing per ASME Y14.5M. 2. This drawing is subject to change without notice. 3. The package thermal pad must be soldered to the printed circuit board for thermal and mechanical performance. www.ti.com EXAMPLE BOARD LAYOUT DSC0010B WSON - 0.8 mm max height PLASTIC SMALL OUTLINE - NO LEAD (1.2) 10X (0.65) SYMM 10 1 10X (0.25) SYMM (2) (0.75) TYP 8X (0.5) 5 ( 0.2) TYP VIA 6 (0.35) TYP (2.75) LAND PATTERN EXAMPLE SCALE:20X 0.07 MIN ALL AROUND 0.07 MAX ALL AROUND SOLDER MASK OPENING METAL SOLDER MASK OPENING NON SOLDER MASK DEFINED (PREFERRED) METAL UNDER SOLDER MASK SOLDER MASK DEFINED SOLDER MASK DETAILS 4214926/A 07/2014 NOTES: (continued) 4. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature number SLUA271 (www.ti.com/lit/slua271). www.ti.com EXAMPLE STENCIL DESIGN DSC0010B WSON - 0.8 mm max height PLASTIC SMALL OUTLINE - NO LEAD 10X (0.65) SYMM METAL TYP 10X (0.25) (0.55) SYMM (0.89) 8X (0.5) (1.13) (2.75) SOLDER PASTE EXAMPLE BASED ON 0.125 mm THICK STENCIL EXPOSED PAD 84% PRINTED SOLDER COVERAGE BY AREA SCALE:25X 4214926/A 07/2014 NOTES: (continued) 5. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate design recommendations. www.ti.com IMPORTANT NOTICE Texas Instruments Incorporated (TI) reserves the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest issue. 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