VIN
18
3
2
5
47
6
RF
1 k:RL
100:
RG
100:V-
V+
VG
-0.5 0 0.5 1 1.5
VG (V)
GAIN (dB)
2
-90
-70
-50
30
-30
-10
10
0
2
4
12
6
8
10
GAIN (V/V)
11
9
7
5
3
1
dB
V/V
125°C
25°C
-55°C
125°C
25°C
-55°C
LMH6505
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SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
LMH6505 Wideband, Low Power, Linear-in-dB, Variable Gain Amplifier
Check for Samples: LMH6505
Near ideal input characteristics (i.e. low input bias
1FEATURES current, low offset, low pin 3 resistance) enable the
2• VS= ±5V, TA= 25°C, RF=1kΩ, RG= 100Ω, RL=device to be easily configured as an inverting
100Ω, AV= AVMAX = 9.4 V/V, Typical Values amplifier as well.
Unless Specified. To provide ease of use when working with a single
•−3 dB BW 150 MHz supply, the VGrange is set to be from 0V to +2V
• Gain Control BW 100 MHz relative to the ground pin potential (pin 4). VGinput
impedance is high in order to ease drive requirement.
• Adjustment Range (<10 MHz) 80 dB In single supply operation, the ground pin is tied to a
• Gain Matching (Limit) ±0.50 dB "virtual" half supply.
• Supply Voltage Range 7V to 12V The LMH6505’s gain control is linear in dB for a large
• Slew Rate (Inverting) 1500 V/μsportion of the total gain control range from 0 dB down
• Supply Current (No Load) 11 mA to −85 dB at 25°C, as shown below. This makes the
device suitable for AGC applications. For linear gain
• Linear Output Current ±60 mA control applications, see the LMH6503 datasheet.
• Output Voltage Swing ±2.4V The LMH6505 is available in either the 8-Pin SOIC or
• Input Noise Voltage 4.4 nV/√Hz the 8-Pin VSSOP package. The combination of
• Input Noise Current 2.6 pA/√Hz minimal external components and small outline
• THD (20 MHz, RL= 100Ω, VO=2VPP)−45 dBc packages allows the LMH6505 to be used in space-
constrained applications.
APPLICATIONS
• Variable Attenuator
• AGC
• Voltage Controlled Filter
• Video Imaging Processing
DESCRIPTION
The LMH6505 is a wideband DC coupled voltage
controlled gain stage followed by a high speed
current feedback operational amplifier which can
directly drive a low impedance load. The gain
adjustment range is 80 dB for up to 10 MHz which is
accomplished by varying the gain control input Figure 1. Gain vs. VG
voltage, VG.
Maximum gain is set by external components, and
the gain can be reduced all the way to cutoff. Power Typical Application
consumption is 110 mW with a speed of 150 MHz
and a gain control bandwidth (BW) of 100 MHz.
Output referred DC offset voltage is less than 55 mV
over the entire gain control voltage range. Device-to-
device gain matching is within ±0.5 dB at maximum
gain. Furthermore, gain is tested and ensured over a
wide range. The output current feedback op amp
allows high frequency large signals (Slew Rate =
1500 V/μs) and can also drive a heavy load current
(60 mA) ensured.
Figure 2. AVMAX = 9.4 V/V
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2005–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings(1)(2)
ESD Tolerance (3)
Human Body Model 2000V
Machine Model 200V
Input Current ±10 mA
Output Current (4) 120 mA
Supply Voltages (V+- V−) 12.6V
Voltage at Input/ Output pins V++0.8V, V−−0.8V
Storage Temperature Range −65°C to 150°C
Junction Temperature 150°C
Soldering Information:
Infrared or Convection (20 sec) 235°C
Wave Soldering (10 sec) 260°C
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For ensured specifications, see the Electrical
Characteristics.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) Human Body Model, applicable std. MIL-STD-883, Method 3015.7. Machine Model, applicable std. JESD22-A115-A (ESD MM std. of
JEDEC). Field-Induced Charge-Device Model, applicable std. JESD22-C101-C (ESD FICDM std. of JEDEC).
(4) The maximum output current (IOUT) is determined by device power dissipation limitations or value specified, whichever is lower.
Operating Ratings(1)
Supply Voltages (V+- V−) 7V to 12V
Temperature Range (2) −40°C to +85°C
Thermal Resistance: (θJC) (θJA)
8 -Pin SOIC 60 165
8-Pin VSSOP 65 235
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For ensured specifications, see the Electrical
Characteristics.
(2) The maximum power dissipation is a function of TJ(MAX),θJA. The maximum allowable power dissipation at any ambient temperature is
PD= (TJ(MAX) – TA)/ θJA. All numbers apply for packages soldered directly onto a PC Board.
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+0.2 VDC
+VIN
RF IN
RG
100:
R1
50:
C1
0.01 PF
+5V
-5V
1V DC
RT
50:
RP1
10 k:
RF
1 k:
ROUT
50:
RL
50:
VG
+
-
LMH6505
PORT 1
PORT 2
LMH6505
www.ti.com
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
Electrical Characteristics(1)
Unless otherwise specified, all limits are ensured for TJ= 25°C, VS= ±5V, AVMAX = 9.4 V/V, RF= 1 kΩ, RG= 100Ω, VIN =
±0.1V, RL= 100Ω, VG= +2V. Boldface limits apply at the temperature extremes.
Symbol Parameter Conditions Min Typ Max Units
(2) (3) (2)
Frequency Domain Response
BW −3 dB Bandwidth VOUT < 1 VPP 150 MHz
VOUT < 4 VPP, AVMAX = 100 38
GF Gain Flatness VOUT < 1 VPP 40 MHz
0.9V ≤VG≤2V, ±0.2 dB
Att Range Flat Band (Relative to Max Gain) ±0.2 dB Flatness, f < 30 MHz 26 dB
Attenuation Range (4) ±0.1 dB Flatness, f < 30 MHz 9.5
BW Gain control Bandwidth VG= 1V (5) 100 MHz
Control
CT (dB) Feed-through VG= 0V, 30 MHz −51 dB
(Output/Input)
GR Gain Adjustment Range f < 10 MHz 80 dB
f < 30 MHz 71
Time Domain Response
tr, tfRise and Fall Time 0.5V Step 2.1 ns
OS % Overshoot 10 %
SR Slew Rate (6) Non Inverting 900 V/μs
Inverting 1500
(1) Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very
limited self-heating of the device such that TJ= TA. No ensured specification of parametric performance is indicated in the Electrical
Tables under conditions of internal self-heating where TJ> TA.
(2) Limits are 100% production tested at 25°C. Limits over the operating temperature range are ensured through correlations using the
Statistical Quality Control (SQC) method.
(3) Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary
over time and will also depend on the application and configuration. The typical values are not tested and are not ensured on shipped
production material.
(4) Flat Band Attenuation (Relative To Max Gain) Range Definition: Specified as the attenuation range from maximum which allows gain
flatness specified (either ±0.2 dB or ±0.1 dB), relative to AVMAX gain. For example, for f < 30 MHz, here are the Flat Band Attenuation
ranges:
±0.2 dB: 19.7 dB down to -6.3 dB = 26 dB range
±0.1 dB: 19.7 dB down to 10.2 dB = 9.5 dB range
(5) Gain control frequency response schematic:
(6) Slew rate is the average of the rising and falling slew rates.
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LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
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Electrical Characteristics(1) (continued)
Unless otherwise specified, all limits are ensured for TJ= 25°C, VS= ±5V, AVMAX = 9.4 V/V, RF= 1 kΩ, RG= 100Ω, VIN =
±0.1V, RL= 100Ω, VG= +2V. Boldface limits apply at the temperature extremes.
Symbol Parameter Conditions Min Typ Max Units
(2) (3) (2)
Distortion & Noise Performance
HD2 2nd Harmonic Distortion 2VPP, 20 MHz −47
HD3 3rd Harmonic Distortion –61 dBc
THD Total Harmonic Distortion −45
En tot Total Equivalent Input Noise f > 1 MHz, RSOURCE = 50Ω4.4 nV/√Hz
INInput Noise Current f > 1 MHz 2.6 pA/√Hz
DG Differential Gain f = 4.43 MHz, RL= 100Ω0.30 %
DP Differential Phase 0.15 deg
DC & Miscellaneous Performance
GACCU Gain Accuracy VG= 2.0V 0 ±0.50 dB
(See Application Information)0.8V < VG< 2V +0.1/−0.53 +4.3/−3.9
G Match Gain Matching VG= 2.0V — ±0.50 dB
(See Application Information)0.8V < VG< 2V — +4.2/−4.0
K Gain Multiplier 0.890 0.940 0.990 V/V
(See Application Information)0.830 1.04
VIN NL Input Voltage Range RGOpen ±3 V
VIN L RG= 100Ω±0.60 ±0.74
±0.50
IRG_MAX RGCurrent Pin 3 ±6.0 ±7.4 mA
±5.0
IBIAS Bias Current Pin 2 (7) −0.6 −2.5 µA
−2.6
TC IBIAS Bias Current Drift Pin 2 (8) 1.28 nA/°C
RIN Input Resistance Pin 2 7 MΩ
CIN Input Capacitance Pin 2 2.8 pF
IVG VGBias Current Pin 1, VG= 2V (7) 0.9 µA
TC IVG VGBias Drift Pin 1 (8) 10 pA/°C
RVG VGInput Resistance Pin 1 25 MΩ
CVG VGInput Capacitance Pin 1 2.8 pF
VOUT L Output Voltage Range RL= 100Ω±2.1 ±2.4
±1.9 V
VOUT NL RL= Open ±3.1
ROUT Output Impedance DC 0.12 Ω
IOUT Output Current VOUT = ±4V from Rails ±60 ±80 mA
±40
VO OFFSET Output Offset Voltage 0V < VG< 2V ±10 ±55 mV
±70
+PSRR +Power Supply Rejection Ratio Input Referred, 1V change, VG= –65 –72 dB
(9) 2.2V
−PSRR −Power Supply Rejection Ratio Input Referred, 1V change, VG= –65 –75 dB
(9) 2.2V
ISSupply Current No Load 9.5 11 14 mA
7.5 16
(7) Positive current corresponds to current flowing into the device.
(8) Drift is determined by dividing the change in parameter distribution at temperature extremes by the total temperature change.
(9) +PSRR definition: [|ΔVOUT/ΔV+| / AV], −PSRR definition: [|ΔVOUT/ΔV−| / AV] with 0.1V input voltage. ΔVOUT is the change in output
voltage with offset shift subtracted out.
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-7
-6
-5
-4
-3
-2
-1
0
1
GAIN (dB)
f (10 MHz/DIV)
-240
-200
-160
-120
-80
-40
0
40
80
PHASE (°)
2V
GAIN
PHASE
1V
0.8V
0
AVMAX = 100V/V
RF = 2.32 k:
RG = 18:
PIN = -24 dBm,
-14
-12
-10
-8
-6
-4
-2
0
2
GAIN (dB)
f (20 MHz/DIV)
-350
-300
-250
-200
-150
-100
-50
0
50
GAIN
PHASE
4 VPP
2 VPP
1 VPP
PHASE (°)
0
-6
-5
-4
-3
-2
-1
0
1
2
3
GAIN (dB)
f (50 MHz/DIV)
-400
-350
-300
-250
-200
-150
-100
-50
0
50
GAIN
PHASE
AVMAX = 2V/V
RF = 1 k:
RG = 510:
PIN = 4 dBm VG = 2V
VG = 1V
VG = 0.8V
VG = 0.8V
VG = 1V
VG = 2V
PHASE (°)
0
-9
-8
-7
-6
-5
-4
-3
-2
-1
0
1
GAIN (dB)
f (50 MHz/DIV) -360
-320
-280
-240
-200
-160
-120
-80
-40
0
40
PHASE (°)
GAIN
PHASE
4 VPP
2 VPP
1 VPP
0
2 VPP 4 VPP
1 VPP
1M 10M 100M 1G
FREQUENCY (Hz)
-9
-8
-7
-6
-5
-4
-3
-2
-1
0
1
GAIN (dB)
GAIN
PHASE
PIN = -22 dBm
85°C
-40°C
25°C
85°C
-40°C
25°C
-350
-300
-250
-200
-150
-100
-50
0
50
100
150
PHASE (°)
1M 10M 100M 1G
FREQUENCY (Hz)
-9
-8
-7
-6
-5
-4
-3
-2
-1
0
1
GAIN (dB)
GAIN
PHASE
PIN = -22 dBm
VG = 2V
-350
-300
-250
-200
-150
-100
-50
0
50
100
150
PHASE (°)
VG = 0.7V
VG = 0.9V
VG = 1V
VG = 0.9V VG = 2V
VG = 1V
VG = 0.7V
LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
www.ti.com
Typical Performance Characteristics
Unless otherwise specified: VS= ±5V, TA= 25°C, VG= VGMAX, RF= 1 kΩ, RG= 100Ω, VIN = 0.1V, input terminated in 50Ω. RL
= 100Ω, Typical values.
Frequency Response Over Temperature Frequency Response for Various VG
Gain/Phase normalized to low frequency value at 25°C. Gain/Phase normalized to low frequency value at each setting.
Figure 4. Figure 5.
Frequency Response (AVMAX = 2) Inverting Frequency Response
Gain/Phase normalized to low frequency value at each setting.
Figure 6. Figure 7.
Frequency Response for Various VG(AVMAX = 100)
(Large Signal) Frequency Response for Various Amplitudes
Gain/Phase normalized to low frequency value at each setting. Gain/Phase normalized to low frequency value at each setting.
Figure 8. Figure 9.
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2.5 3 3.5 4 4.5 5 5.5
±SUPPLY VOLTAGE
0
2
4
6
8
10
12
AVMAX (V/V)
6
-40°C
25°C
85°C
0
100 10k 1M 100M
FREQUENCY (Hz)
-80
-40
PSRR (dB)
10M
100k
1k
-20
-30
-50
-60
-70
-10
+PSRR
-PSRR
3 3.5 4 4.5 5 5.5 6
0
2
4
6
8
10
12
14
16
18
20
IS (mA)
±SUPPLY VOTLAGE (V)
25°C
85°C
-40°C
RL = OPEN
2 3 4 5 6
-0.8
-0.7
-0.6
-0.5
-0.4
IB (PA)
±SUPPLY VOLTAGE (V)
85°C 25°C
-40°C
3 3.5 4 4.5 5 5.5 6
0
2
4
6
8
10
12
14
16
18
20
IS (mA)
±SUPPLY VOTLAGE (V)
25°C
85°C
-40°C
VG = VG_MIN
RL = OPEN
100k 1G
FREQUENCY (Hz)
|S21| (dB)
100M
10M
1M
-40
-20
-5
15
5
-15
-35
10
0
-10
-25
-30
-320
-160
-40
120
40
-120
-280
80
0
-80
-200
-240
ANGLE S21 (°)
40°C
25°C 85°C
40°C
25°C
85°C
VG (AC) = -13.7 dBm
VIN = 0.2V (DC)
VG = 0.98 AVERAGE
MAGNITUDE
ANGLE
LMH6505
www.ti.com
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
Typical Performance Characteristics (continued)
Unless otherwise specified: VS= ±5V, TA= 25°C, VG= VGMAX, RF= 1 kΩ, RG= 100Ω, VIN = 0.1V, input terminated in 50Ω. RL
= 100Ω, Typical values.
Gain Control Frequency Response ISvs. VS
See Electrical Characteristics Note (5).
Figure 10. Figure 11.
ISvs. VSInput Bias Current vs. VS
Figure 12. Figure 13.
PSRR AVMAX vs. Supply Voltage
See Electrical Characteristics Note (9)
Figure 14. Figure 15.
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00.5 11.5 2 2.5
VG (V)
-15
-10
-5
0
5
10
VO_OFFSET (mV)
25°C
85°C
-40°C
0 0.5 1 1.5 2 2.5
VG (V)
0
5
10
15
20
25
30
VO_OFFSET (mV)
85°C
85°C
-40°C
25°C
-0.5 0 0.5 1 1.5
VG (V)
GAIN (dB)
2
-90
-70
-50
30
-30
-10
10
0
2
4
12
6
8
10
GAIN (V/V)
11
9
7
5
3
1
dB
V/V
125°C
25°C
-55°C
125°C
25°C
-55°C
-1.5 -1 -0.5 0 0.5 1 1.5
+8
+6
+4
+2
0
-2
-4
-6
-8
-10
VIN (V)
IRG
(mA)
100k 1M 10M 100M 1G
FREQUENCY (Hz)
-100
-80
-60
-40
-20
0
20
40
60
GAIN (dB)
AVMAX = 100 V/V
AVMAX = 10 V/V
AVMAX = 2 V/V
0.01
0.1
1
10
100
OVER TEMP CHANGE (dB)
00.5 1 1.5 2
VG (V)
TEMP RANGE: -55°C TO 125°C
|GAIN(COLD) ± GAIN (HOT)
LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
www.ti.com
Typical Performance Characteristics (continued)
Unless otherwise specified: VS= ±5V, TA= 25°C, VG= VGMAX, RF= 1 kΩ, RG= 100Ω, VIN = 0.1V, input terminated in 50Ω. RL
= 100Ω, Typical values.
Feed through Isolation for Various AVMAX Gain Variation Over entire Temp Range vs. VG
Figure 16. Figure 17.
IRG vs. VIN Gain vs. VG
See Electrical Characteristics Note (7).
Figure 18. Figure 19.
Output Offset Voltage vs. VG(Typical Unit #1) Output Offset Voltage vs. VG(Typical Unit #2)
Figure 20. Figure 21.
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11k 1M
10k100 10M
FREQUENCY (Hz)
100k10
10
100
1000
10000
eno (nV/
Hz)
VG_MAX
VG_MID
VG_MIN
AVMAX = 2
RG = 510:
RSOURCE = 50:
11k 1M
10k100 10M
FREQUENCY (Hz)
100k10
1
10
100
1000
1
10
100
1000
eni (nV/
Hz)
Ini (pA/
Hz)
VOLTAGE
CURRENT
1100 100k 1M
1k
10 10M
FREQUENCY (Hz)
10k
10
100
1000
10000
eno (nV/
Hz)
VG_MAX
VG_MID
VG_MIN
RSOURCE - 50:
1100 100k
10
100
10000
100000
1M
1k
10 10M
FREQUENCY (Hz)
10k
1000
eno (nV/ Hz)
VG_MAX
VG_MID
VG_MIN
AVMAX = 100
RF = 2.4 k:
RG = 22:
RSOURCE = 50:
VG (V)
0 0.5 1 1.5 2 2.5
0
5
10
15
20
25
30
VO_OFFSET (mV)
25°C
85°C
-40°C
25°C
-55 -35 -25 515 25 55
0
2
8
10
12
14
16
20
22
24
RELATIVE FREQUENCY (%)
OFFSET VOLTAGE (mV)
6
-45 -15 -5 35 45
18
4
LMH6505
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SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
Typical Performance Characteristics (continued)
Unless otherwise specified: VS= ±5V, TA= 25°C, VG= VGMAX, RF= 1 kΩ, RG= 100Ω, VIN = 0.1V, input terminated in 50Ω. RL
= 100Ω, Typical values.
Output Offset Voltage vs. VG(Typical Unit #3) Distribution of Output Offset Voltage
Figure 22. Figure 23.
Output Noise Density vs. Frequency Output Noise Density vs. Frequency
Figure 24. Figure 25.
Output Noise Density vs. Frequency Input Referred Noise Density vs. Frequency
Figure 26. Figure 27.
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 9
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-10 -5 0 510 15 20
POUT (dBm)
-20
-30
-40
-50
-60
-70
-80
-90
-100
THD (dBc)
VG = VG_MAX
100 kHz
20 MHz
-10 -5 0 510 15 20
POUT (dBm)
0
-10
-20
-30
-40
-50
-60
-70
THD (dBc)
1 MHz
20 MHz
VG = VGMID = a1V
-10 -5 0 5 10 15 20
-20
-30
-40
-50
-60
-70
-80
-90
-100
-110
-120
HD (dBc)
POUT (dBm)
HD3, 100 kHz
HD2, 100 kHz
HD2, 20 MHz
VG = VGMAX
HD3, 20 MHz
100k 1M 10M 100M
FREQUENCY (Hz)
-100
-90
-80
-70
-60
-50
-40
-30
HD (dBc)
THD
HD3
HD2
VG = VG_MAX
VOUT = 1 VPP
IOUT (mA)
0
1
2
3
4
5
VOUT FROM V- (V)
020 40 60 80 100 120
85°C
-40°C
25°C
-40°C
25°C
85°C
IOUT (mA)
0
1
2
3
4
5
VOUT FROM V+ (V)
020 40 60 80 100 120
85°C -40°C
25°C
-40°C
85°C
LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
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Typical Performance Characteristics (continued)
Unless otherwise specified: VS= ±5V, TA= 25°C, VG= VGMAX, RF= 1 kΩ, RG= 100Ω, VIN = 0.1V, input terminated in 50Ω. RL
= 100Ω, Typical values.
Output Voltage vs. Output Current (Sinking) Output Voltage vs. Output Current (Sourcing)
Figure 28. Figure 29.
Distortion vs. Frequency HD vs. POUT
Figure 30. Figure 31.
THD vs. POUT THD vs. POUT
Figure 32. Figure 33.
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SS REF
5 ns/DIV
LS REF
0.5 VPP SMALL SIGNAL
4 VPP LARGE SIGNAL
SS REF
5 ns/DIV
LS REF
0.5 VPP SMALL SIGNAL
4 VPP LARGE SIGNAL
VG = VG_MID
00.5 1 1.5 2 2.5 3
VG (V)
820
840
860
880
900
920
940
IG (nA)
-1.4 -0.2 0.2 0.6 1 1.4
DG (%)
VOUT DC (V)
-1 -0.6
-0.1
0
0.2
0.3
0.4
0.5
0.1
-0.15
-0.1
0
0.05
0.1
0.15
-0.05
DP (°)
DP
DG
f = 4.43 MHz
RL = 100:
VG = VGMAX
-5 0 5 10 15 20
0
-10
-20
-30
-40
-50
-60
-70
-80
THD (dBc)
GAIN (dB)
20 MHz
100 kHz
VOUT = 1 VPP
VG VARIED 2 MHz
-15 -10 -5 0 5 10 15 20
0
THD (dBc)
GAIN (dB)
-10
-20
-30
-40
-50
-60
-70
-80
-90
20 MHz
100 kHz
2 MHz
VOUT = 0.25 VPP
VG VARIED
LMH6505
www.ti.com
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
Typical Performance Characteristics (continued)
Unless otherwise specified: VS= ±5V, TA= 25°C, VG= VGMAX, RF= 1 kΩ, RG= 100Ω, VIN = 0.1V, input terminated in 50Ω. RL
= 100Ω, Typical values.
THD vs. Gain THD vs. Gain
Figure 34. Figure 35.
Differential Gain & Phase VGBias Current vs. VG
Figure 36. Figure 37.
Step Response Plot Step Response Plot
Figure 38. Figure 39.
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 11
Product Folder Links: LMH6505
t (10 ns/DIV)
0
0.5
1
1.5
2
2.5
VG (V)
0
2
4
9
10
8
7
1
3
5
6
GAIN (V/V)
GAIN
VG
VIN = 0.3V
LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
www.ti.com
Typical Performance Characteristics (continued)
Unless otherwise specified: VS= ±5V, TA= 25°C, VG= VGMAX, RF= 1 kΩ, RG= 100Ω, VIN = 0.1V, input terminated in 50Ω. RL
= 100Ω, Typical values. Gain vs. VGStep
Figure 40.
12 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LMH6505
VIN (MAX) = IRGMAX · RG
AVMAX = RF
RG· K
RX
5V
0.1 µF
6.8 µF
OUTPUT
6.8 µF
0.1 µF
I-
-VCC
VG
GND
INPUT
SIGNAL
VIN
RGVO
+VCC
RF
RO
RIN
RG
100:
50:
50:50:
-5V
1 k:
X1
CFA
+
-
GAIN
CONTROL
MULT
+-
+
-
LMH6505
www.ti.com
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
APPLICATION INFORMATION
GENERAL DESCRIPTION
The key features of the LMH6505 are:
• Low power
• Broad voltage controlled gain and attenuation range (from AVMAX down to complete cutoff)
• Bandwidth independent, resistor programmable gain range (RG)
• Broad signal and gain control bandwidths
• Frequency response may be adjusted with RF
• High impedance signal and gain control inputs
The LMH6505 combines a closed loop input buffer (“X1” Block in Figure 41), a voltage controlled variable gain
cell (“MULT” Block) and an output amplifier (“CFA” Block). The input buffer is a transconductance stage whose
gain is set by the gain setting resistor, RG. The output amplifier is a current feedback op amp and is configured
as a transimpedance stage whose gain is set by, and is equal to, the feedback resistor, RF. The maximum gain,
AVMAX, of the LMH6505 is defined by the ratio: K · RF/RGwhere “K” is the gain multiplier with a nominal value of
0.940. As the gain control input (VG) changes over its 0 to 2V range, the gain is adjusted over a range of about
80 dB relative to the maximum set gain.
Figure 41. LMH6505 Typical Application and Block Diagram
SETTING THE LMH6505 MAXIMUM GAIN
(1)
Although the LMH6505 is specified at AVMAX = 9.4 V/V, the recommended AVMAX varies between 2 and 100.
Higher gains are possible but usually impractical due to output offsets, noise and distortion. When varying AVMAX
several tradeoffs are made:
RG: determines the input voltage range
RF: determines overall bandwidth
The amount of current which the input buffer can source/sink into RGis limited and is given in the IRG_MAX
specification. This sets the maximum input voltage:
(2)
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 13
Product Folder Links: LMH6505
A(V/V) = K x x
RF
RG
1
1 + e
N - VG
VC
LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
www.ti.com
As the IRG_MAX limit is approached with increasing the input voltage or with the lowering of RG, the device's
harmonic distortion will increase. Changes in RFwill have a dramatic effect on the small signal bandwidth. The
output amplifier of the LMH6505 is a current feedback amplifier (CFA) and its bandwidth is determined by RF. As
with any CFA, doubling the feedback resistor will roughly cut the bandwidth of the device in half.
For more about CFAs, see the basic tutorial, OA-20, Current Feedback Myths Debunked, (literature number
SNOA376), or a more rigorous analysis, OA-13, Current Feedback Amplifier Loop Gain Analysis and
Performance Enhancements, (literature number SNOA366).
OTHER CONFIGURATIONS
1. Single Supply Operation
The LMH6505 can be configured for use in a single supply environment. Doing so requires the following:
(a) Bias pin 4 and RGto a “virtual half supply” somewhere close to the middle of V+and V−range. The other
end of RGis tied to pin 3. The “virtual half supply” needs to be capable of sinking and sourcing the
expected current flow through RG.
(b) Ensure that VGcan be adjusted from 0V to 2V above the “virtual half supply”.
(c) Bias the input (pin 2) to make sure that it stays within the range of 2V above V−to 2V below V+. See the
Input Voltage Range specification in the Electrical Characteristics table. This can be accomplished by
either DC biasing the input and AC coupling the input signal, or alternatively, by direct coupling if the
output of the driving stage is also biased to half supply.
Arranged this way, the LMH6505 will respond to the current flowing through RG. The gain control relationship
will be similar to the split supply arrangement with VGmeasured with reference to pin 4. Keep in mind that
the circuit described above will also center the output voltage to the “virtual half supply voltage.”
2. Arbitrarily Referenced Input Signal
Having a wide input voltage range on the input (pin 2) (±3V typical), the LMH6505 can be configured to
control the gain on signals which are not referenced to ground (e.g. Half Supply biased circuits). This node
will be called the “reference node”. In such cases, the other end of RGwhich is the side not tied to pin 3 can
be tied to this reference node so that RGwill “look at” the difference between the signal and this reference
only. Keep in mind that the reference node needs to source and sink the current flowing through RG.
GAIN ACCURACY
Gain accuracy is defined as the actual gain compared against the theoretical gain at a certain VG, the results of
which are expressed in dB. (See Figure 42).
Theoretical gain is given by:
where
• K = 0.940 (nominal) N = 1.01V
• VC= 79 mV at room temperature (3)
For a VGrange, the value specified in the tables represents the worst case accuracy over the entire range. The
"Typical" value would be the difference between the "Typical Gain" and the "Theoretical Gain." The "Max" value
would be the worst case difference between the actual gain and the "Theoretical Gain" for the entire population.
GAIN MATCHING
As Figure 42 shows, gain matching is the limit on gain variation at a certain VG, expressed in dB, and is specified
as "±Max" only. There is no "Typical." For a VGrange, the value specified represents the worst case matching
over the entire range. The "Max" value would be the worst case difference between the actual gain and the
typical gain for the entire population.
14 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LMH6505
MAXIMUM GAIN = 1 + R2
R1
·RF
RGK
·
2
VIN
R2
VO
VG
RG
RC
R1
25:
+
-
LMH6624 LMH6505
3
1
6
7
4
RF
GAIN (dB)
VG (V)
PARAMETER:
GAIN ACCURACY (TYPICAL) = B-C
GAIN ACCURACY (+MAX) = D-C
GAIN ACCURACY (-MAX) = A-C
GAIN MATCHING (+MAX) = D-B
GAIN MATCHING (-MAX) = A-B
THEORETICAL GAIN
MAX GAIN LIMIT
MIN GAIN LIMIT
TYPICAL GAIN
D
C
B
A
LMH6505
www.ti.com
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
Figure 42. LMH6505 Gain Accuracy & Gain Matching Defined
GAIN PARTITIONING
If high levels of gain are needed, gain partitioning should be considered:
Figure 43. Gain Partitioning
The maximum gain range for this circuit is given by the following equation:
(4)
The LMH6624 is a low noise wideband voltage feedback amplifier. Setting R2at 909Ωand R1at 100Ωproduces
a gain of 20 dB. Setting RFat 1000Ωas recommended and RGat 50Ω, produces a gain of about 26 dB in the
LMH6505. The total gain of this circuit is therefore approximately 46 dB. It is important to understand that when
partitioning to obtain high levels of gain, very small signal levels will drive the amplifiers to full scale output. For
example, with 46 dB of gain, a 20 mV signal at the input will drive the output of the LMH6624 to 200 mV and the
output of the LMH6505 to 4V. Accordingly, the designer must carefully consider the contributions of each stage
to the overall characteristics. Through gain partitioning the designer is provided with an opportunity to optimize
the frequency response, noise, distortion, settling time, and loading effects of each amplifier to achieve improved
overall performance.
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 15
Product Folder Links: LMH6505
-80 -60 -40 -20 0 20
-0.5
0
0.5
1
1.5
2
2.5
3
3.5
4
4.5
DELTA AV (dB)
AV (dB)
4.5 mA SOURCING
4.5 mA SINKING
LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
www.ti.com
LMH6505 GAIN CONTROL RANGE AND MINIMUM GAIN
Before discussing Gain Control Range, it is important to understand the issues which limit it. The minimum gain
of the LMH6505 is theoretically zero, but in practical circuits it is limited by the amount of feedthrough, here
defined as the gain when VG= 0V. Capacitive coupling through the board and package, as well as coupling
through the supplies, will determine the amount of feedthrough. Even at DC, the input signal will not be
completely rejected. At high frequencies feedthrough will get worse because of its capacitive nature. At
frequencies below 10 MHz, the feed through will be less than −60 dB and therefore, it can be said that with
AVMAX = 20 dB, the gain control range is 80 dB.
LMH6505 GAIN CONTROL FUNCTION
In the plot, Gain vs. VG(Figure 19), we can see the gain as a function of the control voltage. The “Gain (V/V)”
plot, sometimes referred to as the S-curve, is the linear (V/V) gain. This is a hyperbolic tangent relationship and
is given by Equation 3. The “Gain (dB)” plots the gain in dB and is linear over a wide range of gains. Because of
this, the LMH6505 gain control is referred to as “linear-in-dB.”
For applications where the LMH6505 will be used at the heart of a closed loop AGC circuit, the S-curve control
characteristic provides a broad linear (in dB) control range with soft limiting at the highest gains where large
changes in control voltage result in small changes in gain. For applications requiring a fully linear (in dB) control
characteristic, use the LMH6505 at half gain and below (VG≤1V).
GAIN STABILITY
The LMH6505 architecture allows complete attenuation of the output signal from full gain to complete cutoff. This
is achieved by having the gain control signal VG“throttle” the signal which gets through to the final stage and
which results in the output signal. As a consequence, the RGpin's (pin 3) average current (DC current) influences
the operating point of this “throttle” circuit and affects the LMH6505's gain slightly. Figure 44 below, shows this
effect as a function of the gain set by VG.
Figure 44. LMH6505 Gain Variation over RGDC Current Capability vs. Gain
This plot shows the expected gain variation for the maximum RGDC current capability (±4.5 mA). For example,
with gain (AV) set to −60 dB, if the RGpin DC current is increased to 4.5 mA sourcing, one would expect to see
the gain increase by about 3 dB (to −57 dB). Conversely, 4.5 mA DC sinking current through RGwould increase
gain by 1.75 dB (to −58.25 dB). As you can see from Figure 44 above, the effect is most pronounced with
reduced gain and is limited to less than 3.75 dB variation maximum.
If the application is expected to experience RGDC current variation and the LMH6505 gain variation is beyond
acceptable limits, please refer to the LMH6502 (Differential Linear in dB variable gain amplifier) datasheet
instead at http://www.ti.com/lit/gpn/LMH6502.
16 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LMH6505
2
OUTPUT
VG
3
1
6
7
4RT
RI
RS
RG
CO
RO
ZO
ZO
RF
SIGNAL
INPUT
LMH6505
+
-
RF
RG·K
-
AVMAX =
2
VO
VG
RG
25:LMH6505
3
1
6
7
4
RF
VIN
LMH6505
www.ti.com
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
AVOIDING OVERDRIVE OF THE LMH6505 GAIN CONTROL INPUT
There is an additional requirement for the LMH6505 Gain Control Input (VG): VGmust not exceed +2.3V (with
±5V supplies). The gain control circuitry may saturate and the gain may actually be reduced. In applications
where VGis being driven from a DAC, this can easily be addressed in the software. If there is a linear loop
driving VG, such as an AGC loop, other methods of limiting the input voltage should be implemented. One simple
solution is to place a 2.2:1 resistive divider on the VGinput. If the device driving this divider is operating off of
±5V supplies as well, its output will not exceed 5V and through the divider VGcan not exceed 2.3V.
IMPROVING THE LMH6505 LARGE SIGNAL PERFORMANCE
Figure 45 illustrates an inverting gain scheme for the LMH6505.
Figure 45. Inverting Amplifier
The input signal is applied through the RGresistor. The VIN pin should be grounded through a 25Ωresistor. The
maximum gain range of this configuration is given in the following equation:
(5)
The inverting slew rate of the LMH6505 is much higher than that of the non-inverting slew rate. This ≈2X
performance improvement comes about because in the non-inverting configuration the slew rate of the overall
amplifier is limited by the input buffer. In the inverting circuit, the input buffer remains at a fixed voltage and does
not affect slew rate.
TRANSMISSION LINE MATCHING
One method for matching the characteristic impedance of a transmission line is to place the appropriate resistor
at the input or output of the amplifier. Figure 46 shows a typical circuit configuration for matching transmission
lines.
Figure 46. Transmission Line Matching
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 17
Product Folder Links: LMH6505
2
VO
VG
RG
R2
10 k:
LMH6505
3
1
6
7
4RF
VIN
+5V
-5V
R1
10 k:R4
10 k:
R3
10 k:
0.1 µF 0.1 µF
+5V
-5V
LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
www.ti.com
The resistors RS, RI, RO, and RTare equal to the characteristic impedance, ZO, of the transmission line or cable.
Use COto match the output transmission line over a greater frequency range. It compensates for the increase of
the op amp’s output impedance with frequency.
MINIMIZING PARASITIC EFFECTS ON SMALL SIGNAL BANDWIDTH
The best way to minimize parasitic effects is to use surface mount components and to minimize lead lengths and
component distance from the LMH6505. For designs utilizing through-hole components, specifically axial
resistors, resistor self-capacitance should be considered. For example, the average magnitude of parasitic
capacitance of RN55D 1% metal film resistors is about 0.15 pF with variations of as much as 0.1 pF between
lots. Given the LMH6505’s extended bandwidth, these small parasitic reactance variations can cause
measurable frequency response variations in the highest octave. We therefore recommend the use of surface
mount resistors to minimize these parasitic reactance effects.
RECOMMENDATIONS
Here are some recommendations to avoid problems and to get the best performance:
• Do not place a capacitor across RF. However, an appropriately chosen series RC combination can be used to
shape the frequency response.
• Keep traces connecting RFseparated and as short as possible.
• Place a small resistor (20-50Ω) between the output and CL.
• Cut away the ground plane, if any, under RG.
• Keep decoupling capacitors as close as possible to the LMH6505.
• Connect pin 2 through a minimum resistance of 25Ω.
ADJUSTING OFFSETS AND DC LEVEL SHIFTING
Offsets can be broken into two parts: an input-referred term and an output-referred term. These errors can be
trimmed using the circuit in Figure 47. First set VGto 0V and adjust the trim pot R4to null the offset voltage at the
output. This will eliminate the output stage offsets. Next set VGto 2V and adjust the trim pot R1to null the offset
voltage at the output. This will eliminate the input stage offsets.
Figure 47. Offset Adjust Circuit
DIGITAL GAIN CONTROL
Digitally variable gain control can be easily realized by driving the LMH6505 gain control input with a digital-to-
analog converter (DAC). Figure 48 illustrates such an application. This circuit employs TI’s eight-bit DAC0830,
the LMC8101 MOS input op amp (Rail-to-Rail Input/Output), and the LMH6505 VGA. With VREF set to 2V, the
circuit provides up to 80 dB of gain control in 256 steps with up to 0.05% full scale resolution. The maximum gain
of this circuit is 20 dB.
18 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LMH6505
2
VIN VO
RG
100:
+
-
LMC8101
LMH6505
3
1
6
7
4
RF
1 k:
Io1
Io2
DIGITAL
INPUT
VREF
RFB
DAC0830
LMH6505
www.ti.com
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
Figure 48. Digital Gain Control
USING THE LMH6505 IN AGC APPLICATIONS
In AGC applications, the control loop forces the LMH6505 to have a fixed output amplitude. The input amplitude
will vary over a wide range and this can be the issue that limits dynamic range. At high input amplitudes, the
distortion due to the input buffer driving RGmay exceed that which is produced by the output amplifier driving the
load. In the plot, THD vs. Gain, total harmonic distortion (THD) is plotted over a gain range of nearly 35 dB for a
fixed output amplitude of 0.25 VPP in the specified configuration, RF=1kΩ, RG= 100Ω. When the gain is
adjusted to −15 dB (i.e. 35 dB down from AVMAX), the input amplitude would be 1.41 VPP and we can see the
distortion is at its worst at this gain. If the output amplitude of the AGC were to be raised above 0.25 VPP, the
input amplitudes for gains 40 dB down from AVMAX would be even higher and the distortion would degrade
further. It is for this reason that we recommend lower output amplitudes if wide gain ranges are desired. Using a
post-amp like the LMH6714/LMH6720/LMH6722 family or the LMH6702 would be the best way to preserve
dynamic range and yield output amplitudes much higher than 100 mVPP. Another way of addressing distortion
performance and its limitations on dynamic range, would be to raise the value of RG. Just like any other high-
speed amplifier, by increasing the load resistance, and therefore decreasing the demanded load current, the
distortion performance will be improved in most cases. With an increased RG, RFwill also have to be increased
to keep the same AVMAX and this will decrease the overall bandwidth. It may be possible to insert a series RC
combination across RFin order to counteract the negative effect on BW when a large RFis used.
AUTOMATIC GAIN CONTROL (AGC)
Fast Response AGC Loop
The AGC circuit shown in Figure 49 will correct a 6 dB input amplitude step in 100 ns. The circuit includes a two
op amp precision rectifier amplitude detector (U1 and U2), and an integrator (U3) to provide high loop gain at low
frequencies. The output amplitude is set by R9. The following are some suggestions for building fast AGC loops:
Precision rectifiers work best with large output signals. Accuracy is improved by blocking DC offsets, as shown in
Figure 49.
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 19
Product Folder Links: LMH6505
LMH6714
2
OUTPUT
20 MHz,
0.1 VPP
-5V
R1
20:
3
1
6
7
4RF
LMH6609
LMH6714
LMH6505
RG
100:
R10
500:
R9
4.22 k:
1N5712
SCHOTTKY
R2
25:
C1
1.0 µF
C2
680 pF
C3
40 pF
+
+
+
+
-
-
-
-
VIN
INCLUDES SCOPE
PROBE CAPACITANCE
R6
300:
R5
25:
R3
300:
R4
300:
R7
300:
R8
500:
U4
U3
U2
U1
LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
www.ti.com
Figure 49. Automatic Gain Control Circuit
Signal frequencies must not reach the gain control port of the LMH6505, or the output signal will be distorted
(modulated by itself). A fast settling AGC needs additional filtering beyond the integrator stage to block signal
frequencies. This is provided in Figure 49 by a simple R-C filter (R10 and C3); better distortion performance can
be achieved with a more complex filter. These filters should be scaled with the input signal frequency. Loops with
slower response time, which means longer integration time constants, may not need the R10 – C3filter.
Checking the loop stability can be done by monitoring the VGvoltage while applying a step change in input signal
amplitude. Changing the input signal amplitude can be easily done with an arbitrary waveform generator.
20 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LMH6505
LMH6505
www.ti.com
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
CIRCUIT LAYOUT CONSIDERATIONS & EVALUATION BOARDS
A good high frequency PCB layout including ground plane construction and power supply bypassing close to the
package is critical to achieving full performance. The amplifier is sensitive to stray capacitance to ground at the I-
input (pin 7) so it is best to keep the node trace area small. Shunt capacitance across the feedback resistor
should not be used to compensate for this effect. Capacitance to ground should be minimized by removing the
ground plane from under the body of RG. Parasitic or load capacitance directly on the output (pin 6) degrades
phase margin leading to frequency response peaking.
The LMH6505 is fully stable when driving a 100Ωload. With reduced load (e.g. 1k.) there is a possibility of
instability at very high frequencies beyond 400 MHz especially with a capacitive load. When the LMH6505 is
connected to a light load as such, it is recommended to add a snubber network to the output (e.g. 100Ωand 39
pF in series tied between the LMH6505 output and ground). CLcan also be isolated from the output by placing a
small resistor in series with the output (pin 6).
Component parasitics also influence high frequency results. Therefore it is recommended to use metal film
resistors such as RN55D or leadless components such as surface mount devices. High profile sockets are not
recommended.
Texas Instruments suggests the following evaluation board as a guide for high frequency layout and as an aid in
device testing and characterization:
Device Package Evaluation Board
Part Number
LMH6505 SOIC LMH730066
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 21
Product Folder Links: LMH6505
LMH6505
SNOSAT4E –DECEMBER 2005–REVISED APRIL 2013
www.ti.com
REVISION HISTORY
Changes from Revision D (April 2013) to Revision E Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 21
22 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LMH6505
PACKAGE OPTION ADDENDUM
www.ti.com 23-Aug-2017
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LMH6505MA/NOPB ACTIVE SOIC D 8 95 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 85 LMH65
05MA
LMH6505MAX/NOPB ACTIVE SOIC D 8 2500 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 85 LMH65
05MA
LMH6505MM/NOPB ACTIVE VSSOP DGK 8 1000 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 85 AZ2A
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
PACKAGE OPTION ADDENDUM
www.ti.com 23-Aug-2017
Addendum-Page 2
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LMH6505MAX/NOPB SOIC D 8 2500 330.0 12.4 6.5 5.4 2.0 8.0 12.0 Q1
LMH6505MM/NOPB VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 24-Aug-2017
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LMH6505MAX/NOPB SOIC D 8 2500 367.0 367.0 35.0
LMH6505MM/NOPB VSSOP DGK 8 1000 210.0 185.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 24-Aug-2017
Pack Materials-Page 2
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