LTC3351 Hot Swappable Supercapacitor Charger, Backup Controller and System Monitor FEATURES DESCRIPTION Integrated Hot Swap Controller with Circuit Breaker nn High Efficiency Synchronous Step-Down CC/CV Charging of One to Four Series Supercapacitors nn Step-Up Mode in Backup Provides Greater Utilization of Stored Energy in Supercapacitors nn 16-Bit ADC for Monitoring System Voltages/ Currents, Capacitance and ESR nn Programmable Undervoltage and Overvoltage Thresholds to 35V nn V : 4.5V to 35V, V IN CAP(n): Up to 5V per Capacitor, Charge/Backup Current: >10A nn Programmable Input Current Limit Prioritizes System Load Over Capacitor Charge Current nn All N-FET Charger Controller and PowerPathTM Controller nn Compact 44-Lead 4mm x 7mm QFN Package The LTC(R)3351 is a backup power controller that charges and monitors a series stack of one to four supercapacitors. The LTC3351's synchronous step-down controller drives N-channel MOSFETs for constant current/constant voltage charging with programmable input current limit. In addition, the step-down converter runs in reverse as a step-up converter to deliver power from the supercapacitor stack to the backup supply rail. Internal balancers eliminate the need for external balance resistors and each capacitor has a shunt regulator for overvoltage protection. nn The LTC3351 monitors system voltages, currents, stack capacitance and ESR which can all be read over the I2C/ SMBus port. The hot swap controller uses N-channel MOSFETs for inrush control and a low loss path from the input to the output. The ideal diode controller uses an N-channel MOSFET for a low loss power path from the supercapacitors to the output. The LTC3351 is available in a thermally enhanced low profile 44-lead 4mm x 7mmx0.75mm QFN surface mount package. APPLICATIONS Swappable PCIE Cards with NVM High Current 12V Ride-Through UPS nn Servers/Mass Storage/High Availability Systems nn nn TYPICAL APPLICATION ICHG (STEP-DOWN) All registered trademarks and trademarks are the property of their respective owners. Patents pending. IBACKUP VOUT VIN HS_GATE UV ISNSP_HS/ ISNSM ISNSP_CHG OUTFB VIN 2V/DIV VCAP > VOUT (DIRECT CONNECT) OV OUTFET TGATE Example Hot Swap from 12V VCAP < VOUT (STEP-UP) VOUT 2V/DIV SW LTC3351 I2C IIN 0.2A/DIV BGATE ICAP VCAP CAP4 CAP3 CAP2 CAP1 CAPRTN CAPFB 5ms/DIV 10F VCAP 3351 TA01 10F 10F 10F 3351 TA01a Rev. A Document Feedback For more information www.analog.com 1 LTC3351 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) CTIMER OV UV VIN HS_GATE SRC CSS TOP VIEW 44 43 42 41 40 39 38 CAP_SLCT0 1 37 ISNS_HS CAP_SLCT1 2 36 ISNSP_CHG 35 ISNSM VINGD 3 SCL 4 34 RETRYB SDA 5 33 VOUT 32 INTVCC SMBALERT 6 CAPGD 7 31 DRVCC 45 PGND VC 8 30 BGATE CAPFB 9 29 BST OUTFB 10 28 TGATE 27 SW SGND 11 26 VCC2P5 RT 12 GPI 13 25 ICAP ITST 14 24 VCAP 23 OUTFET 18 19 20 21 CAP3 CAP4 CFN 22 VCAPP5 17 CFP 16 CAP2 CAPRTN 15 CAP1 VIN, VOUT, ISNSP_HS, ISNSP_CHG, ISNSM, UV, OV, RETRYB, OUTFB........................ -0.3V to 40V VCAP........................................................... -0.3V to 22V CAP4-CAP3, CAP3-CAP2, CAP2-CAP1, CAP1-CAPRTN.................. -0.3V to 5.5V DRVCC, CAPFB, SMBALERT, CAPGD, VINGD, GPI, SDA, SCL............-0.3V to INTVCC + 0.3V BST.......................................................... -0.3V to 45.5V CAP_SLCT0, CAP_SLCT1.......... -0.3V to VCC2P5 + 0.3V BST to SW................................................. -0.3V to 5.5V ISNSP_HS to ISNSM, ISNSP_CHG to ISNSM, ICAP to VCAP.... -0.3V to 0.3V IINTVCC..................................................................100mA ICAP(1,2,3,4), ICAPRTN............................................. 600mA ICAPGD, IVINGD , ISMBALERT.......................................10mA Operating Junction Temperature Range (Notes 2, 3)......................................... -40C to 125C Storage Temperature Range................... -65C to 150C UFF PACKAGE 44-LEAD (4mm x 7mm) PLASTIC QFN TJMAX = 125C, JA = 36.4C/W, JC = 2.6C/W EXPOSED PAD (PIN 45) IS PGND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3351EUFF#PBF LTC3351EUFF#TRPBF 3351 44-Lead (4mm x 7mm) Plastic QFN -40C to 125C LTC3351IUFF#PBF LTC3351IUFF#TRPBF 3351 44-Lead (4mm x 7mm) Plastic QFN -40C to 125C Contact the factory for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Tape and reel specifications. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. 2 Rev. A For more information www.analog.com LTC3351 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TJ = 25C (Note 2). VIN = VOUT = 12V, VDRVCC = VINTVCC unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Switching Regulator VIN Input Supply Voltage IQ Input Quiescent Current, IVOUT (Note 4) VCAPFBHI Maximum Regulated VCAP Feedback Voltage Full Scale (1111b) l 1.188 1.2 1.212 V VCAPFB_DEF Default VCAPFB_DAC Setting (1010b) l 0.997 1.0125 1.028 V VCAPFBLO Minimum Regulated VCAP Feedback Voltage Zero Scale (0000b) l 0.625 0.6375 0.650 V ICAPFB CAPFB Input Leakage Current VCAPFB = 1.2V l -50 50 nA VOUTFB Regulated VOUT Feedback Voltage l 1.182 1.2 1.218 V VOUTFB(TH) OUTFET Turn-Off Threshold Falling Threshold 1.27 1.3 1.33 V IOUTFB OUTFB Input Leakage Current VOUTBST VOUT Voltage in Step-Up Mode VUVLO l 4.5 35 2.25 V mA VOUTFB = 1.2V l -50 50 nA VIN = 0V l 4.5 35 V INTVCC Undervoltage Lockout Rising Threshold Falling Threshold l l 3.85 4.3 4 4.45 V V VDRVUVLO DRVCC Undervoltage Lockout Rising Threshold Falling Threshold l l 3.75 4.2 3.9 4.35 V V VDUVLO VOUT - VCAP Differential Undervoltage Lockout Rising Threshold Falling Threshold l l 160 55 200 90 240 125 mV mV VOVLO Switcher VIN Overvoltage Lockout Rising Threshold Falling Threshold l l 37.7 36.3 38.6 37.2 39.5 38.1 V V VVCAPP5 Charge Pump Output Voltage Relative to VCAP, 0V < VCAP < 20V 5 V Input Current Sense Amplifier VSNSI Regulated Input Current Sense Voltage (ISNSP_CHG - ISNSM) l 31.04 32 32.96 mV l 31.04 32 32.96 mV Charge Current Sense Amplifier VSNSC Regulated Charge Current Sense Voltage (ICAP-VCAP) VCAP = 10V, Charge Mode VCMC Common Mode Range (ICAP, VCAP) VPEAK Peak Inductor Current Sense Voltage Active in Both Step-Up/Step-Down Modes IICAP ICAP Pin Current Step-Down Mode, VSNSC = 32mV Step-Up Mode, VSNSC = 32mV 0 l 51 58 20 V 65 mV 27 100 A A Error Amplifier gMV VCAP Voltage Loop Transconductance 1 mmho gMC Charge Current Loop Transconductance 64 mho gMI Input Current Loop Transconductance 64 mho gMO VOUT Voltage Loop Transconductance 350 mho Oscillator fSW Switching Frequency RT = 107k l DCMAX Maximum Programmable Frequency RT = 53.6k Minimum Programmable Frequency RT = 267k Maximum Duty Cycle Step-Down Mode, 53.6k<RT<267k Step-Up Mode, 53.6k<RT<267k 495 490 500 500 505 510 1 MHz 200 97 87 98 93 kHz kHz kHz 99.5 % % Rev. A For more information www.analog.com 3 LTC3351 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TJ = 25C (Note 2). VIN = VOUT = 12V, VDRVCC = VINTVCC unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Gate Drivers RUP-TG TGATE Pull-Up On-Resistance RDOWN-TG TGATE Pull-Down On-Resistance RUP-BG BGATE Pull-Up On-Resistance RDOWN-BG BGATE Pull-Down On-Resistance 0.6 tNO Non-Overlap Time 50 ns 85 ns V tON(MIN) 2 0.6 2 INTVCC Linear Regulator VINTVCC Internal VCC Voltage 5.2V < VIN < 35V 5 DVINTVCC Load Regulation IINTVCC = 50mA -1.5 -2.5 % Ideal Diode VFTO Fast On Threshold Voltage 65 mV VFR VRTO Forward Regulation Voltage 30 mV Reverse Turn Off Threshold Voltage -30 mV UV/OV Comparator VUV/OVI(TH) UV/OV Input Threshold (Rising Edge) VUV/OVI(HYS) UV/OV Hysteresis IUV/OV UV/OV Input Leakage Current VUV/OV = 0.5V VVINGD VINGD Output Low Voltage ISINK = 5mA IVINGD VINGD High-Z Leakage Current VINGD = 5V l 1.182 1.2 1.218 60 l -50 mV 50 300 nA mV 1 l UV Falling to VINGD Low Delay V 1 A s CAPGD VCAPFB(TH) CAPGD Rising Threshold as % of Regulated VCAP Feedback Voltage vcapfb_dac = Full Scale (1111b) VCAPFB(HYS) CAPGD Hysteresis at CAPFB as a % of Regulated VCAP Feedback Voltage vcapfb_dac = Full Scale (1111b) VCAPGD CAPGD Output Low Voltage ISINK = 5mA ICAPGD CAPGD High-Z Leakage Current VCAPGD = 5V l 90 92 94 % 2.5 % 200 mV 1 l A Analog-to-Digital Converter VRES Measurement Resolution VGPI General Purpose Input Voltage Range Unbuffered Buffered IGPI General Purpose Input Pin Leakage Current Buffered Input RGPI GPI Pin Resistance Buffer Disabled 4 16 0 0 1.25 Bits 5 3.5 V V 1 A M Rev. A For more information www.analog.com LTC3351 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TJ = 25C (Note 2). VIN = VOUT = 12V, VDRVCC = VINTVCC unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS VIN = 0V VIN = 30V 100 1.5 mV % VOUT = 5V VOUT = 30V 100 1.5 mV % VCAP = 0V VCAP = 10V 100 1.5 mV % VGPI = 0V, Unbuffered VGPI = 5V, Unbuffered 2 1 mV % VCAP1 = 0V VCAP1 = 2V 2 1 mV % VCAP2 = 0V VCAP2 = 2V 2 1 mV % VCAP3 = 0V VCAP3 = 2V 2 1 mV % VCAP4 = 0V VCAP4 = 2V 2 1 mV % VSNSI = 0mV VSNSI = 32mV 200 2 V % VSNSC = 0mV VSNSC = 32mV 200 2 V % Measurement System Error VERR Measurement Error (Note 5) CAP1 to CAP4 RSHNT Shunt Resistance DVCAPMAX Maximum Shunt Operating Voltage 0.5 3.6 V 1.209 V Programming Pins VITST ITST Voltage RTST = 20 l 1.182 1.2 I2C/SMBus - SDA, SCL, SMBALERT IIL,SDA,SCL Input Leakage Low -1 1 A IIH,SDA,SCL Input Leakage High -1 1 A VIH Input High Threshold 1.5 VIL Input Low Threshold 0.8 V fSCL SCL Clock Frequency 400 kHz tLOW Low Period of SCL Clock 1.3 s tHIGH High Period of SCL Clock 0.6 s tBUF Bus Free Time Between Start and Stop Conditions 1.3 s tHD,STA Hold Time, After (Repeated) Start Condition 0.6 s tSU,STA Setup Time After a Repeated Start Condition 0.6 s tSU,STO Stop Condition Set-Up Time 0.6 s tHD,DATO Output Data Hold Time 0 tHD,DATI Input Data Hold Time 0 ns tSU,DAT Data Set-Up Time 100 ns tSP Input Spike Suppression Pulse Width VSMBALERT SMBALERT Output Low Voltage V 900 50 ISINK = 5mA 200 ns ns mV Rev. A For more information www.analog.com 5 LTC3351 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TJ = 25C (Note 2). VIN = VOUT = 12V, VDRVCC = VINTVCC unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN ISMBALERT SMBALERT High-Z Leakage Current VSMBALERT = 5V l TYP MAX UNITS 1 A Hot Swap Controller VIN(UVLO_HS) Hot Swap Input Supply Undervoltage Lockout VIN rising, Hot Swap operation l 3.7 3.85 4 V VILIM(TH) Current Limit Threshold (ISNS_HS-ISNSM) VOUT < VIN - 11V VOUT = VIN l l 7 46 10 48 13 50 mV mV I(OV,UV,RETRYB) OV, UV, RETRYB Pin Input Current l -50 0 50 nA VTH(RETRYB) RETRYB Threshold l 185 200 215 mV VTIMER(TH) CTIMER Pin Threshold l 1.188 1.2 0.3 1.212 mV mV ITIMER(DN) CTIMER Pull-Down Current l 1.4 2 2.6 A ITIMER(UP) CTIMER Pull-Up Current l 320 400 480 A ITIMER(RATIO) CTIMER Pull-Down/Pull-Up Current l 0.5 0.6 % IHS_GATE(UP) HS_GATE Pull-Up Current l 24 30 A IHS_GATE(FST) HS_GATE Fast Pull-Down Resistance Rising Falling VHS_GATE = VOUT Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3351 is tested under pulsed load conditions such that TJ TA. The LTC3351E is guaranteed to meet specifications from 0C to 125C junction temperature. Specifications over the -40C to 125C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3351I is guaranteed over the -40C to 125C operating junction temperature range. Note that the maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. The junction temperature (TJ, in C) is calculated from the ambient temperature (TA, in C) and power dissipation (PD, in Watts) according to the formula: TJ = TA + (PD * JA) where JA = 36.4C/W for the UFF package. 6 16 25 Note 3: The LTC3351 includes over temperature protection that is intended to protect the device during momentary overload conditions. When over temperature protection is active the switcher is shutdown. Junction temperature will exceed 125C when over temperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See the Applications Information section. Note 5: Measurement error is the magnitude of the difference between the actual measured value and the ideal value. VSNSI is the voltage between ISNS_CHG and ISNSM, representing input current. VSNSC is the voltage between ICAP and VCAP, representing charge current. Error for VSNSI and VSNSC is expressed in V, a conversion to an equivalent current may be made by dividing by the sense resistors, RSNSI and RSNSC, respectively. Rev. A For more information www.analog.com LTC3351 TYPICAL PERFORMANCE CHARACTERISTICS Hot Swap Current Limit Fold-Back Curve TA = 25C, unless otherwise noted. Example Hot Swap, 12V 50 Hot Swap and Full Charge Cycle VIN 2V/DIV VIN 2V/DIV ISNS_HS - ISNSM (mV) 42 34 26 VCAP 2V/DIV IIN 0.5A/DIV IIN 0.2A/DIV 18 10 VOUT 2V/DIV VOUT 2V/DIV 500ms/DIV 5ms/DIV 3351 G03 3351 G02 0 2 5 7 VIN - VOUT (V) 10 12 3351 G01 Hot Swap Startup into Shorted Output Hot Swap and Begin Charge VIN 2.0V/DIV Hot Swap Short Detailed VIN 2.0V/DIV VOUT 2.0V/DIV IIN 0.2A/DIV VCSS 0.5V/DIV VCTIMER 0.5V/DIV IIN 0.2A/DIV VCSS 0.5V/DIV IIN 0.5A/DIV HS_GATE 0.5V/DIV VCTIMER 0.5V/DIV 100s/DIV 10ms/DIV 10ms/DIV 3351 G06 3351 G05 3351 G04 Backup Operation, 1A Backup Operation, 2A VIN 2.0V/DIV VOUT 2.0V/DIV VCAP 2.0V/DIV VIN 2.0V/DIV IIN 0.5A/DIV VCAP 2.0V/DIV Backup Operation, 3.5A VIN 2V/DIV VOUT 2V/DIV VCAP 2V/DIV VOUT 2.0V/DIV IIN 0.5A/DIV 1.2s/DIV IIN 0.5A/DIV 500ms/DIV 3351 G07 300ms/DIV 3351 G08 3351 G09 Rev. A For more information www.analog.com 7 LTC3351 TYPICAL PERFORMANCE CHARACTERISTICS IIN and ICHARGE vs VIN 2.9 2.3 5.00 5.00 3.75 3.75 2.50 IIN(MAX) = 2A ICHARGE = 4A IOUT = 0A VIN = 12V VIN = 24V VIN = 35V 1.25 IIN 1.7 11 21 16 26 VIN (V) 31 0 36 ICHARGE vs VCAP ICHARGE (A) ICHARGE (A) IIN(MAX) = 2A, ICHARGE = 4A IOUT = 1A, VCAP = 6V 125C 25C -40C ICHARGE 3.5 CURRENT (A) ICHARGE vs VCAP 0 2 4 6 IIN(MAX) = 2A, ICHARGE = 4A VIN = 12V VIN = 24V VIN = 35V VOUT = 10V 60 50 40 0 0.001 3.00 0.1 IOUT (A) 1 10 100 Charger Efficiency vs VCAP 1.196 60 40 VIN = 12V VIN = 24V VIN = 35V vcapfb_dac = 0xF -10 20 50 80 TEMPERATURE (C) 110 140 3351 G16 8 0 0 1.7 3.4 5.1 VCAP (V) 6.8 6 9 vcapfb_dac (CODE) 12 15 Charge Current vs Shunt Voltage IIN(MAX) = 2A IOUT = 0A 20 3 100 80 EFFICIENCY (%) VCAPFB (V) 1.199 0 3351 G15 3351 G14 VCAP vs Temperature 1.197 0.9 0.7 0.01 3351 G13 1.198 1.0 0.8 VCAP = 2V VCAP = 3V VCAP = 4V IOUT (A) 1.195 -40 VCAP vs vcapfb_dac 30 10 1.200 8 1.1 VCAP (V) EFFICIENCY (%) CURRENT (A) 1.2 20 2.25 6 70 1.25 1.50 4 3351 G12 80 IIN 0.75 2 VCAP (V) Efficiency in Boost Mode 90 2.50 0 0 3351 G11 100 ICHARGE 0 IIN(MAX) = 2A ICHARGE = 4A IOUT = 1A VIN = 12V VIN = 24V VIN = 35V VCAP (V) IIN and ICHARGE vs IOUT 3.75 0 8 3351 G10 5.00 2.50 1.25 8.5 3351 G17 CHARGE CURRENT (% OF MAXIMUM) 4.1 TA = 25C, unless otherwise noted. 90 80 70 60 50 40 30 20 10 0 -100 -75 -50 -25 0 25 MAXIMUM CAPACITOR VOLTAGE - VSHUNT (mV) 3351 G18 Rev. A For more information www.analog.com LTC3351 PIN FUNCTIONS CAP_SLCT0, CAP_SLCT1 (Pins 1, 2): CAP_SLCT0 and CAP_SLCT1 set the number of super-capacitors used. Refer to Table1 in the Applications Information section. VINGD (Pin 3): Power-Fail Status Output. This open-drain output is pulled low when VOUT is not powered from VIN. SCL (Pin 4): Clock Pin for the I2C/SMBus Serial Port. SDA (Pin 5): Bidirectional Data Pin for the I2C/SMBus Serial Port. SMBALERT (Pin 6): Interrupt Output. This open-drain output is pulled low when an alarm threshold is exceeded and will remain low until the acknowledgement of the part's response to an SMBus ARA. CAPGD (Pin 7): Capacitor Power Good. This open-drain output is pulled low when CAPFB is below VCAPFB(TH). VC (PIN 8): Control Voltage Pin. This is the compensation node for the charge current, input current, supercapacitor stack voltage and output voltage control loops. An RC network is needed between VC and SGND. There is an internal compensation resistor in series with this pin. It is 1k in buck mode and 2k in boost mode. Nominal voltage range for this pin is 1V to 3V. CAPFB (Pin 9): Capacitor Stack Feedback Pin. This pin closes the feedback loop for constant voltage regulation. An external resistor divider between VCAP and SGND with the center tap connected to CAPFB programs the final supercapacitor stack voltage. This pin is nominally equal to the output of the VCAP DAC when the synchronous controller is charging in constant voltage mode. OUTFB (Pin 10): Step-Up Mode Feedback Pin. This pin closes the feedback loop for voltage regulation of VOUT during input power failure using the synchronous controller in step-up mode. An external resistor divider between VOUT and SGND with the center tap connected to OUTFB programs the minimum backup supply rail voltage when input power is unavailable. This pin is nominally 1.2V when in backup and the synchronous controller is not in current limit. To disable step-up mode tie OUTFB to INTVCC. SGND (Pin 11): Signal Ground. All small-signal and compensation components should be connected to this pin, which in turn connects to PGND. SGND should connect to PGND on top metal under the LTC3351. PGND should be connected to the ground plane with vias under the exposed pad (pin 45). This should be the only connection between SGND and the ground plane. RT (Pin 12): Timing Resistor. The switching frequency of the synchronous controller is set by placing a resistor, RT, from this pin to SGND. This resistor is always required. If not present the synchronous controller will not start. GPI (Pin 13): General Purpose Input. The voltage on this pin is digitized directly by the ADC. For high impedance inputs an internal buffer can be selected and used to drive the ADC. The GPI pin can be connected to a negative temperature coefficient (NTC) thermistor to monitor the temperature of the supercapacitor stack. A low drift bias resistor is required from INTVCC to GPI and a thermistor is required from GPI to ground. Connect GPI to SGND if not used. Read the digitized voltage on this pin in the meas_gpi register. ITST (Pin 14): Programming Pin for Capacitance Test Current. This current partially discharges the capacitor stack at a precise rate for capacitance measurement. This pin servos to 1.2V during a capacitor measurement. A resistor, RTST, from this pin to SGND programs the test current. The resistor on this pin must be at least 20. Current flows from VCAP4 to this pin during this test and must not dissipate more than 300mW in the IC. CAPRTN (Pin 15): Capacitor Stack Shunt Return Pin. Connect this pin to the grounded bottom plate of the first supercapacitor in the stack through a shunt resistor. CAP1 (Pin 16): First Supercapacitor Pin. The top plate of the first supercapacitor and the bottom plate of the second supercapacitor are connected to this pin through a shunt resistor. CAP1 and CAPRTN are used to measure the voltage across the first supercapacitor and shunt current around the capacitor to provide balancing and prevent overvoltage. The voltage between this pin and CAPRTN is digitized and is read in the meas_vcap1 register. Rev. A For more information www.analog.com 9 LTC3351 PIN FUNCTIONS CAP2 (Pin 17): Second Supercapacitor Pin. The top plate of the second supercapacitor and the bottom plate of the third supercapacitor are connected to this pin through a shunt resistor. CAP2 and CAP1 are used to measure the voltage across the second supercapacitor and to shunt current around the capacitor to provide balancing and prevent overvoltage. If not used, this pin should be shorted to CAP1. The voltage between this pin and CAP1 is digitized and is read in the meas_vcap2 register. CAP3 (Pin 18): Third Supercapacitor Pin. The top plate of the third supercapacitor and the bottom plate of the fourth supercapacitor are connected to this pin through a shunt resistor. CAP3 and CAP2 are used to measure the voltage across the third supercapacitor and shunt current around the capacitor to provide balancing and prevent overvoltage. If not used, this pin should be shorted to CAP2. The voltage between this pin and CAP2 is digitized and is read in the meas_vcap3 register. CAP4 (Pin 19): Fourth Supercapacitor Pin. The top plate of the fourth supercapacitor is connected to this pin through a shunt resistor. CAP4 and CAP3 are used to measure the voltage on the capacitor and shunt current around the supercapacitor to provide balancing and prevent overvoltage. If not used, this pin should be shorted to CAP3. The voltage between this pin and CAP3 is digitized and is read in the meas_vcap4 register. The capacitance test current set by the ITST pin is pulled from this pin. CFP (Pin 20): VCAPP5 Charge Pump Flying Capacitor Positive Terminal. Place a 6.3V 0.1F between CFP and CFN. CFN (Pin 21): VCAPP5 Charge Pump Flying Capacitor Negative Terminal. Place a 6.3V 0.1F between CFP and CFN. VCAPP5 (Pin 22): Charge Pump Output. The internal charge pump drives this pin to VCAP + INTVCC. It is used as the high side rail for the OUTFET gate drive and charge current sense amplifier. Connect a 6.3V 0.1F capacitor from VCAPP5 to VCAP. OUTFET (Pin 23): Output Ideal Diode Gate Drive Output. This pin controls the gate of an external N-channel MOSFET used as an ideal diode between VOUT and VCAP. The gate drive receives power from the internal charge pump output VCAPP5. Connect the source of the 10 N-channel MOSFET to VCAP and the drain to VOUT. If the output ideal diode MOSFET is not used, OUTFET should be left floating. VCAP (Pin 24): Supercapacitor Stack Voltage and Charge Current Sense Amplifier Negative Input. Connect this pin to the top of the supercapacitor stack. The voltage at this pin is digitized and is read in the meas_vcap register. ICAP (Pin 25): Charge Current Sense Amplifier Positive Input. The ICAP and VCAP pins measure the voltage across the sense resistor, RSNSC, to provide instantaneous current signals for the control loops and ESR measurement system. The maximum charge current is 32mV/RSNSC. The voltage between this pin and VCAP is the charging/ discharge current as read in the meas_ichg register. VCC2P5 (Pin 26): Internal 2.5V Regulator Output. This regulator provides power to the internal logic circuitry only. Decouple this pin to SGND with a minimum 1F low ESR ceramic capacitor. SW (Pin 27): Switch Node Connection to the Inductor. The negative terminal of the boot-strap capacitor, CB, is connected to this pin. The voltage on this pin is also used as the source reference for the top side N-channel MOSFET gate drive. In step-down mode, the voltage swing on this pin is from a diode (external) forward voltage below ground to VOUT. In step-up mode, the voltage swing is from ground to a diode forward voltage above VOUT. TGATE (Pin 28): Top Gate Driver Output. This pin is the output of a floating gate driver for the top external N-channel MOSFET. The voltage swing at this pin is ground to VOUT + DRVCC. BST (Pin 29): TGATE Driver Supply Input. The positive terminal of the boot-strap capacitor, CB, is connected to this pin. This pin swings from a diode voltage drop below DRVCC up to VOUT + DRVCC. BGATE (Pin 30): Bottom Gate Driver Output. This pin drives the bottom external N-channel MOSFET between PGND and DRVCC. DRVCC (Pin 31): Power Rail for the Bottom Gate Driver. Connect to INTVCC or to an external supply. Decouple this pin to ground with a minimum 6.3V 2.2F low ESR ceramic capacitor. Do not exceed 5.5V on this pin. Rev. A For more information www.analog.com LTC3351 PIN FUNCTIONS INTVCC (Pin 32): Internal 5V Regulator Output. The control circuits and gate drivers (when connected to DRVCC) are powered from this supply. If not connected to DRVCC, decouple this pin to ground with a minimum 4.7F low ESR ceramic capacitor. CSS (Pin 38): Soft Start and Delay Capacitor Pin. A capacitor from this pin to VOUT determines both the maximum dV/dt of VOUT during the power up and the debounce delay from OV and UV becoming good before attempting to reconnect VIN and VOUT. VOUT (Pin 33): Output Voltage Supply. This pin supplies power to the LTC3351 after the hot swap start-up has finished. The switching controller charges the capacitor stack from the voltage at this pin and the LTC3351 backs up the voltage at this pin if the input voltage goes outside the OV/UV range or an input current fault occurs. SRC (Pin 39): Hot Swap/Input FETs Source Pin. This pin senses the source voltage of the hot swap and input FETs. RETRYB (Pin 34): Retry Comparator Input. The hot swap controller will not attempt to connect VIN and VOUT unless this pin is below 200mV. This high voltage capable pin may be connected to VOUT if the system needs to be fully powered down before repowering after a power loss and backup. ISNSM (Pin 35): Input Current Sense Pin. This is the negative input for both the hot swap current sense amplifier and the switching charger input current sense amplifier. ISNSP_CHG (Pin 36): Input Current Sense Pin. This is the positive input for the switching charger input current sense amplifier. The switching charger will reduce charge current to keep the voltage between this pin and ISNSM to 32mV. The current is measured using the sense resistor between this pin and ISNSM and is measured by the ADC and reported in the meas_iin register. ISNSP_HS (Pin 37): Input Current Sense Pin. This is the positive input for the hot swap input current sense amplifier. The hot swap controller will limit the voltage between this pin and ISNSM to 48mV. This pin is also the input current sense for the circuit breaker function. HS_GATE (Pin 40): Hot Swap/Input FETs Gate Pin. This pin controls the external hot swap/input FETs. This pin is pulled up to, at most, VINTVCC above the VOUT pin. VIN (Pin 41): External DC Power Source Input and Sense Pin. For VIN voltages greater than 8V, a 100 resistor in series with this pin is required. The voltage at this pin is digitized and is reported in the meas_vin register. UV (Pin 42): Power-Fail Comparator Input. When the voltage at this pin drops below VUV(TH), the hot swap controller is disconnected, the part enters backup mode and VINGD is pulled low. OV (Pin 43): Power-Fail Comparator Input. When the voltage at this pin exceeds VOV(TH), the hot swap controller is disconnected, the part enters backup mode and VINGD is pulled low. CTIMER (Pin 44): Fault and Retry Timing Capacitor. A capacitor from this pin to SGND programs the fault and retry timing. During an over current fault condition this capacitor is charged with ITIMER(UP) (400A). Once this pin voltage exceeds VTIMER(TH) (1.2V) a fault is declared and VOUT is disconnected from VIN. This pin is continuously discharged with ITIMER(DN) (2A), and once it discharges to 300mV, and RETRYB is low, the hot swap controller will again attempt to reconnect VIN and VOUT. PGND (Exposed Pad Pin 45): Power Ground. For rated thermal performance connect the exposed pad to a continuous ground plane on the second layer of the printed circuit board by several vias directly under the LTC3351. Connect the exposed pad to the SGND pin on top copper. Rev. A For more information www.analog.com 11 LTC3351 BLOCK DIAGRAM 41 40 HS_GATE VIN 38 43 SRC 35 ISNSP_CHG ISNSM HOT SWAP CONTROLLER OV D/A Vcapfb_dac CAPFB VREF OUTFB 1k (BUCK) 2k (BOOST) + - VCAP + - + - ICAP VPEAK VREF BST TGATE IREF ICHG SW DRVCC OSC INTVCC 5V LDO VOUT BANDGAP VREF BGATE START-UP LDO VCC2P5 CAP4 2.5V LDO 22 VFR BIDIRECTIONAL SWITCHING CONTROLLER RT VIN 26 - VCAPP5 CHARGE PUMP x37.5 + - 32 CFN + - 12 - UV VC INTVCC + 8 21 CFP x37.5 IIN VREF 10 20 OUTFET - + CSS vcapfb_dac[3:0] 9 23 VOUT CTIMER VREF 42 33 36 ISNSP_HS + 44 37 39 24 25 29 28 27 31 30 19 INTVCC 3 CAPGD BALANCER vcapfb_dac VINGD VREF 16 34 1 2 6 5 4 13 11 SHUNT CONTROLLER CAP3 CAPFB MULTIPLEXER 7 + - 16-BIT A/D RETRYB LOGIC IIN ICHG VCAP VOUT VIN CAP4 CAP3 CAP2 CAP1 CAPRTN DTEMP BALANCER BALANCER SHUNT CONTROLLER CAP2 SHUNT CONTROLLER CAP1 18 17 16 CAP_SLCT0 BALANCER CAP_SLCT1 SMBALERT + - SDA SCL GPI SHUNT CONTROLLER - + CAPRTN 15 VREF ITST 14 GPIBUF SGND PGND 45 3351 BD 12 Rev. A For more information www.analog.com LTC3351 TIMING DIAGRAM Definition of Timing for F/S Mode Devices on the I2C Bus SDA tLOW tf tSU(DAT) tr tHD(SDA) tf tBUF tr tSP SCL tHD(SDA) S tSU(STA) tHIGH tHD(DAT) tSU(STO) Sr P S 3351 TD S = START, Sr = REPEATED START, P = STOP I2C/SMBus Legend S Sr Rd Wr A N P PEC* START CONDITION REPEATED START CONDITION READ (BIT VALUE OF 1) WRITE (BIT VALUE OF 0) ACKNOWLEDGE NACK STOP CONDITION PACKET ERROR CODE MASTER TO SLAVE SLAVE TO MASTER SMBus WRITE WORD PROTOCOL S SLAVE ADDRESS Wr A COMMAND CODE A DATA BYTE LOW A DATA BYTE HIGH A DATA BYTE LOW A DATA BYTE HIGH A P SMBus WRITE WORD WITH PEC PROTOCOL S SLAVE ADDRESS Wr A COMMAND CODE A PEC* A P SMBus READ WORD PROTOCOL S SLAVE ADDRESS Wr A COMMAND CODE A Sr SLAVE ADDRESS Rd A DATA BYTE LOW A DATA BYTE HIGH N A Sr SLAVE ADDRESS Rd A DATA BYTE LOW A DATA BYTE HIGH A P SMBus READ WORD WITH PEC PROTOCOL S SLAVE ADDRESS Wr A COMMAND CODE PEC* N P SMBus ALERT RESPONSE ADDRESS PROTOCOL S ALERT RESPONSE ADDRESS Rd A DEVICE ADDRESS Rd N P SMBus ALERT RESPONSE ADDRESS PROTOCOL WITH PEC S ALERT RESPONSE ADDRESS Rd A DEVICE ADDRESS Rd A PEC* N P *USE OF PACKET ERROR CHECKING IS OPTIONAL Rev. A For more information www.analog.com 13 LTC3351 OPERATION Introduction The LTC3351 is a highly integrated backup power controller and system monitor. It features a bidirectional switching controller, hot swap controller, output ideal diode, supercapacitor shunts/balancers, under and over voltage comparators, a 16-bit ADC, and I2C/SMBus programmability with status reporting. If VIN is within externally programmable UV/OV threshold voltages, the hot swap controller connects VIN to VOUT and the synchronous switching controller operates in step-down mode charging a stack of supercapacitors. A programmable input current limit ensures that the supercapacitors will automatically be charged at the highest possible charge current that the input can support. If VIN goes outside the UV/OV thresholds, or if the hot swap controller's circuit breaker trips, or if a simulated failure is requested, then the hot swap controller will disconnect VOUT from VIN and the synchronous controller will run in reverse as a step-up converter to deliver power from the supercapacitor stack to VOUT. An ideal diode controller drives an external MOSFET to provide a low loss power path from VCAP to VOUT. This ideal diode works seamlessly with the bidirectional controller to provide power from the supercapacitors to VOUT. The hot swap controller utilizes two back-to-back MOSFETs to control inrush, provide a short circuit breaker function and prevent back driving VIN while in backup mode. The LTC3351 provides balancing and overvoltage protection to a series stack of one to four supercapacitors. The internal capacitor voltage balancers eliminate the need for external balance resistors. Overvoltage protection is provided by shunt regulators that use an internal switch and an external resistor across each supercapacitor. The LTC3351 monitors system voltages, currents, and its own die temperature. A general purpose input (GPI) pin is provided to measure an additional system parameter or implement a thermistor measurement. In addition, the LTC3351 can measure the capacitance and equivalent series resistance of the supercapacitor stack. This provides indication of the health of the supercapacitors 14 and, along with the VCAP voltage measurement, provides information on the total energy stored and the maximum power that can be delivered. Operations Example The LTC3351 is a highly integrated circuit with many features and operating modes. To better explain the operations of the LTC3351, a simplified example will be used. This example is graphically shown in Figure1 and will be referred to throughout. Due to the widely varying time scales of the events with which the LTC3351 operates, the time axis of Figure1 is not to scale. The example begins with VIN and VCAP at 0V. VIN is applied suddenly at the point labeled "hot plug". There is a very small inrush current into the drain capacitance of the hot swap FET connected to VIN, this is shown as a small "spike" on the IIN waveform. This "spike" is very small in either duration or amplitude, depending on the rise rate of VIN. During the time labeled "debounce", the LTC3351 qualifies the input as good using the UV and OV comparators and expires an input debounce timer using the CSS pin. Once this debounce time has passed, the LTC3351 begins turning on the hot swap FETs to charge the capacitance on VOUT in a controlled way, during which time both the input current and rise rate of VOUT are controlled. This results in a low constant IIN current while the VOUT capacitance is charged. Once VOUT has been charged to VIN, the charger is allowed to start charging the supercapacitors. For this example; at the beginning of the charge cycle the supercapacitors are fully discharged. The charger will begin with constant current charging of the supercapacitors. Since the capacitor voltage is very low, the power delivered is very low. This low power delivery results in a low input current despite high charge current. As the voltage on the supercapacitors rises, the delivered power also rises and thus the input current also rises. This constant current phase of charging is labeled "CC Charging." In this example IOUT, the downstream system load, turns on during the constant current phase of charging. When this happens is outside the control of the LTC3351 and its timing in this example is arbitrary. Since output current Rev. A For more information www.analog.com LTC3351 OPERATION is supplied from VIN via the hot swap FETs this step in load current directly causes a step in input current. The charge current is unaffected by this load step because the LTC3351 is not in input current limit. As the supercapacitors charge, their voltage increases and the input current increases due to the increasing power being delivered to the supercapacitors. In this example the increasing input current reaches the input current limit at the beginning of the time labeled "CP charging." With a constant input voltage, input current limit causes the LTC3351 to effectively have an input power limit. Depending on the settings of the input current limit, charge current limit, charge voltage, system load current and input voltage the charging phase of operation may or may not reach the input current limit. During the constant power phase of charging, the charge current decreases as the charge voltage increases to maintain constant input power. This results in a slowing rate of charge as the charger approaches constant voltage charging. Once VCAP reaches the programmed charge voltage, the constant voltage phase of charging begins. This is labeled "CV charging." During constant voltage charging, current is being delivered to the supercapacitors and there is a voltage drop across their internal ESR (equivalent series resistance). The LTC3351 holds the VCAP voltage constant during this phase. Holding the VCAP voltage constant allows the voltage across this ESR to fall towards zero as the internal capacitance is charged to VCAP. During this time the charge current decays toward the leakage current of the capacitors. The LTC3351 CV charges forever, waiting for a power failure, while keeping the capacitors charged, balanced and ready for backup. While fully charged and standing by for backup, the LTC3351 can also measure the ESR and capacitance of the batteries if requested (not shown in Figure1). This is the condition the LTC3351 is likely to spend most time in, assuming input power loss is infrequent as is typically the case. When power does fail, as labeled "power failure" in Figure1, the OV or UV comparators detect the power failure at the input. The hot swap FET is turned off to isolate VOUT from VIN. Once the FET is off, VOUT immediately falls to VCAP (or the programmed boost voltage if DEBOUNCE CC CHARGING POWER FAILURE INRUSH HOT CP CHARGING IDEAL DIODE BACKUP CV CHARGING AND BALANCING PLUG CONTROL BOOST BACKUP VIN VOLTAGE VOUT VCAP ICAP CURRENT 0 IIN 0 IOUT TIME (NOT TO SCALE) 3351 F01 Figure1. Rev. A For more information www.analog.com 15 LTC3351 OPERATION it is higher, however in this example it is not). The ideal diode FET between VCAP and VOUT is turned on during the time labeled "ideal diode backup." While operating in ideal diode backup, power is supplied directly from the capacitors without conversion. VCAP is discharged during this time and VOUT falls along with VCAP. Bidirectional Switching Controller--Step-Down Mode The bidirectional switching controller is designed to charge a series stack of supercapacitors (Figure 2). Charging proceeds at a constant current until the supercapacitors reach their maximum charge voltage determined by the CAPFB servo voltage and the resistor divider between VCAP and CAPFB. The maximum charge current is determined by the value of the sense resistor, RSNSC, connected in series with the inductor. The charge current loop servos the voltage across the sense resistor to 32mV. When charging begins, an internal soft-start ramp gradually increases the charge current. The VCAP voltage and charge current are read from the meas_vcap and meas_ichg registers, respectively. Once the VOUT voltage approaches the programmed boost voltage, the boost converter is turned on and the ideal diode turned off. The boost converter then supplies the output at constant voltage. As the capacitor voltage falls, the capacitor current must increase to supply the constant power load at VOUT. This continues until either the boost reaches its current limit or exhausts the available energy in the capacitors. This time is labeled "boost backup." VIN VOUT (TO SYSTEM) RSNSI ISNSM ISNSP_CHG ISNSP_HS HS_GATE SRC VIN LTC3351 HOT SWAP CONTROLLER INPUT CURRENT CONTROLLER + - VREF + - 37.5 IIN TGATE BIDIRECTIONAL SWITCHING CONTROLLER CHARGE CURRENT CONTROLLER CAPACITOR VOLTAGE CONTROLLER BGATE STEP-DOWN MODE + - IREF + - ICHG + - ICAP 37.5 VCAP D/A VREF vcapfb_dac[3:0] CAPFB VC RSNSC + + + + 3351 F02 Figure2. Power Path Block Diagram - Power Available from VIN 16 Rev. A For more information www.analog.com LTC3351 OPERATION The LTC3351 provides constant power charging (for a fixed VIN) by limiting the input current drawn by the switching controller in step-down mode. The charger input current limit will reduce charge current to limit the voltage between ISNSP_CHG and ISNSM, typically across RSNSI, to 32mV. If the combined system load plus supercapacitor charge current is large enough to cause the switching controller to reach the programmed input current limit, the input current limit loop will reduce the charge current by precisely the amount necessary to enable the external load to be satisfied. Even if the charge current is programmed to exceed the allowable input current, the input current limit will not be violated; the supercapacitor charger will reduce its current as needed. The input current is read from the meas_iin register. Bidirectional Switching Controller--Step-Up Mode The bidirectional switching controller acts as a step-up converter to provide power from the supercapacitors to VOUT when input power is unavailable (Figure3). VINGD low enables step-up mode. VOUT regulation is set by a resistor divider between VOUT and OUTFB. To disable stepup mode tie OUTFB to INTVCC. Step-up mode is often used with the output ideal diode. If the VOUT regulation voltage is set below the capacitor stack voltage, upon removal of input power, power to VOUT is provided from the supercapacitor stack via the output ideal diode. VCAP and VOUT will decrease as the load current discharges the supercapacitor stack. The output ideal diode will shut off and VOUT will fall a PN diode (~700mV) below VCAP when the voltage on OUTFB falls below 1.3V (VOUTFB(TH)). If OUTFB falls below 1.2V when the output ideal diode shuts off, the synchronous step-up controller will turn on immediately to regulate OUTFB to 1.2V by providing power from the supercapacitor stack. If OUTFB is above 1.2V when the output ideal diode shuts off, the load current will flow through the body diode of the output ideal diode N-channel MOSFET for a period of time until OUTFB falls to 1.2V. VCAP < VOUT VOUT (TO SYSTEM) OUTPUT VOLTAGE CONTROLLER VOUT - + LTC3351 VCAP > VOUT OUTFB VREF +- - + OUTFET VFR BIDIRECTIONAL SWITCHING CONTROLLER STEP-UP MODE TGATE RSNSC BGATE + + ICAP + VCAP + 3351 F03 VC Figure3. Power Path Block Diagram - Power Backup Rev. A For more information www.analog.com 17 LTC3351 OPERATION The synchronous controller in step-up mode will run nonsynchronously when VCAP is less than 90mV (VDUVLO falling) below VOUT. It will run synchronously when VCAP falls 200mV (VDUVLO rising) below VOUT. Hot Swap Controller Upon applying power to VIN, the LTC3351 will immediately turn on a strong pull-down on the HS_GATE pin to prevent the external FETs from conducting current from VIN to VOUT. During the initial VIN rise the LTC3351 may limit the rise rate of the voltage at the VIN pin by drawing current through the 100 resistor in series with the pin. The LTC3351 will then drive the INTVCC pin to 3.6V using current from the VIN pin. Once the INTVCC voltage is greater than 3.3V the hot swap turn-on sequence will begin. If both the under voltage (UV) and (OV) comparators are in range, the CSS pin will begin to source 1A of current, charging the CSS capacitor. Once the CSS pin reaches 1.2V and the RETRYB pin is below 200mV (VTH(RETRYB)) the LTC3351 will begin the process of connecting VIN and VOUT using the external FETs. The LTC3351 will begin pulling up on the HS_GATE pin using 24A (IHS_GATE(UP)) of current. As VOUT begins to rise, the capacitor from VOUT to the CSS pin provides feedback to the LTC3351 about the rise rate of the output. Therefore, the sizing of the CSS capacitor determines the maximum slew rate of VOUT and the inrush current. During the ramp up of HS_GATE and VOUT, both the current limit and the circuit breaker are active. VIN to VOUT differential foldback reduces both the current limit and circuit breaker threshold while charging the output capacitor; this reduces the required SOA of the hot swap FETs. If, at any time, the under voltage (UV) or overvoltage (OV) comparators go out of range, or if the CTIMER pin reaches 1.2V (VTIMER(TH)), the LTC3351 will declare a fault and quickly turn off the external FET by grounding HS_GATE. The FET sources will be kept within a safe voltage of the gate by the SRC pin. Once a fault is declared the LTC3351's backup controller will begin supplying power to VOUT from the energy stored in the capacitor stack. This will continue until either a new turn-on sequence occurs, the supercapacitor stored energy is depleted, or the boost is disabled via I2C/SMBus. 18 The LTC3351 will limit the current from VIN to VOUT through the external FETs using the voltage across the current sense resistor(s) sensed using the ISNSP_HS and ISNSM pins. When VOUT is within 1V of VIN the LTC3351 will limit the voltage across the ISNSP_HS and ISNSM pins to 48mV (VILIM(TH)). To limit power in the external FET this limit is folded back to 10mV as the voltage from VIN to VOUT increases from 1V to 10V. Above 10V the limit remains at 10mV. This current limit is separate from the switching charger's input current limit. The switching charger's input current limit is unaffected by the above described fold back. The LTC3351's CTIMER pin will source current when the input current is within 1.66% of the input current limit. As with the input current limit, this threshold is folded back as the voltage from VIN to VOUT increases. The current sourced from the CTIMER pin to the CTIMER capacitor is about 400A (ITIMER(UP)). Once the voltage at the CTIMER pin exceeds 1.2V (VTIMER(TH) rising) a fault is declared and the turn-off sequence is initiated. The CTIMER pin has a static 2A (ITIMER(DN)) load that discharges the CTIMER capacitor. Once a fault is declared the CTIMER pin must fall below 300mV before the turn-on sequence is re-attempted. RETRYB Pin The LTC3351's RETRYB pin determines if the LTC3351 tries to connect VIN and VOUT after a fault. The faults are UV, OV, circuit breaker (CTIMER), or a simulated fault programmed via the I2C/SMBus port by setting ctl_hotswap_ disable. If the RETRYB pin is low (VTH(RETRYB) below 200mV) the LTC3351 will try to connect VIN and VOUT using the hot swap controller. The RETRYB pin is high voltage tolerant and high impedance. A divider from VOUT to RETRYB allows a precise threshold to be set. This can be used to ensure the system completely powers down following a failure before re-powering. VINGD Pin The VINGD pin indicates that the input voltage is within the OV/UV range and the system is powered from VIN. For VINGD to be high, the voltage at UV must be above 1.2V (VOV rising) the voltage at OV must be below 1.17V (VUV falling), the circuit breaker must not be tripped, the hot Rev. A For more information www.analog.com LTC3351 OPERATION swap must not be disabled via I2C/SMBus, and the hot swap controller must have completed connecting VIN and VOUT. The state of the VINGD pull-down is read from the vingd bit in the sys_status register. If UV is set to a level near or less than the charge voltage of the capacitors, VIN becomes high impedance and the VOUT load is very low, it is possible for a small amount of current to flow from VCAP to VIN through VOUT due to the maximum duty cycle operation. In this condition, the high duty cycle buck is effectively a reverse low duty cycle boost. The boost has a small amount of output current that holds VIN above VCAP and possibly UV, causing the part to falsely indicate VINGD. Eventually VCAP will be discharged below the programmed UV threshold and VINGD will indicate correctly. This situation can be avoided by programming the UV threshold at least 3% above the capacitor charge voltage. Switcher/Charger Undervoltage Lockout (UVLO) Internal undervoltage lockout circuits monitor both the INTVCC and DRVCC pins. The switching controller is kept off until INTVCC rises above VUVLO (4.3V) and DRVCC rises above VDRVUVLO (4.2V). The controller is disabled if either INTVCC falls below 4V or DRVCC falls below 3.9V. Charging is disabled until VOUT is VDUVLO (200mV) above the supercapacitor voltage and VINGD is high. Charging is disabled when VOUT falls to within 90mV of the supercapacitor voltage or when VINGD is low. RT Oscillator and Switching Frequency The RT pin is used to program the switching frequency. A resistor, RT, from this pin to ground sets the switching frequency according to: fSW (MHz ) = 53.5 R T (k ) Ideal Diode The LTC3351 has an ideal diode controller that drives an external N-channel MOSFET between VCAP and VOUT. The ideal diode consists of a precision amplifier that drives the gates of N-channel MOSFETs whenever the voltage at VOUT is approximately 30mV (VFR) below the voltage at VCAP. Within the amplifier's linear range, the small-signal resistance of the ideal diode will be quite low, keeping the forward drop near 30mV. At higher current levels, the MOSFETs will be in full conduction. RT also sets the scale factor for the capacitor measurement value reported in the meas_cap register, described in the ESR and Capacitance Measurement section of this data sheet. The ideal diode provides a path for the supercapacitors to power VOUT when VIN is unavailable or the hot swap controller is disconnected. In addition to a Fast-Off comparator, the ideal diode also has a Fast-On comparator that turns on the external MOSFET when VOUT drops 65mV (VFTO) below VCAP. The ideal diode will shut off when OUTFB is just above regulation allowing the synchronous controller to power VOUT in step-up mode. Switching Controller Input Overvoltage Protection Input overvoltage protection turns off both switching controller switches if VIN exceeds VOVLO (38.6V). The controller will resume switching if VIN falls below 37.2V. The hot swap controller is unaffected by this and uses its own programmable OV threshold. VCAP DAC Gate Drive Supply (DRVCC) The feedback reference for the CAPFB servo point is programmed using an internal 4-bit digital-to-analog converter (DAC). The reference voltage is programmable from 0.6375V (VCAPFBLO) to 1.2V (VCAPFBHI) in 37.5mV increments. The DAC defaults to 0xA (VCAPFB_DEF 1.0125V) and is programmed via the vcapfb_dac register. The bottom gate driver is powered from the DRVCC pin, which is normally connected to the INTVCC pin. An external LDO can also be used to power the gate drivers to minimize power dissipation inside the LTC3351. See the Applications Information section for details. Supercapacitors lose capacitance as they age. By initially setting the VCAP DAC to a low setting, the final charge voltage on the supercapacitors can be increased as they age to maintain a constant level of stored backup energy throughout the lifetime of the supercapacitors. The Rev. A For more information www.analog.com 19 LTC3351 OPERATION capacitance and ESR measurement system may temporarily increase this DAC to a value as much as full scale (1.2V) during the ESR test. If using the capacitance and ESR test, the highest usable DAC setting will be determined by the voltage increase between that setting and 1.2V at the CAPFB pin, wherein the capacitor stack's voltage increases by 1.25 times the voltage specified in cap_delta_v_setting. Charge Status Indication The LTC3351 includes a comparator to report the status of the supercapacitors via an open-drain NMOS transistor on the CAPGD pin. This pin pulls to ground until the CAPFB pin voltage rises to within nominally 8% of the VCAP DAC setting. Once the CAPFB pin is above this threshold, the CAPGD pin goes high impedance. The output of this comparator may also be read from the cappg bit in the sys_status register. Capacitor Voltage Balancer The LTC3351 has an integrated active stack balancer. This balancer slowly balances all of the capacitor voltages to within approximately 10mV of each other. This maximizes the life of the supercapacitors by keeping the voltage on each as low as possible to achieve the needed total stack voltage. When the difference between any two capacitor voltages exceeds approximately 10mV, the capacitor with the largest voltage is discharged with a resistive balancer (approximately 75 until all capacitor voltages are within 10mV). The balancers can be disabled by setting the ctl_disable_balancer bit. Capacitor Shunt Regulators During charging, the capacitors are protected from overvoltage. The capacitors in the stack will not have exactly the same capacitance due to manufacturing tolerances or uneven aging. This will cause the capacitor voltages to increase at different rates with the same charge current. If this mismatch is severe enough or if the capacitors are being charged to near their maximum voltage, it becomes necessary to limit the voltage increase on some capacitors while still charging the other capacitors. Up to 500mA of current may be shunted around a capacitor 20 whose voltage is approaching the programmable shunt voltage. This shunt current reduces the charge rate of that capacitor relative to the other capacitors. If a capacitor continues to approach its shunt voltage, the stack charge current is reduced. This protects the capacitor from overvoltage while still charging the other capacitors, although at a reduced rate of charge. When shunting, the internal switch may be on more than 96% duty cycle. The shunts are disabled by setting the ctl_disable_shunt bit. The shunt voltage is programmable in the vshunt register. Shunt voltages may be programmed in 183.5V increments. If a voltage greater than 3.6V is programmed, the charge current will be reduced as that voltage is approached but the shunt will not turn on. The default value is 0x3999, resulting in a shunt voltage of approximately 2.7V. See Register Map for more information. I2C/SMBus and SMBALERT The LTC3351 contains an I2C/SMBus compatible port. This port allows communication with the LTC3351 for configuration and reading back telemetry data. The port supports two SMBus formats, read word and write word. These may be used with or without the packet error code (PEC) feature. Refer to the SMBus specification for details of these formats and PEC. The registers accessible via this port are organized on an 8-bit address bus and each register is 16 bits wide. The "command code" (or subaddress) of the SMBus read/write word formats is the 8-bit address of each of these registers. The address of the LTC3351 is 0b0001001. The SMBALERT pin is asserted (pulled low) whenever an enabled limit is exceeded or when an enabled status event happens (see the Limit Checking and Alarms and the Monitor Status Register sections of this data sheet). The LTC3351 will de-assert the SMBALERT pin only after responding to a SMBus alert response address (ARA), an SMBus protocol used to respond to a SMBALERT. The host will read from the ARA (0b0001100) and each part asserting SMBALERT will begin to respond with its address. The responding parts arbitrate in such a way that only the part with the lowest address responds completely. Only when a part has responded with its entire address does it release the SMBALERT signal. If multiple parts are asserting the SMBALERT signal then multiple Rev. A For more information www.analog.com LTC3351 OPERATION reads from the ARA are needed. For more information refer to the SMBus specification. Details on the registers accessible through this interface are available in the Register Map section of this data sheet. For I2C masters unable to create the repeated start needed for the read and write word protocols, a stop followed by a start may be substituted. Analog-to-Digital Converter The LTC3351 has an integrated 16-bit sigma-delta analogto-digital converter (ADC). This converter is automatically multiplexed between the measured channels. Its results are stored in registers accessible via the I2C/SMBus port. There are 11 channels measured by the ADC, each of which takes approximately 800s to measure. In addition to providing status information about the system voltages and currents, some of these measurements are used by the LTC3351 to balance, protect (shunt), and measure the capacitors in the stack. The result of each analog-to-digital conversion is stored in a 16-bit register as a signed, two's complement number. To reduce average quiescent current, the effective duty cycle of the ADC can be reduced by programming adc_ wait_vin and/or adc_wait_backup. Each register inserts a delay in the ADC's measurement cycle during its respective mode of operation. Each LSB of these registers has a weight of approximately 400s. At some times, such as when shunting or making capacitance and ESR measurement, these settings may be temporarily ignored. Measurements of individual channels may be enabled or disabled by setting the appropriate bit in adc_backup_ch_ en_reg and adc_vin_ch_en_reg. The measurements from the ADC are stored to meas_ vcap1, meas_vcap2, meas_vcap3, meas_vcap4, meas_ gpi, meas_vin, meas_vcap, meas_vout, meas_iin, meas_ ichg and meas_dtemp. ESR and Capacitance Measurement The LTC3351 monitors the health of a supercapacitor stack by measuring the capacitance and the ESR. Both the capacitance and ESR are measured in a single test. The LTC3351 measures ESR by applying and measuring a current step with the high efficiency charger and measuring the change in voltage. The capacitance is measured by discharging a fixed voltage with a known current and measuring time. The ESR and capacitance measurement sequence, initiated by setting ctl_start_cap_esr_meas, is: 1.The mon_meas_active and mon_esr_meas_active bits become high if the capacitors are charged, otherwise, mon_capesr_pending becomes high. 2.The charger is configured to charge at a pre-set current up to a full vcapfb_dac setting, this current is set using an 8 bit DAC controlled either by: a.An internal algorithm that selects the optimal current based on the previous capacitor measurement (assuming the LTC3351 is not in input current limit) b. An override setting, programmed in esr_i_override, is used if non-zero. If it is likely the LTC3351 will operate in input current limit while charging, then this should be set low enough to avoid input current limit. 3.The measurement system waits esr_i_on_settling for the current and capacitor effects to stabilize. Each LSB of esr_i_on_settling is 1024 switcher periods. 4. A series of measurements of capacitor voltages and charge currents are made. The charger is then temporarily shut off. 5.The measurement system waits esr_i_off_settling for the current and capacitor effects to stabilize. Each LSB of esr_i_off_settling is 1024 switcher periods. 6.A series of measurements of capacitor voltages and charger currents are made. From these measurements, and the previous measurements, the ESR is calculated and stored in meas_esr. 7.The capacitors will be charged to at least 1.25 * cap_ delta_v_setting above their initial voltage (as measured at step 1). If the capacitor voltage reaches the maximum charge voltage (VCAPFB = 1.2V), then the test will stop trying to charge and continue without fully charging. The test may fail later due to this. If the charger is unable to reach 1.25*cap_delta_v_setting above the initial Rev. A For more information www.analog.com 21 LTC3351 OPERATION voltage with capfb less than 1.2V, it will continue trying to charge indefinitely. This may occur if the charger is limited by the input voltage and maximum duty cycle of the buck charger. This may also occur if there is no charge current available due to system load exceeding the input current limit. If either of these conditions occurs, the test will remain in this condition indefinitely. 8.The charger is temporarily disabled. The mon_cap_ meas_active bit becomes high, the mon_esr_meas_ active bit becomes low and the ITST current is enabled. After a time set by cap_i_on_settling, a series of voltage measurements is made. 9.The capacitor stack is discharged a fixed voltage (set by cap_delta_v_setting) from the voltage measured in the previous step using the ITST current (1.2V/RTST, up to 60mA or 300mW). This voltage is measured using the CAP1-4 pins. 10.The time required to discharge by this fixed voltage is measured. It is then scaled for cap_delta_v_setting and stored as meas_cap. 11.The charger stays off and the ITST current stays on until the stack voltage returns to the voltage set by vcapfb_dac. 12.The charger is turned back on and the ITST current is turned off. The mon_meas_active bit goes low. VCAP There is a corresponding monitor status mask register (monitor_status_mask_reg). Writing a one to any of these bits will cause the SMBALERT pin to pull low when the corresponding bit in monitor_status_reg has a rising edge. This allows reduced polling of the LTC3351 when waiting for a capacitance or ESR measurement to complete. cap_m0 (DIFFERENCE) 1.25 * cap_delta_v_setting cap_delta_v_setting vcapfb_dac VCAP SETTING chrg_cv INDICATION esr_i_off_settling cap_i_on_settling ESR TEST CURRENT (AUTOMATIC OR esr_i_override) ICAP The LTC3351 has a monitor status register (monitor_ status_reg) containing status bits to indicate the state of the capacitance and ESR monitoring system. These bits are set and cleared by the capacitor monitor upon certain events during a capacitor and ESR measurement, as described in the ESR and Capacitance Measurement section. esr_m3 IDEAL VOLTAGE esr_i_on_settling Monitor Status Register EXTRAPOLATED MEASUREMENT REAL VOLTAGE esr_m0 The measurement of Capacitance and ESR can fail if power fails during the test or if the capacitor stack is discharged below the CAPGD threshold. The test will also fail if ctl_stop_cap_esr_meas is set. If it does fail, mon_meas_failed will be set. Details of monitor_status_reg and monitor_status_ mask_reg can be found in the Register Map section of this datasheet. Figure4 shows this sequence graphically. esr_m1, esr_m2 This measurement is only initiated when the ctl_start_ cap_esr_meas bit is set. The results of the measurement can be checked against limits and issue a SMBALERT if limits are exceeded, see the Limit Checking and Alarms section of this data sheet. SOFTSTART ESR MEASUREMENT meas_cap CAPACITANCE MEASUREMENT PRECHARGE CURRENT (1/4 FULL SCALE) 0 -ITST SOFTSTART 3351 F04 CAPACITANCE MEASUREMENT Figure4. 22 Rev. A For more information www.analog.com LTC3351 OPERATION System Status Register The sys_status register contains data about the state of the charger, switcher and comparators. Details of this register may be found in the Register Map section of this data sheet. Limit Checking and Alarms The LTC3351 has a limit checking function that will check each measured value against I2C/SMBus programmable limits. This feature is optional and all of the limits are disabled by default. The limit checking is designed to simplify system monitoring, eliminating the need to continuously poll the LTC3351 for measurement data. If a measured parameter goes outside of the programmed level of an enabled limit, the associated bit in the alarm_ reg register is set high and the SMBALERT pin is pulled low. This informs the I2C/SMBus host that a limit has been exceeded. The alarm_reg may then be read to determine exactly which programmed limits have been exceeded. A single ADC is shared between the 11 channels with about 9ms between consecutive measurements of the same channel. In a transient condition, it is possible for these parameters to exceed their programmed levels in between consecutive ADC measurements without setting the alarm. Once the LTC3351 has responded to an SMBus ARA the SMBALERT pin is released. The LTC3351 will not pull the pin low again until another limit is exceeded. To reset a limit that has been exceeded write a zero to the respective bit in the alarm_reg register. When writing alarm_reg, zeros will clear their respective bits in the register, ones will be ignored. A number of the LTC3351's registers are used for limit checking. Individual limits are enabled or disabled in alarm_mask_reg. Once an enabled alarm's measured value exceeds the programmed level for that alarm the alarm is set. That alarm may only be cleared by writing a zero to the appropriate bit of alarm_reg. All alarms that have been set and have not yet been cleared may be read in the alarm_reg. All of the individual measured voltages have a corresponding undervoltage (UV) and overvoltage (OV) alarm level. All of the individual capacitor voltages are compared to the same alarm levels, set in cap_ov_lvl and cap_uv_lvl. The input current measurement has an overcurrent (OC) alarm programmed in iin_oc_lvl. The charge current has an undercurrent alarm programmed in ichg_uc_lvl. Die Temperature Sensor The LTC3351 has an integrated die temperature sensor monitored by the ADC and digitized to meas_dtemp. An alarm is configured on die temperature by setting dtemp_cold_lvl and/or dtemp_hot_lvl and enabling their respective alarms in alarm_mask_reg. To convert the code in the meas_dtemp register to degrees Celsius use the following: TDIE (C) = 0.0295 * meas_dtemp - 274C General Purpose Input The general purpose input (GPI) pin is used to measure an additional system parameter where the voltage on this pin is digitized by the ADC. For high impedance inputs, an internal buffer may be selected and used to drive the ADC. This buffer is enabled by setting the ctl_gpi_buffer_en bit in the ctl_reg register. With this buffer, the input range is limited from 0V to 3.5V. If this buffer is not used, the range is from 0V to 5V, however, the input stage of the ADC will draw about 0.8A per volt from this pin. The ADC input is a switched capacitor amplifier running at about 2MHz, so this current draw will be at that frequency. The pin current can be eliminated at the cost of reduced range and increased offset by enabling the buffer. Alarms are available for this pin voltage with levels programmed using gpi_uv_lvl and gpi_ov_lvl. These alarms are enabled using the mask_alarm_gpi_uv and mask_ alarm_gpi_ov bits in alarm_mask_reg. To monitor the temperature of the supercapacitor stack, the GPI pin can be connected to a negative temperature coefficient (NTC) thermistor. A low drift bias resistor is required from INTVCC to GPI and a thermistor is required from GPI to ground. Connect GPI to SGND if not used. Rev. A For more information www.analog.com 23 LTC3351 OPERATION Internal Diodes The LTC3351 has numerous internal diodes as part of its circuits and ESD protection structures. In normal operation, these diodes are reverse biased. Figure5 shows all HS_GATE SRC the diodes except the substrate diodes. These substrate diodes have their anode connect to ground and their cathodes connect to every pin except SW. ISNSP_CHG ISNSP_HS ISNSM VOUT DRVCC VIN OV BGATE UV BST OUTFB TGATE RETRYB SW INTVCC CAPSLCT0 ICAP CAP_SLCT1 VINGD VCAP SCL CFN SDA LTC3351 INTERNAL DIODES SMBALERT CAPGD VC CFP VCAPP5 CAPFB RT OUTFET GPI CAPRTN VCC2P5 CSS CAP1 CTIMER CAP2 CAP3 ITST CAP4 SUBSTRATE DIODES NOT SHOWN 3351 F05 Figure5. 24 Rev. A For more information www.analog.com LTC3351 APPLICATIONS INFORMATION Digital Configuration Although the LTC3351 has extensive digital features, none are mandatory for basic use. The shunt voltage is programmed via vshunt, which has a default value of 2.7V. The capacitor voltage feedback reference defaults to 1.0125V (VCAPFB_DEF) and is set using vcapfb_dac. If these values are acceptable, no software is required for basic use. All other digital features are optional, most for system monitoring. The ADC automatically runs and stores conversions to registers (e.g., meas_vcap). Capacitance and ESR measurements only run if requested. Each measured parameter has programmable limits (e.g., vcap_uv_lvl and vcap_ov_lvl) which may trigger an alarm and SMBALERT when enabled. All alarms are disabled by default. Capacitor Configuration The LTC3351 is used with one to four supercapacitors. If fewer than four capacitors are used, the capacitors must be populated from CAPRTN to CAP4, and the unused CAP pins must be tied to the highest used CAP pin. For example, if three capacitors are used, tie CAP4 to CAP3. If only two capacitors are used, tie both CAP4 and CAP3 to CAP2. The number of capacitors used must be programmed on the CAP_SLCT0 and CAP_SLCT1 pins by tying the pins to VCC2P5 for a one and ground for a zero as shown in Table1. The value programmed on these pins is read back from num_caps via I2C/SMBus. Table1. Number of Capacitors num_caps register CAP_SLCT1 CAP_SLCT0 1 0 0 0 2 1 0 1 3 2 1 0 4 3 1 1 Capacitor Shunt Regulator Programming VSHUNT is programmed via the vshunt register and defaults to 2.7V at initial power-up. VSHUNT serves to limit the voltage on any individual capacitor by turning on a shunt around that capacitor as the voltage approaches VSHUNT. CAPRTN, CAP1, CAP2, CAP3 and CAP4 are connected to the supercapacitors through resistors which serve as ballasts for the internal shunts. The shunt current is approximately VSHUNT divided by twice the shunt resistance value. For a VSHUNT of 2.7V, 2.7 resistors should be used for 500mA of shunt current. If the shunts are disabled, the shunt resistors must be populated with 100. Since the shunt current is less than what the switcher can supply, the on-chip logic will automatically reduce the charging current to allow the shunt to protect the capacitor. This greatly reduces the charge rate once any shunt is activated. For this reason, program VSHUNT as high as possible to reduce the likelihood of it activating during a charge cycle. Ideally, VSHUNT is set high enough so that any likely capacitor mismatches would not cause the shunts to turn on. This keeps the charger operating at the highest possible charge current and reduces the charge time. If the shunts never turn on, the charge cycle completes quickly and the balancers eventually equalize the voltage on the capacitors. The shunt setting may also be used to discharge the capacitors for testing, storage or other purposes. Simulated Power Failure The LTC3351 has the ability to simulate a power failure by setting ctl_hotswap_disable. This causes the hot swap controller to disconnect VOUT from VIN and indicate power has failed exactly as if it would had power actually failed. In this configuration all power consumption downstream will be supplied by the supercapacitors either through the ideal diode or the boost converter. If, during this test, the stored energy is exhausted, then VOUT will collapse, just as in a real power failure. At the end of the simulated failure test, the ctl_hotswap_ disable bit must be cleared to allow the hot swap to reconnect VIN to VOUT. The min_vout_hs_disable register may be used to automatically clear ctl_hotswap_disable if VOUT falls below the programmed voltage. Clearing the ctl_hotswap_disable does not force the hot swap to reconnect, it only allows it to reconnect if its usual conditions are met, mainly that OV, UV and RETRYB voltages are correct. If the hot swap is re-enabled while there is system load current, it is considered a hot reconnect and is discussed in the Hot Reconnects section of this datasheet. Rev. A For more information www.analog.com 25 LTC3351 APPLICATIONS INFORMATION Hot Swap Component Selection The hot swap controller will servo the HS_GATE pin to regulate the voltage across the sense resistor(s) between ISNSP_HS and ISNSM to be, at most, 48mV (VILIM(TH)) . This current limit is folded back as the voltage between VIN and VOUT increases to 12V, at which point the regulation voltage drops to 12mV and no further. The CSS capacitor is used both to set an input qualification delay (debounce) and to limit the VOUT dV/dt rate to limit the inrush current. dVOUT/dt = 48A/CSS t DELAY = 1.2V * C SS Setting Switcher Input and Charge Currents 1A The primary concern when selecting a CSS capacitor value is to select a value large enough to slow the VOUT rise rate such that the input current stays below the minimum hot swap current limit due to foldback. The following equations are for input voltages above 10V and assume a 12mV minimum current limit voltage. The minimum CSS capacitor could be reduced further for lower voltage inputs due to the minimum current limit voltage being higher due to less foldback. The following equations assume any VOUT load remain off until after the hot swap completes, if loads are present on VOUT the CSS capacitor must be further increased to set a VOUT rise rate such that the dV/dt*COUT current and the load current do not exceed the folded back current limit at any point. The maximum dV/dt of the output without reaching current limit is dVOUT dt = 12mV Minimum C SS = 12mV IIN(MAX) = 32mV R SNSI ICHG(MAX) = 32mV R SNSC The peak inductor current limit for both buck and boost modes, IPEAK, is 80% higher than the maximum charge current and is equal to: IPEAK = 58mV R SNSC This current limit is active in both charging and backup modes. In backup mode, it is the only control limitation on inductor and output current. * R SNS * C OUT = 4mmho * R SNS * C OUT Low Current Charging and High Current Backup The CSS capacitance may be increased to any value to achieve a longer delay, however it must be larger than the minimum CSS computed above to avoid current limit and tripping the circuit breaker. 26 The maximum switcher input current is determined by the resistance across the ISNSP_CHG and ISNSM pins, typically RSNSI. The maximum charge current is determined by the value of the sense resistor, RSNSC, connected in series with the inductor. The input and charge current loops servo the voltage across their respective sense resistor to 32mV. Therefore, the maximum input and charge currents are: R SNS * C OUT 48A The switcher and hot swap controller both share the negative terminal for their current sense amplifiers. The switcher reduces charger current so that there is at most 32mV between ISNSP_CHG and ISNSM and the hot swap controller will limit the input current to at most 48mV between ISNSP_HS and ISNSM. This allows a single sense resistor to be used in many applications, resulting in a hot swap circuit breaker that is 50% higher than the switcher's input current limit. Any two values may be selected by using two current sense resistors, see the Input Sense Resistors Selection section of this data sheet for more information. The LTC3351 accommodates applications requiring low charge currents and high backup currents. In these applications, program the desired charge current using RSNSI. The higher current needed during backup is set using Rev. A For more information www.analog.com LTC3351 APPLICATIONS INFORMATION RSNSC. The input current limit will override the charge current limit when the supercapacitors are charging while the charge current limit provides sufficient current capability for backup operation. Note that the backup current will flow through RSNSI2. Size the RSNSI2 resistor package to handle the powerdissipation. The charge current will be limited to ICHG(MAX) at low VCAP (i.e., low duty cycles). As VCAP rises, the switching controller's input current will increase until it reaches IIN(MAX). The input current will be maintained at IIN(MAX) and the charge current will decrease as VCAP rises further. VOUT (TO SYSTEM) ILOAD Some applications may want to use only a portion of the input current limit to charge the supercapacitors. Two input current sense resistors placed in series can be used to accomplish this as shown in Figure6. ISNSP_CHG is kelvin connected to the positive terminal of RSNSI1 and ISNSM is kelvin connected to the negative terminal of RSNSI2. The load current is pulled through RSNSI1 while the input current to the charger is pulled through RSNSI1 and RSNSI2. The input current limit is: 32mV = RSNSI1 * ILOAD + (RSNSI1 + RSNSI2) * IINCHG For example, suppose that only 2A of input current is desired to charge the supercapacitors but the system load and charger combined can pull a total of up to 4A from the supply. Setting RSNSI1 = RSNSI2 = 8m will set a 4A current limit for the load and charger, while setting a 2A limit for the charger. With no system load, the charger can pull up to 2A of input current. As the load pulls 0A to 4A of current, the charger's input will drop from 2A to 0A. The following equation can be used to determine charging input current as a function of system load current: IINCHG = 32mV R SNSI1 +R SNSI2 - R SNSI1 R SNSI1 +R SNSI2 VIN RSNSI1 RSNSI2 VIN HS_GATE ISNSP_CHG IINCHG ISNSM TGATE LTC3351 BGATE 3351 F06 Figure6. Input Sense Resistors Selection Any combination of hot swap current limit and switching charger input current limit can be achieved with two resistors. In Figure7 below, three resistors are shown, however, in all configurations at least one will be replaced with a short. R1 IHS R2 VOUT (TO SYSTEM) R3 IINCHG (TO CHARGER) ISNSP_HS ISNSP_CHG ISNSM LTC3351 3351 F07 *I LOAD The contact resistance of the negative terminal of RSNSI1 and the positive terminal of RSNSI2 as well as the resistance of the trace connecting them will contribute error to the input current limit. To minimize the error, place both input current sense resistors close together with a large PCB pad area between them as the system load current is pulled from the trace connecting the two sense resistors. Figure7. If the desired hot swap current limit is 1.5 time the charger input current limit, then only R2 is needed and R1 and R3 are replaced with shorts. R2 = 48mV/IHS If the desired hot swap current limit is greater than 1.5 times the charger input current limit, then R1 is replaced with a short. This will typically be the case when a higher backup current than charge current is needed. R2 = 48mV/IHS Rev. A For more information www.analog.com 27 LTC3351 APPLICATIONS INFORMATION R3 = 32mV/IINCHG - R2 VCAP In this configuration, R3 adds to the output impedance of the boost. Alternatively, the resistors may be reconfigured as shown in Figure8. RFBC1 LTC3351 CAPFB RFBC2 3351 F9 R3 R2 ISNSP_CHG ISNSP_HS VOUT IINCHG Figure9. Setting VOUT Voltage in Backup Mode ISNSM LTC3351 3351 F08 Figure8. If the desired charger input current limit is more than 2/3 of the hot swap input current limit R3 is replaced with a short. R2 = 32mV/IINCHG The output voltage for the switching controller in stepup mode is set by an external feedback resistor divider, as shown in Figure10. The regulated output voltage is determined by: R VOUT = 1+ FBO1 1.2V R FBO2 Take great care to route the OUTFB line away from noise sources, such as the SW line, BST or TGATE. R1 = 48mV/IHS - R2 VOUT Note that the circuit breaker timer (the CTIMER pin) may run as low as 2% below the current limit, setting the charger's input current limit too close to the hot swap current limit will trip the circuit breaker. Operation with the charger's input current set close to the hot swap current limit requires careful attention to the LTC3351's tolerance for both VILIM(HS) and VSNSI, the tolerance of both current sense resistors, the layout, the worst case switching charger's input current ripple, and how quickly the switching charger can reduce its current due to the fastest increase in downstream VOUT current. RC_INT in Figure10 is 1k in buck mode and 2k in boost mode. Setting VCAP Voltage Compensation The LTC3351 VCAP voltage is set by an external feedback resistor divider, as shown in Figure9. The regulated output voltage is determined by: The input current, charge current, VCAP voltage, and VOUT voltage loops all require a 1nF to 10nF capacitor from the VC node to ground. When using the output ideal diode and backing up to low voltages (<8V), use 8.2nF to 10nF on VC. When not using the output ideal diode, 4.7nF to 10nF on VC is recommended. For very high backup voltages (>15V), 1nF to 4.7nF is recommended. R VCAP = 1+ FBC1 CAPFBREF R FBC2 where CAPFBREF is the output of the VCAP DAC, programmed via vcapfb_dac. Take great care to route the CAPFB line away from noise sources, such as the SW line, BST, TGATE or BGATE. 28 LTC3351 OUTFB VREF - + RC_INT VC RFO (OPT) RFBO1 CFO (OPT) RFBO2 CFBO1 RC (OPT) CC 3351 F10 Figure10. VOUT Voltage Divider and Compensation Network In addition to the VC node capacitor, the VOUT voltage loop requires a phase-lead capacitor, CFBO1, for stability and improved transient response during input power Rev. A For more information www.analog.com LTC3351 APPLICATIONS INFORMATION failure (Figure10). The product of the top divider resistor and the phase-lead capacitor is used to create a zero at approximately 2kHz: R FBO1 * CFBO1 1 2 ( 2kHz ) Choose RFBO1, such that CFBO1 100pF, to minimize the effects of parasitic pin capacitance. Because the phaselead capacitor introduces a larger ripple at the input of the VOUT transconductance amplifier, an additional RC lowpass filter from the VOUT divider to the OUTFB pin may be needed to eliminate voltage ripple spikes. The filter time constant should be located at the switching frequency of the switching controller: R FO * CFO = 1 2fSW with CFO > 10pF to minimize the effects of parasitic pin capacitance. For backup applications, where the VOUT regulation voltage is low (~5V to 6V), an additional 1k to 3k resistor, RC, in series with the VC capacitor improves stability and transient response. Minimum VCAP Voltage in Backup Mode In backup mode, power is provided to the output from the supercapacitors either through the output ideal diode or the switching controller operating in step-up mode. The output ideal diode provides a low loss power path from the supercapacitors to VOUT. The minimum internal (open-circuit) supercapacitor voltage will be equal to the minimum VOUT necessary for the system to operate plus the voltage drops due to the output ideal diode and equivalent series resistance, RSC, of each supercapacitor in the stack. Example: System needs 5V to run and draws 1A during backup. There are four supercapacitors in the stack, each with an RSC of 45m. The output ideal diode forward regulation voltage is 30mV (OUTFET RDS(ON) < 30m). The minimum open-circuit supercapacitor voltage is: VCAP(MIN) = 5V + 0.030V + (1A * 4 * 45m) = 5.21V Using the switching controller in step-up mode allows the supercapacitors to be discharged to a voltage much lower than the minimum VOUT needed to run the system. The amount of power that the supercapacitor stack can deliver at its minimum internal (open-circuit) voltage should be greater than what is needed to power the output and the step-up converter. According to the maximum power transfer rule: PCAP(MIN) = VCAP(MIN)2 P > BACKUP 4 * n * R SC In the equation above is the efficiency of the switching controller in step-up mode and n is the number of supercapacitors in the stack. Example: System needs 5V to run and draws 1A during backup. There are four supercapacitors in the stack (n= 4), each with an RSC of 45m. The converter efficiency is 90%. The minimum open-circuit supercapacitor voltage is: VCAP(MIN) = 4 * 4 * 45m * 5V * 1A 0.9 = 2.0V In this case, the voltage seen at the terminals of the capacitor stack is half this voltage, or 1V, according to the maximum power transfer rule. Note the minimum VCAP voltage can also be limited by the peak inductor current limit (180% of maximum charge current) and the maximum duty cycle in step-up mode(~90%). Optimizing Supercapacitor Energy Storage Capacity In most systems the supercapacitors will provide backup power to one or more DC/DC converters. A DC/DC converter presents a constant power load to the supercapacitor stack. When the supercapacitors are near their maximum voltage, the loads will draw little current. As the capacitors discharge, the current drawn from supercapacitors will increase to maintain constant power to the load. The amount of energy required in back up mode is the product of this constant backup power, PBACKUP, and the backup time, tBACKUP. Rev. A For more information www.analog.com 29 LTC3351 APPLICATIONS INFORMATION The energy stored in a stack of n supercapacitors available for backup is: 1 2 ( 2 2 - V CELL(MIN) nC SC V CELL(MAX) MAX = 1+ 1- ) where CSC, VCELL(MAX) and VCELL(MIN) are the capacitance, maximum voltage and minimum voltage of a single capacitor in the stack, respectively. The maximum voltage on the stack is VCAP(MAX) = n * VCELL(MAX). The minimum voltage on the stack is VCAP(MIN) = n * VCELL(MIN). Some of this energy will be dissipated as conduction loss in the ESR of the supercapacitor stack. A higher backup power requirement leads to a higher conduction loss for a given stack ESR. The amount of capacitance needed is found by solving the following equation for CSC: 1 PBACKUP * tBACKUP = nCSC 4 30 where: 2 MAX * VCELL(MAX) - Min = 1+ 1- 4R SC * PBACKUP 2 n V CELL(MAX) and, 4R SC * PBACKUP 2 n V CELL(MIN) RSC is the equivalent series resistance (ESR) of a single supercapacitor in the stack. Note that the maximum power transfer rule limits the minimum cell voltage to: VCELL(MIN) = VCAP(MIN) n 4R SC * PBACKUP n A calculator for this is available on the LTC3350 website. 2 MIN * VCELL(MIN) - 4RSC *PBACKUP ln n MAX * VCELL(MAX) MIN * VCELL(MIN) Rev. A For more information www.analog.com LTC3351 APPLICATIONS INFORMATION To minimize the size of the capacitance for a given amount of backup energy, increase the maximum voltage on the stack, VCELL(MAX). However, the voltage is limited to a maximum of 2.7V and higher than this may lead to an unacceptably low capacitor lifetime. Capacitor Selection Procedure An alternative option is to keep VCELL(MAX) at a voltage that leads to reasonably long lifetime and increase the capacitor utilization ratio of the supercapacitor stack. The capacitor utilization ratio, B, can be defined as: 3.Choose number of capacitors in the stack. 2 B = 2 V CELL(MAX) - V CELL(MIN) 5.Solve for capacitance, CSC: PBACKUP n B+ tBACKUP B 2 - 1- 2 1 2 nCSC * VCELL(MAX) * 2 B ln 1+ 1- B+ 2 B 1- - 2 B ln (1+ B 1- ) -1 B 6.Find supercapacitor with sufficient capacitance CSC and minimum RSC: 4R SC * PBACKUP VCELL(MIN) = 2PBACKUP * tBACKUP * 2 n VCELL(MAX) CSC where is the efficiency of the boost converter (~90% to 96%). For the backup equation, MAX and MIN, substitute PBACKUP/ for PBACKUP. In this case the energy needed for backup is governed by the following equation: R SC 2 (1- B ) n V CELL(MAX) 4PBACKUP 7.If a suitable capacitor is not available, iterate by choosing more capacitance, a higher cell voltage, more capacitors in the stack and/or a lower utilization ratio. 8.Make sure to take into account the lifetime degradation of ESR and capacitance, as well as the maximum discharge current rating of the supercapacitor. A list of supercapacitor suppliers is provided in Table2. B B Table2. Supercapacitor Suppliers Once a capacitance is found using the above equation the maximum ESR allowed needs to be checked: 4.Choose a desired utilization ratio, B, for the supercapacitor (e.g., 80%). 2 If the synchronous controller is used in step-up mode, then the supercapacitors can be run down to a voltage set by the maximum power transfer rule to maximize the utilization ratio. The minimum voltage in this case is: 2.Determine maximum cell voltage that provides acceptable capacitor lifetime. V CELL(MAX) 1.Determine backup requirements PBACKUP and tBACKUP. R SC 2 (1- B ) n V CELL(MAX) 4PBACKUP AVX www.avx.com Bussmann www.cooperbussmann.com CAP-XX www.cap-xx.com Illinois Capacitor www.illcap.com Kemet Corp. www.kemet.com Maxwell www.maxwell.com Murata www.murata.com NESS CAP www.nesscap.com Tecate Group www.tecategroup.com Rev. A For more information www.analog.com 31 LTC3351 APPLICATIONS INFORMATION Inductor Selection The switching frequency and inductor selection are interrelated. Higher switching frequencies allow the use of smaller inductor and capacitor values, but generally results in lower efficiency due to MOSFET switching and gate charge losses. In addition, the effect of inductor value on ripple current must also be considered. The inductor ripple current decreases with higher inductance or higher frequency and increases with higher VIN. Accepting larger values of ripple current allows the use of low inductances but results in higher output voltage ripple and greater core losses. ripple current and consequent output voltage ripple. Do not allow the core to saturate. The saturation current for the inductor should be at least 80% higher than the maximum regulated current, ICHG(MAX). A list of inductor suppliers is provided in Table3. Table3. Inductor Vendors For the LTC3351, the best overall performance will be attained if the inductor is chosen to be: L= VENDOR URL Coilcraft www.coilcraft.com Murata www.murata.com Sumida www.sumida.com TDK www.tdk.com Toko www.toko.com Vishay www.vishay.com Wurth Electronic www.we-online.com VIN(MAX) ICHG(MAX) * fSW COUT and CCAP Capacitance for VIN(MAX) 2VCAP and: VCAP VCAP L = 1- VIN(MAX) 0.25 *ICHG(MAX) * fSW for VIN(MAX) 2VCAP, where VCAP is the final supercapacitor stack voltage, VIN(MAX) is the maximum input voltage, ICHG(MAX) is the maximum regulated charge current, and fSW is the switching frequency. Using these equations, the inductor ripple will be at most 25% of ICHG(MAX). Using the above equation, the inductor may be too large to provide a fast enough transient response to hold up VOUT when input power goes away. This occurs in cases where the maximum VIN is high (e.g. 25V) and the backup voltage low (e.g. 6V). In these situations it would be best to choose an inductor that is smaller resulting in maximum peak-to-peak ripple as high as 40% of ICHG(MAX). Once the value for L is known, the type of inductor core is selected. Ferrite cores are recommended for their very low core loss. Selection criteria should concentrate on minimizing copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This causes an abrupt increase in inductor 32 VOUT serves as the input to the synchronous controller in step-down mode and as the output in step-up (backup) mode. If step-up mode is used, place 100F of bulk (aluminum electrolytic, OS-CON, POSCAP) capacitance for every 2A of backup current desired. For 5V system applications, 100F per 1A of backup current is recommended. In addition, a certain amount of high frequency bypass capacitance is needed to minimize voltage ripple. The voltage ripple in step-up mode is: VOUT = 1- VCAP V 1 + OUT *RESR IOUT(BACKUP) VOUT COUT * fSW VCAP Maximum ripple occurs at the lowest VCAP that can supply IOUT(BACKUP). Multilayer ceramics are recommended for high frequency filtering. If step-up mode is unused, then the specification for COUT will be determined by the desired ripple voltage in stepdown mode: VOUT = ICHG(MAX) VCAP V 1- CAP +ICHG(MAX) *RESR VOUT VOUT COUT * fSW Rev. A For more information www.analog.com LTC3351 APPLICATIONS INFORMATION In continuous conduction mode, the source current of the top MOSFET is a square wave of duty cycle VCAP/VOUT. To prevent large voltage transients, a low ESR capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: V IRMS ICHG(MAX) CAP VOUT VOUT VCAP on VCAP can be a combination of bulk and high frequency capacitors. Aluminum electrolytic, OS-CON and POSCAP capacitors are suitable for bulk capacitance while multilayer ceramics are recommended for high frequency filtering. Power MOSFET Selection -1 This formula has a maximum at VOUT = 2VCAP, where IRMS = ICHG(MAX)/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Medium voltage (20V to 35V) ceramic, tantalum, OS-CON and switcher-rated electrolytic capacitors can be used as input capacitors. Sanyo OS-CON SVP, SVPD series, Sanyo POSCAP TQC series, or aluminum electrolytic capacitors from Panasonic WA series or Cornell Dubilier SPV series in parallel with a couple of high performance ceramic capacitors can be used as an effective means of achieving low ESR and high bulk capacitance. VCAP serves as the input to the switching controller in step-up mode and as the output in step-down mode. The purpose of the VCAP capacitor is to filter the inductor current ripple. The VCAP ripple (VCAP) is approximated by: Two external power MOSFETs are selected for the LTC3351's synchronous controller: one N-channel MOSFET for the top switch and one N-channel MOSFET for the bottom switch. The selection criteria of the external N-channel power MOSFETs include maximum drainsource voltage (VDSS), threshold voltage, on-resistance (RDS(ON)), reverse transfer capacitance (CRSS), total gate charge (QG), and maximum continuous drain current. Select VDSS of both MOSFETs to be higher than the maximum input supply voltage (including transient). The peakto-peak drive levels are set by the DRVCC voltage. Logiclevel threshold MOSFETs should be used because DRVCC is powered from either INTVCC (5V) or an external LDO whose output voltage must be less than 5.5V. MOSFET power losses are determined by RDS(ON), CRSS and QG. The conduction loss at maximum charge current for the top and bottom MOSFET switches are: V PCOND(TOP) = CAP ICHG(MAX)2 * R DS(ON) (1+ T ) VOUT 1 VCAP I PP +R ESR 8C CAP * fSW where fSW is the switching frequency, CCAP is the capacitance on VCAP and IPP is the ripple current in the inductor. The output ripple is highest at maximum input voltage since IPP increases with input voltage. Because supercapacitors have low series resistance, it is important that CCAP be sized properly so that the bulk of the inductor current ripple flows through the filter capacitor and not the supercapacitor. It is recommended that: n *R 1 SC +R ESR 8C * f 5 CAP SW where n is the number of supercapacitors in the stack and RSC is the ESR of each supercapacitor. The capacitance PCOND(BOT) = 1- VCAP ICHG(MAX)2 * R DS(ON) (1+ T ) VOUT The term (1+ T) is generally given for a MOSFET in the form of a normalized RDS(ON) vs Temperature curve, but = 0.005/C can be used as an approximation for low voltage MOSFETs. Both MOSFET switches have conduction loss. However, transition loss occurs only in the top MOSFET in stepdown mode and only in the bottom MOSFET in step-up mode. These losses are proportional to VOUT2 and can be considerably large in high voltage applications (VOUT > 20V). The maximum transition loss is: PTRAN k 2 VOUT 2 *ICHG(MAX) * CRSS * fSW Rev. A For more information www.analog.com 33 LTC3351 APPLICATIONS INFORMATION where k is related to the drive current during the Miller plateau and is approximately equal to one. The synchronous controller can operate in both stepdown and step-up mode with different voltages on VOUT in each mode. If VOUT is 12V in step-down mode (input power available) and 10V in step-up mode (backup mode) then both MOSFETs can be sized to minimize conduction loss. If VOUT can be as high as 25V while charging and VOUT is held to 6V in backup mode, then the MOSFETs should be sized to minimize losses during backup mode. This may lead to choosing a high side MOSFET with significant transition loss which may be tolerable when input power is available so long as thermal issues do not become a limiting factor. The bottom MOSFET can be chosen to minimize conduction loss. If step-up mode is unused, then choosing a high side MOSFET that that has a higher RDS(ON) device and lower CRSS would minimize overall losses. Another power loss related to switching MOSFET selection is the power lost to driving the gates. The total gate charge, QG, must be charged and discharged each switching cycle. The power is lost to the internal LDO and gate drivers within the LTC3351. The power lost due to charging the gates is: PG (QGTOP + QGBOT) * fSW * VOUT where QGTOP is the top MOSFET gate charge and QGBOT is the bottom MOSFET gate charge. Whenever possible, utilize MOSFET switches that minimize the total gate charge to limit the internal power dissipation of the LTC3351. Schottky Diode Selection Optional Schottky diodes can be placed in parallel with the top and bottom MOSFET switches. These diodes clamp SW during the non-overlap times between conduction of the top and bottom MOSFET switches. This prevents the body diodes of the MOSFET switches from turning on, storing charge during the non-overlap time and requiring a reverse recovery period that could cost as much as 3% in efficiency at high VIN. One or both diodes can be omitted if the efficiency loss can be tolerated. Rate the diode for about one-third to one-fifth of the full load current since it is on for only a fraction of the duty cycle. Larger 34 diodes result in additional switching losses due to their larger junction capacitance. In order for the diodes to be effective, the inductance between them and the top and bottom MOSFETs must be as small as possible. Place these components next to each other on the same layer of the PC board. Top MOSFET Driver Supply (CB, DB) An external bootstrap capacitor, CB, connected to the BST pin supplies the gate drive voltage for the top MOSFET. Capacitor CB, in Figure11, is charged though an external diode, DB, from DRVCC when the SW pin is low. The value of the bootstrap capacitor, CB, needs to be 20 times that of the total input capacitance of the top MOSFET. The bottom MOSFET, MN2 in Figure 11, turning on ensures that the SW pin goes low. If the bottom MOSFET is on for less than 50S for eight consecutive switching cycles, the bottom MOSFET will turn on for 100nS to 250nS at the end of the eighth switching cycle to refresh the voltage on CB. With the top MOSFET on, the BST voltage is above the system supply rail: VBST = VOUT + VDRVCC The reverse break down of the external diode, DB, must be greater than VOUT(MAX) + VDRVCC(MAX). The step-up converter briefly runs non-synchronously when used with the output ideal diode. During this time the BST to SW voltage can pump up to voltages exceeding 5.5V if DB is a Schottky diode. Fast switching PN diodes are recommended due to their low leakage and junction capacitance. A Schottky diode can be used if the step-up converter runs synchronous throughout backup mode. BST CB LTC3351 DB SW DRVCC INTV CC 0.1F 1F OPT >2.2F 3351 F11 Figure11. Rev. A For more information www.analog.com LTC3351 APPLICATIONS INFORMATION INTVCC/DRVCC and IC Power Dissipation The LTC3351 features a low dropout linear regulator (LDO) that supplies power to INTVCC from the VOUT supply. INTVCC powers the gate drivers (when connected to DRVCC) and much of the LTC3351's internal circuitry. The LDO regulates the voltage at the INTVCC pin to 5V. The LDO can supply a maximum current of 50mA and must be bypassed to ground with a minimum of 1F when not connected to DRVCC. DRVCC should have at least a 2.2F ceramic or low ESR electrolytic capacitor. No matter what type of bulk capacitor is used on DRVCC, an additional 0.1F ceramic capacitor placed directly adjacent to the DRVCC pin is highly recommended. Good bypassing is needed to supply the high transient currents required by the MOSFET gate drivers. High input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3351 to be exceeded. The INTVCC current, which is dominated by the gate charge current, is supplied by the 5V LDO. Power dissipation for the IC in this case is highest and is approximately equal to (VOUT) * (IQ + IG), where IQ is the non-switching quiescent current of ~4mA and IG is gate charge current. The junction temperature is estimated by using the equations given in Note 2 of the Electrical Characteristics. For example, the IG supplied by the INTVCC LDO is limited to less than 42mA from a 35V supply in the QFN package at a 70C ambient temperature: TJ = 70C + (35V)(4mA + 42mA)(36.4C/W) = 125C To prevent the maximum junction temperature from being exceeded, the INTVCC LDO current must be checked while operating in continuous conduction mode at maximum VOUT. The power dissipation in the IC is drastically reduced if DRVCC is powered from an external LDO. In this case the power dissipation in the IC is equal to power dissipation due to IQ and the power dissipated in the gate drivers, (VDRVCC) * (IG). Assuming the external DRVCC LDO output is 5V and is supplying 42mA to the gate drivers, at 70C ambient the junction temperature rises to only 80.5C: TJ = 70C + [(35V)(2.25mA)+(5V)(42mA)](36.4C/W) = 80.5C Power the external LDO from VOUT. It must be enabled after the INTVCC LDO has powered up and its output must be less than 5.5V. INTVCC should no longer be tied to DRVCC. Minimum On-Time Considerations Minimum on-time, tON(MIN), is the smallest time duration that the LTC3351 is capable of turning on the top MOSFET in step-down mode. It is determined by internal timing delays and the gate charge required to turn on the top MOSFET. The minimum on-time for the LTC3351 is approximately 85ns. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that: t ON(MIN) < VCAP VOUT * fSW If the duty cycle falls below what can be accommodated by the minimum on-time, the switching controller will begin to skip cycles. The charge current and VCAP voltage will continue to be regulated, but the ripple voltage and current will increase. Ideal Diode MOSFET Selection An external N-channel MOSFET is required for the output ideal diode. Important parameters for the selection of this MOSFET is the maximum drain-source voltage, VDSS, gate threshold voltage and on-resistance (RDS(ON)). When the supercapacitors are at 0V, the input voltage is applied across the output ideal diode MOSFET. Therefore, the VDSS of the output ideal diode MOSFET must withstand the highest voltage on VIN. The gate drive for the ideal diode is 5V. Use logic-level threshold N-channel MOSFET. As a general rule, select a MOSFET with a low enough RDS(ON) to obtain the desired VDS while operating at full load current. The LTC3351 will regulate the forward voltage drop across the output ideal diode MOSFET to 30mV Rev. A For more information www.analog.com 35 LTC3351 APPLICATIONS INFORMATION if RDS(ON) is low enough. The required RDS(ON) can be calculated by dividing 0.030V by the load current in amps. Simplified SOA Requirement Calculation: Achieving forward regulation will minimize power loss and heat dissipation, but it is not a necessity. If a forward voltage drop of more than 30mV is acceptable, then a smaller MOSFET can be used but must be sized compatible with the higher power dissipation. Care should be taken to ensure that the power dissipated is never allowed to rise above the manufacturer's recommended maximum level. 2.Using this slew rate and the output capacitance determine the inrush current. During backup mode, the output ideal diode shuts off when the voltage on OUTFB falls below 1.3V. For high VOUT backup voltages (>8.4V), the output ideal diode will shut off when VCAP is more than a diode drop (~700mV) above the VOUT regulation point (i.e., OUTFB > 1.2V). The body diode of the output ideal diode N-channel MOSFET will carry the load current until VCAP drops to within a diode drop of the VOUT regulation voltage at which point the switching controller takes over. During this period the power dissipation in the output ideal diode MOSFET increases significantly. Diode conduction time is small compared to the overall backup time but can be significant when discharging very large supercapacitors (>600F). Care should be taken to properly heat sink the MOSFET to limit the temperature rise. Hot Swap Input FET Selection In addition to RDS(ON) requirements, the hot swap input FET must be sized for safe operating area (SOA). This is done by sizing for both the start-up characteristic of the system and to handle a short circuit of the duration set by CTIMER. Typically, the FET should be sized for the worst case startup condition and CTIMER should be set such that a short circuit will require less SOA than the startup condition. Logic level gate FETs are required. The foldback curve reduces the SOA requirement of the FET. The LTC3351's foldback curve reduces the effective current limit as the voltage across the FET (as measured from VIN to VOUT) increases. The current is linearly reduced from 100% at 1V to about 20% at 11V, voltages larger than 11V will remain at 20%. 36 1.Determine the output slew rate (see page 26). 3.Add any enabled downstream loads to this inrush current to determine the input current. 4.Using the slew rate and maximum input voltage, determine the output capacitor charge time. Using the output capacitor charging time, the input current and the maximum input voltage, select a FET with an appropriate SOA. The CTIMER capacitor value should be set to not exceed the FETs SOA when the output is shorted with maximum input voltage. Hot Reconnects Return of input power when operating in backup mode must be considered. The RETRYB pin provides a mechanism to prevent returning to input power based on either the VOUT voltage or digital signal, typically indicating the backup has completed. The RETRYB pin may also be grounded causing the LTC3351 to attempt to reconnect the system to the input once all other reconnect criteria are met. If the RETRYB pin is grounded, and a handover from backup operation back to VIN operation is required, it is essential that the load line of the system remain below the foldback curve of the hot swap controller. If the system load exceeds the foldback curve of the hot swap controller, the controller will be unable to support the load and an over-current fault will occur. After the CTIMER cool down time has passed, the cycle will repeat indefinitely until the energy storage is depleted and the load shuts down, returning the system to conditions resembling an initial power-up. This may be prevented by keeping the foldback curve above the systems load line. Boosting to a voltage near VIN will allow the hot swap to start at a high current point on the foldback curve and may allow the hot swap controller to reconnect VIN without running into current limit. Rev. A For more information www.analog.com LTC3351 APPLICATIONS INFORMATION Alternate Hot Swap Controller Configuration Typically, the CSS capacitor sets both the debounce time and the maximum rise rate of VOUT. If the fixed relationship between the debounce time and the maximum rise rate is not satisfactory the hot swap controller has an alternate configuration to decouple these two parameters, however this comes at the expense of reduced current limit speed. In this alternate configuration, the CSS capacitor is connected from the CSS pin to ground. In this configuration, the CSS capacitor will only function as a debounce timer. The rise rate of the output is controlled by IHS_GATE(UP) into the capacitance of the HS_GATE node. Adding a capacitor from the HS_GATE pin to ground allows the output rise rate to be programmed independently of all other parameters. However, this capacitance represents a large load that the current limit amplifier must drive, resulting in a slower current limit. A 1k resistor should be placed in series with this capacitor. This resistor will limit current into the HS_GATE when the HS_GATE pin is pulled to ground due to a fault being detected. Increased Capacitance Test Current The LTC3351 can sink up to 60mA of capacitance test current dissipating up to 300mW. This constant current sink is capable of testing large capacitors, however the test time may become unreasonably long. To increase the test current, a simple circuit using a low Vt (<1.2VVGS) NMOS and a resistor may be used. The gate of the NMOS is connected to the ITST pin such that when the ITST circuit is turned on the NMOS is also turned on. This circuit allows much higher capacitance test current than the LTC3351 alone. LTC3351 a modified equation for converting meas_cap to capacitance. This equation is: C= -56 *10 -9 *R t *meas _ cap R *ln 1- VCAP R 1.2V * + VCAP R TST Where R is the resistance in the added ITST circuit, VCAP is the charge voltage at the beginning of the test and VCAP is the voltage set using the cap_delta_v_setting register. R, RTST and Rt are in Ohms, C is in Farads, and VCAP and VCAP are in volts. The above equation is valid when ctl_cap_scale is 1 (the small setting); if the large setting is used, the above equation should be multiplied by 100. Increased Shunt Current The LTC3351 can shunt up to 500mA around an individual cell that has reached VSHUNT as set by the vshunt register. This limits the charge rate of the other capacitors because the charge current is reduced to near the shunt current while shunting to prevent each cap from exceeding VSHUNT. To enable faster charging while shunting, higher shunt current is needed. LTC3351 CAP4 VCAP R4 C4 MN4 CAP3 10 R3 C3 MN3 CAP2 VCAP 10 10 VCAP R2 R C2 MN2 ITST CAP1 RTST 10 R1 3351 F12 Figure12. Increased Capacitance Test Current When using this circuit, the test current will be the original ITST circuit current plus an additional current due to the capacitor stack voltage across the resistor. This requires C1 MN1 CAPRTN 10 3351 F13 Figure13. Increased Shunt Current Circuit Addition For more information www.analog.com Rev. A 37 LTC3351 APPLICATIONS INFORMATION Higher shunt current may be accommodated with an external NMOS and resistor across each capacitor as shown in Figure13. The shunt current will be VSHUNT divided by Rx (assuming the RDS(ON) of MNx is small compared to Rx). The logic level NMOS must be selected so that its VGS at the desired shunt current is less than one half the VSHUNT voltage. When using this circuit the shunt resistors should be increased to 10 to minimize the drop across the LTC3351's internal shunt FET and thus maximize the VGS for MNx. Using 10 resistors will also reduce the power dissipation allowing a smaller package size compared to the typical application's 2.7. When using 10 resistors the internal shunt FET will turn on the external NMOS, however the internal balancer's current will not create enough voltage across the 10 resistors to turn on the external NMOS. Additionally, increasing the 10 resistors to 500 will allow the internal balancers to turn on the external FETs, thus also increasing balance current. Input Short Protection The LTC3351's input current sense amplifier, ISNSP_HS to ISNSM and ISNSP_CHG to can have at most 0.3V between them. This is not typically an issue since the LTC3351 regulates input current to 48mV (hot swap current limit) or 32mV (charger input current limit) using these pins. The LTC3351 hot swap does not regulate reverse input current. If the input voltage is rapidly reduced (the input directly shorted to ground for instance) large currents will flow from the VOUT capacitance to the VIN short. Unless external protection is used, these large currents can quickly produce many volts of drop across the input sense resistor(s), overstressing the input current sense pins. If the input is shorted after it has fallen below the UV threshold, this is not an issue since the hot swap FETs have already been turned off. However, if the input supply can be shorted to ground while the LTC3351 is connected, the input protection circuit in Figure14 is recommended. Also if the system load is so low that the LTC3351 circuit can be disconnected from the input and then shorted to ground before the input falls below UV, then the same circuit is also recommended. 38 VIN VOUT 1 1 ISNSP_X ISNSP_X 1 ISNSM LTC3351 3351 F14 Figure14. Input Short Protection Circuit Since there are several possible configurations for the input current sense resistors, not all possible protection circuits are shown. To construct a circuit for another configuration of input current sense resistors, there should be one Schottky diode for each current sense amplifier (ISNSP_HS-ISNSM and ISNSP_CHG-ISNSM). There should be 1 between the current sense resistor and the current sense pins, and the diode should be connected such that it can conduct current in the direction from VOUT to VIN, this means the anodes will be connected to ISNSM. In this circuit, the Schottky diode is a simple clamp to limit the voltage across the pins. The 1 resistors are needed to limit the current in each Schottky diode. Clearly, if only one sense resistor is used, only one protection diode is needed. Supercapacitor Settling The ESR-only supercapacitor model using a capacitor in series with an equivalent series resistance (ESR) is over-simplified. Real supercapacitors have an additional settling time due to their internal physics To explain the implications of this settling, assume a super capacitor of infinite capacitance such that any voltage change due to charging or discharging can be ignored. In the simple ESR-only model, a pulse of current would simply result in a step in voltage at its rising edge and a step back to its initial voltage at its falling edge. A real supercapacitor's response to the same pulse of current's rising edge will be an instantaneous step in voltage due to the high frequency ESR, followed by a settling to a higher voltage due to the "DC" ESR. On the falling edge there will be an Rev. A For more information www.analog.com LTC3351 APPLICATIONS INFORMATION instantaneous drop in voltage due to the high frequency ESR and a slower settling back to the original voltage due to the "DC" ESR. The following figure illustrates this. ESR ONLY MODEL REAL VCAP t ICAP 3351 F15 t Figure15. This results in an ESR measurement dependent on the frequency at which it is measured. There is a high frequency ESR and a larger "DC" ESR. In energy storage applications where the capacitors will supply a sustained load, the "DC" ESR limits the deliverable power. Supercapacitor manufacturers have various methods for measuring ESR. Even when specifying a "DC" ESR, the measurement is often done at a frequency significantly higher than "DC", often near 100Hz, thereby reporting an intermediate frequency ESR. ESR Measurement Timing Adjustments The LTC3351 default ESR measurement timing is set to measure between the DC and high frequency ESR, closer to the high frequency ESR for a wide variety of supercapacitors. The timing can be adjusted to measure closer to the DC ESR, however this must done empirically based on the exact switching controller configuration and capacitors used. Values appropriate for some capacitors would result in failed measurements for others and there are no meaningful defaults preprogrammed in the LTC3351. Using an DC coupled oscilloscope with a precision offset, observe the supercapacitor stack voltage (VCAP) during the ESR test. AC coupling will not work as the signal time scales are too long to cross the oscilloscope's AC coupling high pass filter without distortion. Set the oscilloscope to trigger on SMBALERT falling and enabling the mon_ meas_active alert using the mask_mon_meas_active bit. Then set the ctl_start_cap_esr_meas to start the ESR measurement. The SMBAlert will need to be cleared each time to re-trigger the oscilloscope. Alternatively, a current probe can be used to trigger on input current, since the input current will be high during the ESR test. The esr_i_override setting is available to prevent the automatic adjustment of the ESR test current. The automatically selected ESR test current is available in the next_ esr_i register. After several measurements have been completed, next_esr_i may be used as a starting point to manually set esr_i_override. While adjusting the timing of the ESR measurement, either set esr_i_override so the test current won't change between successive measurements or do each measurement twice for each setting and ignore the first result. Referring to Figure4, using the oscilloscope you can now see the effects of esr_i_on_settling and esr_i_off_settling on the captured waveform. Begin by adjusting esr_i_off_ settling so that VCAP is no longer decreasing significantly after esr_i_off_settling time. This time will likely be significantly longer than the default time. The esr_i_on_settling time is a little more complicated to adjust as the capacitors are being charged during this time. If the esr_i_on_settling time is too short, the internal chemistry of the supercapacitor will not have settled and the ESR measured will be closer to the high frequency ESR than the DC ESR. If esr_i_on_settling is too long and esr_i_override is not set, then charge current will be automatically adjusted downward, resulting in reduced signal for the ESR measurement. If esr_i_on_settling is too long while esr_i_override is set, the capacitors will charge to constant voltage and charge current will fall, corrupting the measurement. This can be observed in the input current waveform. During an ESR test, it should not decrease, it should turn off cleanly. The charge current can also be observed by comparing esr_m1_i and esr_m2_i. They should be about the same, esr_m2_i should not be significantly less than esr_m1_i. If it is, esr_i_on_settling is too long. If esr_i_override is set, monitor the increase in stack voltage during the ESR test. Ideally it should be about the same as the v in the capacitance test (as set by cap_ delta_v_setting). If the change in voltage during the ESR test is less than in the capacitance test, esr_i_override can be increased to maximize the signal available for Rev. A For more information www.analog.com 39 LTC3351 APPLICATIONS INFORMATION ESR test. If the voltage increase during the ESR test is more than the voltage set by cap_delta_v_setting, cap_ delta_v_setting can be increased for more resolution in the capacitance test. When configuring the test, there must be enough voltage between the vcapfb_dac setting and 1.2V to perform the test. If configuring the ESR test to measure the high frequency ESR, the only adjustment necessary is to reduce the esr_i_off_settling to 0. The default setting for esr_i_on_ settling is set to fully allow the switcher to settle under all normal conditions and should not be further reduced when measuring the high frequency ESR. Individual Capacitor ESR and Capacitance Calculation The LTC3351 reports total stack capacitance and ESR. From the stored voltage and current measurements, it is possible to calculate each individual capacitor's ESR and capacitance. For the calculation algorithm contact the factory. Capacitance Measurement Timing Adjustments Like the ESR measurement, the capacitance measurement is subject to non-ideal effects of the supercapacitors. The LTC3351 measures capacitance by pre-charging the capacitor stack by 1.25*cap_delta_v_setting (the ESR test contributes to this charging), turning on a test discharge current, waiting the time specified by cap_i_ on_settling, measuring a first voltage, then waiting for the stack to discharge cap_delta_v_setting from the first measurement. In this test the capacitors are discharged by a known voltage with a known current and the time is proportional to capacitance. Since the capacitors are charged by 1.25 times the change in voltage during the capacitance test, there is an "extra" 25 percent of the measurement time. Some of this time can be used to allow the capacitors to settle into a constant rate of voltage decrease. A starting point would be to use 15 percent of the expected discharge time as cap_i_on_settling. In the equation below, C should be the minimum expected stack capacitance due to aging and tolerance and v is change in voltage. 40 t DISCHARGE = v * C * R TST 1.2V Combining the above equation, the equation for switcher frequency and cap_i_on_settling LSB weight results in: cap _ i _ on _ settling = v * C * 6.5 * 10 6 * R TST Rt The above equation is only a starting point. Adequate settling can be confirmed by triggering an oscilloscope on the rising edge of ITST (or alternately on SMBALERT using the mon_esr_done alarm). Observe the response of the discharging capacitors. They should have clearly settled into a linear discharge by the time specified in cap_i_on_settling after the trigger. If they have not settled in time, cap_i_on_settling needs to be increased. cap_i_ on_settling cannot be increased beyond 25 percent of the expected minimum measurement time. If 25 percent of the expected measurement time is inadequate to settle, the expected measurement time may be increased by increasing v via cap_delta_v_setting. PCB Layout Considerations When laying out the printed circuit board, the following guidelines should be used to ensure proper operation of the IC. Check the following in your layout: 1.The VCC2P5 bypass capacitor must return to SGND or the ground plane. If returning to the ground plane keep away from the switcher's high di/dt loop. 2.Referring to Figure16, keep MN1, MN2 and COUT close together. The high di/dt loop formed by the MOSFETs, Schottky diodes and the VOUT capacitance should have short, wide traces to minimize high frequency noise and voltage stress from inductive ringing. Surface mount components are preferred to reduce parasitic inductances from component leads. Connect the drain of the top MOSFET directly to the positive terminal of COUT. Connect the source of the bottom MOSFET directly to the negative terminal of COUT. This capacitor provides the AC current to the MOSFETs. Rev. A For more information www.analog.com LTC3351 VOUT COUT L1 MN1 RSNSC VCAP + HIGH FREQUENCY CIRCULATING PATH + MN2 CCAP + + 3351 F16 Figure16. resistor connections should not be run along the high current feeds from the COUT capacitor. 6.Route ICAP and VCAP sense lines together, keep them short. Apply this rule to ISNSP_HS, ISNSP_CHG and ISNSM as well. Filter components should be placed near the part and not near the sense resistors. Ensure accurate current sensing with Kelvin connections at the sense resistors. See Figure17. 3.Ground is referenced to the negative terminal of the VCAP decoupling capacitor in step-down mode and to the negative terminal of the VOUT decoupling capacitor in step-up mode. The combined IC SGND pin/PGND paddle and the ground returns of CINTVCC and CDRVCC must return to the combined negative terminal of COUT and CCAP. 4.Effective grounding techniques are critical for successful DC/DC converter layouts. Orient power components such that switching current paths in the ground plane do not cross through the SGND pin and exposed pad on the backside of the LTC3351. Switching path currents can be controlled by orienting the MOSFET switches, the inductor, and VOUT and VCAP decoupling capacitors in close proximity to each other. It is important to keep SGND and the components connected to SGND away from interference due to switching currents. To do this, an island of SGND is formed on top metal. This SGND island should only connect to PGND on top metal under the LTC3351, between the SGND pin and the exposed pad of the LTC3351. This should be the only connection between SGND and PGND. This area of top metal is where the LTC3351's small signal components such as RT, VC, feedback dividers and VCC2P5 bypass capacitor should be connected. Power components and all other bypass capacitors should not connect to SGND. 5.Locate VCAP and VOUT dividers near the LTC3351 and away from switching components. Kelvin the top of resistor dividers to the positive terminals of CCAP and COUT, respectively. The bottom of the resistive dividers should return directly to the SGND pin. The feedback DIRECTION OF SENSED CURRENT RSNSC OR RSNSI 3351 F17 TO VCAP TO ICAP OR OR ISNSM ISNSP_HS OR ISNSP_CHG Figure17. 7.Locate the DRVCC and BST decoupling capacitors in close proximity to the LTC3351. These capacitors carry the MOSFET drivers' high peak currents. An additional 0.1F ceramic capacitor placed immediately next to the DRVCC pin can help improve noise performance substantially. 8.Locate the small-signal components away from high frequency switching nodes (BST, SW, TG, and BG). All of these nodes have very large and fast moving signals and should be kept on the output side of the LTC3351. 9.The output ideal diode senses the voltage between VOUT and VCAP. VCAP is used for Kelvin sensing the charge current. Place the output ideal diode MOSFET near the charge current sense resistor, RSNSC, with a short, wide trace to minimize resistance between the source of the ideal diode MOSFET and RSNSC. 10. The OUTFET pin for the external ideal diode controller has extremely limited drive current. Care must be taken to minimize leakage to adjacent PC board traces. 100nA of leakage from this pin will introduce an additional ideal diode offset of approximately 10mV. Rev. A For more information www.analog.com 41 LTC3351 REGISTER MAP Symbol Name ctl_reg Command Code Access Bit Range 0 R/W [10:0] 0 Control Register: Several independent control bits are grouped into this register. [0] 0 Begin a capacitance and ESR measurement when possible; this bit clears itself once a measurement cycle begins or becomes pending. ctl_start_cap_esr_meas Default Description Enum: start_measurement = 1 ctl_gpi_buffer_en [1] 0 A one in this bit location enables the input buffer on the GPI pin. With a zero in this location the GPI pin is measured without the buffer. ctl_stop_cap_esr_meas [2] 0 Stops an active capacitance/ESR measurement; this bit clears itself once a measurement cycle has been stopped. ctl_cap_scale [3] 0 Enum: stop_measurement = 1 Increases capacitor measurement resolution 100 times, this is used when measuring smaller capacitors. Enums: large_cap = 0, small_cap = 1 ctl_disable_shunt [4] 0 Disables the shunt feature. ctl_hotswap_disable [5] 0 Disables the HotSwap controller. The gate of the hotswap FET is forced low, disconnecting VIN and VOUT and forcing the switcher into backup mode. This can be used to simulate a power failure for testing. ctl_force_boost_off [6] 0 This bit disables the boost. ctl_force_charger_off [7] 0 This bit disables the charger. ctl_force_itst_on [8] 0 This bit forces the ITST current on. This can be used to discharge the capacitor stack or manually measure capacitance. Note that this only enables the test current, it does not disable the charger. Set ctl_force_charger_off to disable the charger. ctl_disable_balancer [10] 0 Disables the balancer. [15:0] 0 Mask Alarms Register: Writing a one to any bit in this register enables a rising edge of its respective bit in alarm_reg to trigger an SMBALERT. mask_alarm_gpi_uv [0] 0 GPI Under Voltage alarm mask mask_alarm_gpi_ov [1] 0 GPI Over Voltage alarm mask mask_alarm_vin_uv [2] 0 VIN Under Voltage alarm mask mask_alarm_vin_ov [3] 0 VIN Over Voltage alarm mask mask_alarm_vcap_uv [4] 0 VCAP Under Voltage alarm mask mask_alarm_vcap_ov [5] 0 VCAP Over Voltage alarm mask mask_alarm_vout_uv [6] 0 VOUT Under Voltage alarm mask mask_alarm_vout_ov [7] 0 VOUT Over Voltage alarm mask mask_alarm_dtemp_ cold [8] 0 Die temperature cold alarm mask mask_alarm_dtemp_ hot [9] 0 Die temperature hot alarm mask mask_alarm_ichg_uc [10] 0 Charge undercurrent alarm mask mask_alarm_iin_oc [11] 0 Input overcurrent alarm mask mask_alarm_cap_uv [12] 0 Capacitor Under Voltage alarm mask mask_alarm_cap_ov [13] 0 Capacitor Over Voltage alarm mask mask_alarm_cap_lo [14] 0 Capacitance low alarm mask mask_alarm_esr_hi [15] 0 ESR high alarm mask alarm_mask_reg 42 1 R/W Rev. A For more information www.analog.com LTC3351 REGISTER MAP Command Code Access Bit Range 2 R/W [14:0] 0 Mask Monitor Status Register: Writing a one to any bit in this register enables a rising edge of its respective bit in monitor_status_reg to trigger an SMBALERT. mask_mon_meas_ active [0] 0 Set the SMBALERT when there is a rising edge on mon_meas_active mask_mon_capesr_ pending [2] 0 Set the SMBALERT when there is a rising edge on mon_capesr_pending mask_mon_cap_done [3] 0 Set the SMBALERT when there is a rising edge on mon_cap_done mask_mon_esr_done [4] 0 Set the SMBALERT when there is a rising edge on mon_esr_done mask_mon_meas_ failed [5] 0 Set the SMBALERT when there is a rising edge on mon_meas_failed mask_mon_disable_ charger [7] 0 Set the SMBALERT when there is a rising edge on mon_disable_charger mask_mon_cap_ meas_active [8] 0 Set the SMBALERT when there is a rising edge on mon_cap_meas_active mask_mon_esr_meas_ active [9] 0 Set the SMBALERT when there is a rising edge on mon_esr_meas_active mask_mon_power_ failed [10] 0 Set the SMBALERT when there is a rising edge on mon_power_failed mask_mon_power_ returned [11] 0 Set the SMBALERT when there is a rising edge on mon_power_returned mask_mon_balancing [12] 0 Set the SMBALERT when there is a rising edge on mon_balancing mask_mon_shunting [13] 0 Set the SMBALERT when there is a rising edge on mon_shunting mask_mon_cap_ precharge [14] 0 Set the SMBALERT when there is a rising edge on mon_cap_precharge Symbol Name monitor_status_ mask_reg Default Description vcapfb_dac 3 R/W [3:0] 10 VCAP Regulation Reference: This register is used to program the capacitor voltage feedback loop's reference voltage. Only bits 3:0 are active. VCAPFB_DAC = 37.5mV * vcapfb_dac + 637.5mV vshunt 5 R/W [15:0] 14744 Shunt Voltage Register: This register programs the shunt voltage for each capacitor in the stack. When set below 3.6V, the charger will limit current and the active shunts will shunt current to prevent this voltage from being exceeded. As a capacitor voltage nears this level, the charge current will be reduced. Current will be shunted when the capacitor voltage is within 25mV of vshunt. Vshunt should be programmed at least 50mV higher than the intended final balanced individual capacitor voltage. When programmed above 3.6V no current will be shunted, however the charge current will be reduced as described. 182.8V per LSB. adc_vin_ch_en_reg 6 R/W [11:1] 3842 adc_vin_ichg_en [1] 1 Enables ADC measurement of charge current while in charging mode. adc_vin_dtemp_en [2] 0 Enables ADC measurement of die temperature while in charging mode. adc_vin_gpi_en [3] 0 Enables ADC measurement of GPI (general purpose input) while in charging mode. adc_vin_iin_en [4] 0 Enables ADC measurement of input current while in charging mode. adc_vin_vout_en [5] 0 Enables ADC measurement of vout while in charging mode. adc_vin_vcap_en [6] 0 Enables ADC measurement of vcap while in charging mode. adc_vin_vin_en [7] 0 Enables ADC measurement of vin while in charging mode. adc_vin_vcap1_en [8] 1 Enables ADC measurement of vcap1 while in charging mode. This bit must be set for capacitance and ESR measurement. Rev. A For more information www.analog.com 43 LTC3351 REGISTER MAP Symbol Name Command Code Access Bit Range Default Description adc_vin_vcap2_en [9] 1 Enables ADC measurement of vcap2 while in charging mode. This bit must be set for capacitance and ESR measurement if there are two or more capacitors in the stack. adc_vin_vcap3_en [10] 1 Enables ADC measurement of vcap3 while in charging mode. This bit must be set for capacitance and ESR measurement if there are three or more capacitors in the stack adc_vin_vcap4_en [11] 1 Enables ADC measurement of vcap4 while in charging mode. This bit must be set for capacitance and ESR measurement if there are four capacitors in the stack [11:1] 0 adc_backup_ichg_en [1] 0 Enables ADC measurement of charge current while in backup mode. adc_backup_dtemp_en [2] 0 Enables ADC measurement of die temperature while in backup mode. adc_backup_gpi_en [3] 0 Enables ADC measurement of GPI (general purpose input) while in backup mode. adc_backup_iin_en [4] 0 Enables ADC measurement of input current while in backup mode. adc_backup_vout_en [5] 0 Enables ADC measurement of vout while in backup mode. adc_backup_vcap_en [6] 0 Enables ADC measurement of vcap while in backup mode. adc_backup_vin_en [7] 0 Enables ADC measurement of vin while in backup mode. adc_backup_vcap1_en [8] 0 Enables ADC measurement of vcap1 while in backup mode. adc_backup_vcap2_en [9] 0 Enables ADC measurement of vcap2 while in backup mode. adc_backup_vcap3_en [10] 0 Enables ADC measurement of vcap3 while in backup mode. adc_backup_vcap4_en [11] 0 Enables ADC measurement of vcap4 while in backup mode. adc_backup_ch_en_ reg 7 R/W adc_wait_vin 8 R/W [15:0] 100 Sets the wait time between ADC measurement groups while in charging mode. The LSB of this register has a weight of 400uS. The ADC measures all enabled channels then waits this time before measuring all channels again. The ADC data is used for balancing and shunting, increasing this time reduces the shunt and balancer update rate and is not typically recommended if shunting or balancing is enabled. If shunting or measuring capacitance/ESR this time may be ignored by the ADC. 400uS per LSB adc_wait_backup 9 R/W [15:0] 100 Sets the wait time between ADC measurement groups while in backup mode. The LSB of this register has a weight of 400uS. The ADC measures all enabled channels then waits this time before measuring all channels again. 400uS per LSB gpi_uv_lvl 10 R/W [15:0] 0 General Purpose Input Under Voltage Level: This is an alarm threshold for the GPI pin. If enabled, the GPI pin voltage falling below this level will trigger an alarm and an SMBALERT. 182.8V per LSB gpi_ov_lvl 11 R/W [15:0] 0 General Purpose Input Over Voltage Level: This is an alarm threshold for the GPI pin. If enabled, the GPI pin voltage rising above this level will trigger an alarm and an SMBALERT. 182.8V per LSB vin_uv_lvl 12 R/W [15:0] 0 VIN Under Voltage Level: This is an alarm threshold for the input voltage. If enabled, the input pin voltage falling below this level will trigger an alarm and an SMBALERT. 2.19mV per LSB vin_ov_lvl 13 R/W [15:0] 0 VIN Over Voltage Level: This is an alarm threshold for the input voltage. If enabled, the input pin voltage rising above this level will trigger an alarm and an SMBALERT. 2.19mV per LSB vcap_uv_lvl 14 R/W [15:0] 0 VCAP Under Voltage Level: This is an alarm threshold for the capacitor stack voltage. If enabled, the VCAP pin voltage falling below this level will trigger an alarm and an SMBALERT. 1.46mV per LSB vcap_ov_lvl 15 R/W [15:0] 0 VCAP Over Voltage Level: This is an alarm threshold for the capacitor stack voltage. If enabled, the VCAP pin voltage rising above this level will trigger an alarm and an SMBALERT. 1.46mV per LSB 44 Rev. A For more information www.analog.com LTC3351 REGISTER MAP Command Code Access Bit Range vout_uv_lvl 16 R/W [15:0] 0 VOUT Under Voltage Level: This is an alarm threshold for the output voltage. If enabled, the VOUT pin voltage falling below this level will trigger an alarm and an SMBALERT. 2.19mV per LSB vout_ov_lvl 17 R/W [15:0] 0 VOUT Over Voltage Level: This is an alarm threshold for the output voltage. If enabled, the VOUT pin voltage rising above this level will trigger an alarm and an SMBALERT. 2.19mV per LSB dtemp_cold_lvl 18 R/W [15:0] 0 Die Temperature Cold Level: This is an alarm threshold for the die temperature. If enabled, the die temperature falling below this level will trigger an alarm and an SMBALERT. Temperature = 0.0295C per LSB - 274C dtemp_hot_lvl 19 R/W [15:0] 0 Die Temperature Hot Level: This is an alarm threshold for the die temperature. If enabled, the die temperature rising above this level will trigger an alarm and an SMBALERT. Temperature = 0.0295C per LSB - 274C ichg_uc_lvl 20 R/W [15:0] 0 Charge Undercurrent Level: This is an alarm threshold for the charge current. If enabled, the charge current falling below this level will trigger an alarm and an SMBALERT. 1.955V/Rsnsc per LSB iin_oc_lvl 21 R/W [15:0] 0 Input Overcurrent Level: This is an alarm threshold for the input current. If enabled, the input current rising above this level will trigger an alarm and an SMBALERT. 1.955V/Rsnsi per LSB cap_uv_lvl 22 R/W [15:0] 0 Capacitor Under Voltage Level: This is an alarm threshold for each individual capacitor voltage in the stack. If enabled, any capacitor voltage falling below this level will trigger an alarm and an SMBALERT. 182.8V per LSB. cap_ov_lvl 23 R/W [15:0] 0 Capacitor Over Voltage Level: This is an alarm threshold for each individual capacitor in the stack. If enabled, any capacitor voltage rising above this level will trigger an alarm and an SMBALERT. 182.8V per LSB cap_lo_lvl 24 R/W [15:0] 0 Capacitance Low Level: This is an alarm threshold for the measured stack capacitance. If the measured stack capacitance is less than this level it will trigger an alarm and an SMBALERT, if enabled. When ctl_cap_scale is set to 1, capacitance is 3.36F * RT/RTST per LSB. When ctl_cap_scale is set to 0 it is 336F * RT/RTST per LSB. esr_hi_lvl 25 R/W [15:0] 0 ESR High Level: This is an alarm threshold for the measured stack ESR. If enabled, a measurement of stack ESR exceeding this level will trigger an alarm and an SMBALERT. Rsnsc/64 per LSB. esr_i_on_settling 26 R/W [15:0] 2 Time to allow the charging current to settle before measuring the charge voltage and current for ESR. Each LSB is 1024 switcher periods. esr_i_off_settling 27 R/W [15:0] 8 Time to wait after turning the charge current off before measuring the charge voltage and current for ESR. Each LSB is 1024 switcher periods. esr_i_override 28 R/W [15:0] 0 This value overrides the LTC3351's adaptive test current selection for the ESR test. If this register is non-zero, the lower 8 bits will be used as an 8 bit DAC value to set the charge current during the ESR test. Typically this register will not need to be set. ITEST = 32mV * (esr_i_override[7:0] + 1) / 256 / Rsnsc cap_i_on_settling 29 R/W [15:0] 8 Time to wait after turning the test current on before measuring the first voltage of the capacitance measurement. Each LSB is 1024 switcher periods. cap_delta_v_setting 30 R/W [15:0] 550 Symbol Name Default Description The target delta V for the capacitance test. The scale is 182.8V per LSB. The default is approximately 100mV. Rev. A For more information www.analog.com 45 LTC3351 REGISTER MAP Command Code Access Bit Range min_boost_cap_ voltage 31 R/W [15:0] 0 If this register is non-zero, it sets the minimum capacitor voltage the boost will operate at. If any capacitor voltage falls below this value in boost mode the boost will be forced off, the boost will not turn back on even if the capacitor voltage rises above this voltage. Only after input power returns will the boost be re-enabled. This prevents the boost from cycling on and off many times once the capacitors' voltage has discharged to the point it can no longer support the system load through the boost. To use this feature vcap[1:num_caps+1] measurements must be enabled in backup mode, see adc_backup_ch_en_reg. Also the capacitor voltages are only measured as often as set by adc_wait_backup. min_vout_hs_disable 32 R/W [15:0] 0 If this register is non-zero, it sets the minimum voltage VOUT is allowed to reach while the HotSwap is disabled. If the voltage falls below this level the ctl_hotswap_disable bit will be cleared, re-enabling the HotSwap controller. To use this feature the VOUT measurement must be enabled in boost mode, see adc_backup_ch_en_reg. Also the VOUT voltage is only measured as often as set by adc_wait_backup. alarm_reg 35 R/W [15:0] 0 Alarms Register: A one in any bit in the register indicates its respective alarm has triggered. All bits are active high. Alarms are cleared by clearing (writing 0) the appropriate bit in this register. Setting (writing 1) bits has no effect. For example to clear the alarm_gpi_uv alarm, write 0xFFFD. alarm_gpi_uv [0] 0 GPI Under Voltage alarm alarm_gpi_ov [1] 0 GPI Over Voltage alarm alarm_vin_uv [2] 0 VIN Under Voltage alarm alarm_vin_ov [3] 0 VIN Over Voltage alarm alarm_vcap_uv [4] 0 VCAP Under Voltage alarm alarm_vcap_ov [5] 0 VCAP Over Voltage alarm alarm_vout_uv [6] 0 VOUT Under Voltage alarm alarm_vout_ov [7] 0 VOUT Over Voltage alarm alarm_dtemp_cold [8] 0 Die temperature cold alarm alarm_dtemp_hot [9] 0 Die temperature hot alarm alarm_ichg_uc [10] 0 Charge undercurrent alarm alarm_iin_oc [11] 0 Input overcurrent alarm alarm_cap_uv [12] 0 Capacitor Under Voltage alarm alarm_cap_ov [13] 0 Capacitor Over Voltage alarm alarm_cap_lo [14] 0 Capacitance low alarm alarm_esr_hi [15] 0 ESR high alarm [15:0] N/A Monitor Status: This register provides real time status information about the state of the monitoring system. Each bit is active high. mon_meas_active [0] N/A Capacitance/ESR measurement is active. This bit becomes one at the begining of a capacitance/ESR measurement and remains 1 after the measurement has completed until the capacitors have been discharged back to their regualtion voltage. mon_capesr_scheduled [1] N/A Indicates that the LTC3351 is waiting programmed time to begin a capacitance/ ESR measurement mon_capesr_pending [2] N/A Indicates that the LTC3351 is waiting for satisfactory conditions to begin a capacitance/ESR measurement mon_cap_done [3] N/A Indicates that the capacitance measurement has completed mon_esr_done [4] N/A Indicates that the ESR Measurement has completed Symbol Name monitor_status_reg 46 36 R Default Description Rev. A For more information www.analog.com LTC3351 REGISTER MAP Symbol Name Command Code Access Bit Range Default Description mon_meas_failed [5] N/A Indicates the last attempted capacitance and ESR measurement was unable to complete mon_boost_shutdown [6] N/A This bit is set in boost mode when any capacitor falls below min_boost_cap_ voltage_reg. It is cleared when power returns. mon_disable_charger [7] N/A Indicates the capacitance and ESR measurement system has temporarily disabled the charger. mon_cap_meas_active [8] N/A Indicates the capacitance and ESR measurement system is measuring capacitance. mon_esr_meas_active [9] N/A Indicates the capacitance and ESR measurement system is measuring ESR. mon_power_failed [10] N/A This bit is set when VIN is outside the UV/OV range or the HotSwap controller is disabled by setting the ctl_hotswap_disable. It is cleared only when mon_power_ returned is set. mon_power_returned [11] N/A This bit is set when the output is powered by the input and the charger is able to charge. It is cleared only when mon_power_failed is set. mon_balancing [12] N/A Indicates the LTC3351 is balancing the capacitor voltage. mon_shunting [13] N/A Indicates a capacitor voltage is approaching vshunt and a shunt is turned on. mon_cap_precharge [14] N/A Indicates the capacitor stack is being precharged for a capacitance measurement. mon_reset [15] N/A This bit is set during a power on reset. It is cleared on any I2C/SMBus write. It can be used to determine if the chip has reset during a power loss followed by a power return. meas_gpi 37 R [15:0] N/A Measurement of GPI pin voltage. 182.8V per LSB meas_vin 38 R [15:0] N/A Measured Input Voltage. 2.19mV per LSB meas_vcap 39 R [15:0] N/A Measured Capacitor Stack Voltage. 1.46mV per LSB. meas_vout 40 R [15:0] N/A Measured Output Voltage. 2.19mV per LSB. meas_dtemp 41 R [15:0] N/A Measured die temperature. Temperature = 0.0295C per LSB - 274C meas_ichg 42 R [15:0] N/A Measured Charge Current. 1.955V/Rsnsc per LSB meas_iin 43 R [15:0] N/A Measured Input Current. 1.955V/Rsnsi per LSB lo_vcap 44 R [15:0] N/A The lowest measured capacitor voltage from the last measurement set. hi_vcap 45 R [15:0] N/A The highest measured capacitor voltage from the last measurement set. meas_cap 46 R [15:0] N/A Measured capacitor stack capacitance value. When ctl_cap_scale is set to 1, capacitance is 3.36F * RT/RTST per LSB. When ctl_cap_scale is set to 0 it is 336F * RT/RTST per LSB. meas_esr 47 R [15:0] N/A Measured capacitor stack equivalent series resistance (ESR) value. Rsnsc/64 per LSB meas_vcap1 48 R [15:0] N/A Measured voltage between the CAP1 and CAPRTN pins. 182.8V per LSB meas_vcap2 49 R [15:0] N/A Measured voltage between the CAP2 and CAP1 pins. 182.8V per LSB meas_vcap3 50 R [15:0] N/A Measured voltage between the CAP3 and CAP2 pins. 182.8V per LSB meas_vcap4 51 R [15:0] N/A Measured voltage between the CAP4 and CAP3 pins. 182.8V per LSB. When the ITST current is on, either due to ctl_force_itst_on or during a capacitance measurement, this voltage measurement will temporarily be low due to the ITST current flowing in the shunt resistor. cap_m0_vc1 52 R [15:0] N/A The voltage change on cap1 due to the capacitance measurement. The relative voltage change on each capacitor during the capacitance measurement and the total capacitance can be used to calculate the capacitance of each individual capacitor. Rev. A For more information www.analog.com 47 LTC3351 REGISTER MAP Symbol Name Command Code Access Bit Range cap_m0_vc2 53 R [15:0] N/A The voltage change on cap2 due to the capacitance measurement. The relative voltage change on each capacitor during the capacitance measurement and the total capacitance can be used to calculate the capacitance of each individual capacitor. cap_m0_vc3 54 R [15:0] N/A The voltage change on cap3 due to the capacitance measurement. The relative voltage change on each capacitor during the capacitance measurement and the total capacitance can be used to calculate the capacitance of each individual capacitor. cap_m0_vc4 55 R [15:0] N/A The voltage change on cap4 due to the capacitance measurement. The relative voltage change on each capacitor during the capacitance measurement and the total capacitance can be used to calculate the capacitance of each individual capacitor. esr_m0_vc1 56 R [15:0] N/A A measurement of VCAP1 just before turning current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m0_vc2 57 R [15:0] N/A A measurement of VCAP2 just before turning current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m0_vc3 58 R [15:0] N/A A measurement of VCAP3 just before turning current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m0_vc4 59 R [15:0] N/A A measurement of VCAP4 just before turning current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m1_vc1 60 R [15:0] N/A The first VCAP1 voltage measurement with charge current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m1_vc2 61 R [15:0] N/A The first VCAP2 voltage measurement with charge current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m1_vc3 62 R [15:0] N/A The first VCAP3 voltage measurement with charge current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m1_vc4 63 R [15:0] N/A The first VCAP4 voltage measurement with charge current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m1_i 64 R [15:0] N/A The first charge current measurement with charge current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m2_vc1 65 R [15:0] N/A The second VCAP1 voltage measurement with the charge current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m2_vc2 66 R [15:0] N/A The second VCAP2 voltage measurement with the charge current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m2_vc3 67 R [15:0] N/A The second VCAP3 voltage measurement with the charge current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m2_vc4 68 R [15:0] N/A The second VCAP4 voltage measurement with the charge current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m2_i 69 R [15:0] N/A The second charge current measurement with charge current on for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m3_vc1 70 R [15:0] N/A The VCAP1 voltage measurement with charge current off for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m3_vc2 71 R [15:0] N/A The VCAP2 voltage measurement with charge current off for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m3_vc3 72 R [15:0] N/A The VCAP3 voltage measurement with charge current off for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m3_vc4 73 R [15:0] N/A The VCAP4 voltage measurement with charge current off for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. esr_m3_i 74 R [15:0] N/A The charge current measurement with charge current off for the ESR measurement. This value is used by the LTC3351 in the calculation of meas_esr. 48 Default Description Rev. A For more information www.analog.com LTC3351 REGISTER MAP Command Code Access Bit Range rev_code 80 R [15:0] 5 The LTC3351 revision code. next_esr_i 84 R [7:0] 32 The 8 bit DAC setting for the charge current that the LTC3351 has calculated for the next ESR measurement based on the previous ESR measurement. The first ESR measurement will use a setting of 32. If esr_i_override is non-zero, this register will be calculated but esr_i_override will be used instead. If non-zero ITEST = 32mV * (next_esr_i[7:0] + 1) / 256 / Rsnsc num_caps 237 R [1:0] 0 Number of Capacitors. This register shows the state of the CAP_SLCT1, CAP_SLCT0 pins. The value read in this register is the number of capacitors programmed minus one. sys_status 238 R [11:0] N/A System Status Register: This register provides real time status information about the instantaneous state of the system. Each bit is active high. stepdown_mode [0] N/A The synchronous controller is in step-down mode (charging) stepup_mode [1] N/A The synchronous controller is in step-up mode (backup) chrg_cv [2] N/A The charger is in constant voltage mode chrg_uvlo [3] N/A The charger is in under-voltage lockout or has been disabled by ctl_force_charger_off. chrg_input_ilim [4] N/A The charger is in input current limit cappg [5] N/A The capacitor voltage is above power good threshold boost_en [7] N/A Indicates the boost is enabled buck_en [8] N/A Indicates the charger is enabled chrg_ci [9] N/A Indicates the charger is in constant current mode vingd [11] N/A Indicates the input voltage is inside the UV/OV range. Symbol Name Default Description Revision: 149 Date: 2017-08-18 15:51:04 -0400 (Fri, 18 Aug 2017) Note: All registers are 16 bits. Unused bits not shown above should be written as 0 and ignored when reading Rev. A For more information www.analog.com 49 50 For more information www.analog.com T C5 1F 10k NTC R4 10k VINGD CAPGD SMBALERT SCL SDA VIN 20V TO 30V D1 SMBJ30A CC 1.2nF R1 10k VDD RPF3 28k RPF3 14.3k RPF1 665k D2 RT 107k R2 10k RVIN 100 10 RG2 10 MN5 SiS434DN D4 R4 1 R5 1 D5 RSNSH RSNSI 0.004 0.012 R6 1 CTIMER GND PGND VINGD CAPGD SMBALERT SCL SDA VCC2P5 CAP_SLCT0 CAP_SLCT1 GPI VC RT ITST OV UV VIN Ct 33nF RETRYB LTC3351 CAPFB CAPRTN CAP1 CAP2 CAP3 CFN VCAPP5 CAP4 VCAP CFP ICAP BGATE SW TGATE BST VOUT OUTFET OUTFB INTVCC DRVCC SRC HS_GATE ISNSP_CHG ISNSP_HS ISNSM CSS D3 RTST 35.7 R3 10k RG1 10 MN1 SiS434DN Css 22nF CF 0.1F 220pF CB 0.1F DB 1N4448HWT 24V Input, 18V 36W Backup MN3 SiS434DN L1 6.8H RCAPRTN 2.7 RCAP1 2.7 RCAP2 2.7 RCAP3 2.7 RSNSC 0.005 COUT1 33F x2 RCAP4 2.7 RFB02 46.4k MN2 SiS434DN C3 4.7F RFB01 649k CAP1-4: NESSCAP ESHSR-0010C0-002R7 L1: COILCRAFT XAL7070-682ME D2: PDS1040L-13 D3-D5: DIODES INC 1N5819HW CCP5 0.1F C4 0.1F CFBO1 120pF CAP1 10F CAP2 10F CAP3 10F + + + + CCAP 22F x4 CAP4 10F COUT2 10F x4 36W IN BACKUP 3351 TA02 RFBC2 118k RFBC1 866k VOUT 18V LTC3351 TYPICAL APPLICATIONS Rev. A C5 1F For more information www.analog.com NTC R4 10k VINGD CAPGD SMBALERT SCL SDA VIN 22V TO 30V D1 SMBJ30A RT 107k R2 10k RVIN 100 BACKUP_DONE CC 4.7nF R1 10k VDD RPF3 28k RPF3 11k RPF1 665k D2 R3 10k RG1 10 10 RG2 10 MN5 SiS434DN D4 R4 1 R5 1 D5 RSNSH RSNSI 0.004 0.012 R6 1 CTIMER GND PGND VINGD CAPGD SMBALERT SCL SDA VCC2P5 CAP_SLCT0 CAP_SLCT1 GPI VC RT ITST OV UV VIN Ct 33nF RETRYB LTC3351 CAPFB CAPRTN CAP1 CAP2 CAP3 CFN VCAPP5 CAP4 VCAP CFP ICAP BGATE SW TGATE BST VOUT OUTFET OUTFB INTVCC DRVCC SRC HS_GATE ISNSP_CHG ISNSP_HS ISNSM CSS D3 RTST 35.7 MN1 SiS434DN Css 22nF CF 0.1F 220pF CB 0.1F DB 1N4448HWT RFB02 73.2k RCAP1 100k RCAP2 100k RCAP3 100k RCAP4 100k MN3 SiS434DN MN2 SiS434DN C3 4.7F RFB01 665k L1 6.8H NTC CP1 8.33F 16.2V T RT1 10k COUT2 10F x4 + + + + + + CCAP 22F x4 36W IN BACKUP MN4 SiS434DN RSNSC 0.005 COUT1 33F x2 CP1: TECATE GROUP PBLS-8.33/16.2 L1: COILCRAFT XAL7070-682ME D2: PDS1040L-13 D3-D5: DIODES INC 1N5819HW CCP5 0.1F C4 0.1F CFBO1 120pF 24V Input, 12V 36W Backup, 6 Capacitor Stack 3351 TA03 RFBC2 100k RFBC1 1.15M VOUT 12V LTC3351 TYPICAL APPLICATIONS Rev. A 51 52 C5 1F VIN -18V TO 18V For more information www.analog.com T RT1 100k R4 100k PFO CAPGD SMBALERT SCL SDA RPF3 49.9k RPF3 33.2k RPF1 665k D1 CC 10nF R1 10k VDD DIN BATS4 RVIN 1k DQ2 1n4148 RT 71.5k R2 10k RQ2 10k Q1 2n3904 R3 10k RHS_GATE 100k RGATE2 10 RSNSI 0.008 CTIMER GND PGND VINGD CAPGD SMBALERT SCL SDA VCC2P5 CAP_SLCT0 CAP_SLCT1 GPI VC RT ITST OV UV VIN Ct 33nF RETRYB LTC3351 CAPFB CAPRTN CAP1 CAP2 CAP3 CFN VCAPP5 CAP4 ICAP VCAP CFP BGATE SW TGATE BST VOUT OUTFET OUTFB INTVCC DRVCC SRC HS_GATE ISNSP_CHG ISNSP_HS ISNSM CSS RSRC 100k RTST 35.7 D2 RGATE1 10 MN5 MN1 SiS438DN SiS438DN Css 22nF CF 0.1F CB 0.1F DB 1N4448HWT MN3 BSZ060NE2LS L1 3.3H RCAPRTN 2.7 RCAP1 2.7 RCAP2 2.7 RCAP3 2.7 CAP1 10F CAP2 10F CAP3 10F CAP4 10F COUT2 2.2F x2 + + + + CCAP 22F x4 25W IN BACKUP MN4 SiS438DN RSNSC 0.006 COUT1 47F x2 RCAP4 2.7 RFB02 162k MN2 BSZ060NE2LS C3 4.7F RFB01 649k CAP1-4: NESSCAP ESHSR-0010C0-002R7 L1: COILCRAFT XAL7030-332ME D1: SMBJ18CA D2: 1N4148 CCP5 0.1F C4 0.1F CFBO1 120pF 12V Backup Controller with Reverse Input Protection RRT2 52.3k RFBC2 118k 3351 TA04 RRT1 1M RFBC1 866k VOUT 6V LTC3351 TYPICAL APPLICATIONS Rev. A C5 1F For more information www.analog.com T RT1 100k R4 100k PFO CAPGD SMBALERT SCL SDA SMAJ10A VIN 4.8V TO 12V CC 4.7nF 2k R1 10k VDD RPF3 4.02k RPF3 6.04k RPF1 30.1k R2 10k RTST 20 R3 10k RGATE 10 RT 88.7k RVIN 100 MN1 SiS452DN RSNSI 0.005 CTIMER GND PGND VINGD CAPGD SMBALERT SCL SDA VCC2P5 CAP_SLCT0 CAP_SLCT1 GPI VC RT ITST OV UV VIN Ct 33nF RTRY2 10k RTRY1 200k RETRYB LTC3351 VOUT CAPFB CAPRTN CAP1 CAP2 CAP3 CFN VCAPP5 CAP4 ICAP VCAP CFP BGATE SW TGATE BST VOUT OUTFET OUTFB INTVCC DRVCC SRC HS_GATE ISNSP_CHG ISNSP_HS ISNSM CSS 10 MN5 SiS452DN Css 22nF CF 0.1F CB 0.1F DB 1N4448HWT MN3 SiS452DN L1 1H RCAPRTN 2.7 RCAP1 2.7 RSNSC 0.003 COUT1 100F x6 RCAP2 2.7 RFB02 210k MN2 SiS452DN C3 4.7F RFB01 665k CAP1-4: NESSCAP ESHSR-0010C0-002R7 L1: COILCRAFT XAL7030-332ME CCP5 0.1F C4 0.1F CFBO1 100pF 4.8V to 12V 10A Supercap Charger with 5V 30W Backup Mode CAP1 50F + + CCAP 22F x4 CAP2 50F COUT2 2.2F x2 5V 30W 3351 TA05 RFBC2 118k RFBC1 866k VOUT 5V LTC3351 TYPICAL APPLICATIONS Rev. A 53 LTC3351 PACKAGE DESCRIPTION UFF Package 44-Lead Plastic QFN (4mm x 7mm) (Reference LTC DWG # 05-08-1500 Rev O) 0.70 0.05 4.50 0.05 3.10 0.05 2.40 REF 5.64 0.05 2.64 0.05 PACKAGE OUTLINE 0.20 0.05 5.60 REF 6.10 0.05 7.50 0.05 0.40 BSC RECOMMENDED SOLDER PAD LAYOUT APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 4.00 0.10 0.75 0.05 PIN 1 NOTCH R = 0.30 TYP OR 0.35 x 45 CHAMFER 2.40 REF 43 0.00 - 0.05 44 0.40 0.10 1 2 PIN 1 TOP MARK (SEE NOTE 6) 2.64 0.10 5.64 0.10 7.00 0.10 5.60 REF R = 0.10 TYP R = 0.10 TYP 0.74 0.10 (UFF44) QFN REV 0 0415 0.200 REF R = 0.10 TYP 0.20 0.05 0.40 BSC BOTTOM VIEW--EXPOSED PAD NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 54 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE Rev. A For more information www.analog.com LTC3351 REVISION HISTORY REV DATE DESCRIPTION A 08/19 Split FSW specification into room temperature and overtemperature and changed limits Hot Swap Component Selection: Changed verbiage from "reduced" to "increased" PAGE NUMBER 3 26 Rev. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license For is granted implication or otherwise under any patent or patent rights of Analog Devices. more by information www.analog.com 55 LTC3351 TYPICAL APPLICATION 12V PCLE Backup Controller MN5 SiS438DN MN1 SiS438DN VIN 11V TO 18V RSNSI 0.008 25W IN BACKUP RGATE 10 D1 SMBJ18A RVIN 100 Css 22nF SRC HS_GATE ISNSP_CHG ISNSP_HS ISNSM CSS VOUT OUTFET OUTFB INTVCC DRVCC VIN RPF1 665k UV RPF3 33.2k OV RPF3 49.9k BST TGATE VDD CFBO1 120pF DB 1N4448HWT C4 0.1F CB 0.1F R2 10k R3 10k PFO CAPGD SMBALERT SCL SDA R4 100k C5 1F T RT1 100k CC 10nF RT 71.5k RTST 35.7 LTC3351 ICAP VCAP CFP CFN VCAPP5 CAP4 C3 4.7F CF 0.1F L1 3.3H RSNSC 0.006 CCAP 22F x4 RCAP4 2.7 CAP2 CAP1 CAPRTN CTIMER RFB02 162k CCP5 0.1F CAP3 GND PGND COUT2 2.2F x2 COUT1 47F x2 MN3 BSZ060NE2LS BGATE VINGD CAPGD SMBALERT SCL SDA VCC2P5 CAP_SLCT0 CAP_SLCT1 GPI VC RT ITST RFB01 649k MN2 BSZ060NE2LS SW R1 10k VOUT 6V MN4 SiS438DN RCAP3 2.7 CAP4 10F RCAP2 2.7 CAP3 10F RCAP1 2.7 CAP2 10F RCAPRTN 2.7 CAP1 10F + + + + CAPFB RETRYB Ct 33nF CAP1-4: NESSCAP ESHSR-0010C0-002R7 L1: COILCRAFT XAL7030-332ME RFBC1 866k RRT1 1M RFBC2 118k RRT2 52.3k 3351 TA06 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS Power Management LTC3350 Bidirectional Controller, Monitor and PowerPath for Supercapacitors ICHARGE = 10A+, VIN: 4.5V to 35V, VOUT: 4V to 35V LTC3225 Boost Charge Pump Supercapacitor Charger ICHARGE = 0.15A, VIN: 2.8V to 5.5V, VOUT: 4.8V to 5.3V LTC3226 Boost Charge Pump and PowerPath Manager for Supercapacitors ICHARGE = 0.33A, VIN: 2.5V to 5.5V, VOUT: 2.5V to 5.5V LTC3625 Buck and Boost Supercapacitor Charger ICHARGE = 1A, VIN: 2.7V to 5.5V, VOUT: 4V to 5.3V LTC3355 Buck Supercapacitor Charger and Boost Backup ICHARGE = 1A, VIN: 3V to 20V, VOUT: 2.7V to 5.5V LTC3110 Bidirectional Buck-Boost and PowerPath Manager for Supercapacitors ICHARGE = 2A, VIN: 0.1V to 5.5V, VOUT: 1.8V to 5.5V LTC3643 Bidirectional Boost Charger/Buck Backup, Electrolytic Capacitors ICHARGE = 2A, VIN: 3V to 17V, VOUT: Up to 40V LTC4040 Buck Battery Charger and Boost Backup for Li Batteries ICHARGE = 2.5A, VIN: 2.5V to 5.5V, VOUT: 3.5V to 5V LTC3128 Buck-Boost Supercapacitor Charger ICHARGE = 3A, VIN: 1.7V to 5.5V, VOUT: 1.8V to 5.5V LTC4425 Linear/Ideal Diode Supercapacitor Charger ICHARGE = 3A, VIN: 1.7V to 5.5V, VOUT: 2.7V to 5.5V LTC4110 Bidirectional Buck-Boost Flyback Controller ICHARGE = 3A, VIN: 4.5V to 19V, VOUT: 2.7V to 19V 56 Rev. A 08/19 www.analog.com For more information www.analog.com ANALOG DEVICES, INC. 2017-2019