Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Power Devices in TV & Monitor Applications
(including selection guides)
319
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
4.1.1 An Introduction to Horizontal Deflection
Introduction
This section starts with the operation of the power
semiconductors in asimple deflection test circuit leading to
afunctionalexplanationofa typicalTV horizontaldeflection
circuit. Theoperationofthe commoncorrectioncircuitsare
discussed and the secondary function of the horizontal
deflection circuit described.
Deflection Test Circuit
The horizontal deflection test circuit used to assess Philips
deflection transistors is shown in Fig. 1 below. Lc
represents the horizontal deflection coils.
Fig. 1. Test Circuit for Deflection Transistors
This circuit is a simplification of a practical horizontal
deflectioncircuit. Itcan beused toproducethe voltageand
currentwaveformsseenbyboththetransistorandthediode
in a real horizontal deflection circuit. It is, therefore, very
useful as a test circuit for switching times and power
dissipation. Thewaveformsproducedbythetestcircuitare
shown in Fig. 2.
Fig. 2. Test Circuit Waveforms
Cycle of Operation
Briefly going through one cycle of operation, the sequence
of events is as follows. (This can be followed through on
the waveforms shown in detail in Fig. 3, by starting on the
left and following the stages numbered 1 to 8).
1. Turn on the deflection transistor by applying a positive
current drive to the base. The voltage on the collector is
nowapproximately0.5Vbecausethedeviceisfullyon. This
means that the voltage across the coil, Lc, is the full line
voltage; in this case 150V.
2. According to the law, V = L • dI/dt, the current in the coil
Lc will now start to rise with a gradient given by 150V/Lc.
This portion of the coil current (ILc), is the sawtooth portion
of the collector current in the transistor (Ic).
3. Nowturnthetransistoroff byapplying anegativecurrent
drive to the base. Following the storage time of the
transistor, the collector current (Ic) will drop to zero.
4. The current in Lc (ILc) is still flowing! This current,
typically 4.5Afor testing the BU2508A, cannotflow through
the transistor any more, nor can it flow through the reverse
biased diode, BY228. It, therefore, flows into the flyback
capacitor, Cfb, and so the capacitor voltage rises as ILc
falls. Because Cfb is connected across the transistor, the
rise in capacitorvoltage is seen as a rise in Vce acrossthe
transistor.
Lc will transfer all its energy to Cfb. The capacitor voltage
reaches its peak value, typically 1200V,at the point where
ILc crosses zero.
5. Now we have a situation where there is zero energy in
Lc but there is a very large voltage across it. So ILc will
rise, and since this current is supplied by Cfb, the voltage
across Cfb falls. This is, of course, a resonant LC circuit
and essentially it is energy which is flowing, first from the
inductor, Lc, to the capacitor, Cfb, and then from the
capacitor, Cfb, to the inductor, Lc. Note that the current in
Lc is now flowing in the opposite direction to what it was
previously. It is, therefore, a negative current.
6. Thisresonance wouldcontinue,with thecoil currentand
the capacitor voltage followingsinusoidal paths, were it not
for the diode, BY228. When the capacitor voltage starts to
go negative the diode becomes forward biased and
effectively clamps the capacitor voltage to approximately
-1.5V, the diode VF drop. This also clamps the voltage
across Lc to approximately the same value as it was when
the transistor was conducting, ie the line voltage (150V).
Note that the coil current is now being conducted by the
diode, and hence ILc = Idiode.
IBon
-VBB
LB
Lc
HVT
Cfb BY228
+150V nominal
adjust for Icm
Ib Ib
Ic Ic Ic Ic
ILc ILc
Tscan Tfb
Idiode
Vce Vce
ILc=Idiode
Idiode
Tfb
ILc=Ic
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7. So we have again a current ramp in Lc with a dI/dt equal
to150/Lc. Thiscurrentstartswithavalueequal tothevalue
it had at the end of the transistor on time (neglecting circuit
losses). It is, however, flowing in the opposite (negative)
direction and so the positive dI/dt will bring it back towards
zero.
8. Before ILc actually reaches zero, the base drive is
re-appliedto the transistor. This meansthat whenILc does
reach zero, we arrive back at the same conditions we had
atthebeginningofstage1;iethetransistorison,thecurrent
in Lc is zero and the voltage across Lc is the line voltage
(150V).
TV Operating Principle
In a television set, or a computer monitor, the picture
information is written onto the screen one line at a time.
Each of these horizontal lines of picture information is
written onto the screen by scanning the screen from left to
rightwithanelectronbeam. Thiselectronbeamisproduced
byagunsituatedatthebackofthetube,anditisaccelerated
towardsthescreen byahighpotential(typically25kV). The
beam is deflected from left to right magnetically, by varying
the current in a set of horizontal deflection coils positioned
between the gun and the screen.
The screen is phosphor coated, andwhen the high energy
electron beam strikes the phosphor coating the phosphor
givesoffvisiblelight. Thedensityofelectronsintheelectron
beam can be varied: phosphor brightness depends on
beam density, and so the instantaneous brightness of the
scanning spot can be varied at a fast rate as each line of
picture information is written onto the screen. A set of
vertical deflection coils deflect the beam vertically at the
end of each horizontal scan and so lines of picture
informationcanbebuiltup,one aftertheother. The vertical
deflection frequency (or
field rate
) for European sets is
50 Hz(alternate line scanning, giving 25complete screens
of information per second).
With no current in the horizontal deflection coils, the
magnetic field between them is zero and so the electron
beam hits the centre ofthe screen. With a negative current
inthecoils,theresultantmagneticfielddeflectstheelectron
beam to the left side of the screen. With a positive coil
current the deflection is to the right.
Now consider the characteristic deflection waveforms,
Fig. 3. The current ILc represents the current in the
horizontal deflection coils. During the period where the
current in the deflection coils is ramping linearly from its
peak negative value to its peak positive value, the electron
beam is scanning the screen from left to right. This is the
scan time, Tscan.
Fig. 3. Test Circuit Waveforms
Ib Ib
Ic Ic Ic Ic
ILc ILc
Tscan Tfb
Idiode
Vce Vce
ILc=Idiode
Idiode
Tfb
1
2
3
4
5
6
7
8
ILc=Ic
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During the period where the horizontal deflection coil
current flows into the flyback capacitor, and then back into
thecoil (thehalfcosinecurve attheend ofthescanperiod),
the electron beam is rapidly moving from the right side of
the screen tothe left. This is called the flyback period, Tfb,
and no information is written onto the screen during this
part of the cycle.
S-Correction
The actual TV horizontal deflection circuit differs from the
test circuit in a number of ways that improve the picture
quality. Thesimplifieddeflectioncircuit showninFig. 1 can
be redrawn as shown in Fig. 4 where Lc is the horizontal
deflectionyokeandCsischargedtothelinevoltage(150V).
Fig. 4. Simplified TV Deflection Circuit
The advantage of this arrangement is that, by carefully
selecting the value of Cs, one form of picture distortion is
corrected for as follows.
The front of the TV tube is flat, rather than curved, and so
during each horizontal scan the electron beam travels a
greater distance to the edges of the screen than it does to
the middle. A linear deflection coil current would tend to
over deflect the beam as it travelled towards the edges of
the screen. This would result in a set of ’equidistant’ lines
appearing on the screen as shown in Fig. 5 below.
Fig. 5. Distortion Caused By Flat Screens
The voltage on the capacitor, Cs, will be modulated by the
deflection coil current, ILc. When the diode is in forward
conduction and the current in Lc is ’negative’, the voltage
on Cs will rise as Cs becomes more charged. When the
transistor is conducting and the current in Lc is ’positive’,
the voltageon Cswill drop as Csdischarges. This isshown
in Fig. 6 below.
Fig. 6. S-Correction
This will give an ‘S’ shape to the current ramp in the
deflection coils which corrects for the path difference
between the centre and the edge of a flat screen tube.
Hence the value of the capacitor, Cs, is quite critical. Cs
is known as the S-correction capacitor.
Linearity Correction
Fig. 7. Asymmetric Picture Distortion
Caused by Coil Resistance
The voltage across the deflection coil is also modulated by
the voltage drop across the series resistance of the coil.
This parasitic resistance (RLc) causes an asymmetric
picture distortion. A set of ‘equidistant’ vertical lines would
appear on the screen as shown in Fig. 7. The voltage
across the coils is falling as the beam scans the screen
VCs
ILc
150V
ILc
ILc=0
Lc
Cfb
Cs
+
150V
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from left to right. The beam, therefore, travels more slowly
towards theright side of the screen and the lines are drawn
closer together, see Fig. 8.
Fig. 8. Effect of Coil Resistance on Voltage Across Coil
VRLc is the voltage drop across the resistive component
of Lc. Subtracting this from the voltage across Cs (VCs)
gives the voltage across the inductive component on the
deflection coils (VLc). To compensate for the voltage drop
across the parasitic coil resistance we need a component
with a negative resistance to place in series with the coil.
This negative resistance effect is mimicked by using a
saturableinductance,Lsat,inserieswiththedeflectioncoils
as shown in Fig. 9 below.
Fig. 9. Position of Saturable Inductor in Circuit
For an inductor with a low saturation current, the
relationship between inductance and current is as shown
in Fig. 10. As the current is increased much above zero,
the core saturates and so the inductance drops. This
happens if the current is conducted in either direction.
Fig. 10. L/I Characteristic of a Saturable Inductance
By taking a saturable inductance and premagnetising the
core, we add a dc bias to this characteristic as shown in
Fig. 11 below.
Fig. 11. DC Shift Produced by Premagnetised Core
Since Lsat has a much lower inductance than Lc, the dI/dt
through Lsat is governed by the deflection coils, and is
thereforedILc/dt. The voltagedropacross Lsatistherefore
given by V = Lsat•dILc/dt. During the scan time, Tscan,
dILc/dt is approximately constant in value, and so the
voltage/currentcharacteristics of Lsat during thescan time
are as shown in Fig. 12 below.
This is the characteristic required and so the voltage
developed across Lsat, the linearity correction coil,
compensates for the series resistance of the deflection
coils.
VCs
ILc
150V
0V
VRLc
VLc
ILc
ILc=0
150V
L
Icoil
I=0
L
Icoil
I=0
Cfb +Vcc (150V)
Lsat
Lc
Cs
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Fig. 12. V/I Characteristic of Lsat During Tscan
Cs Losses
So the circuit shown in Fig. 9 now gives the desired
deflectionwaveforms. Theelectronbeamscansthescreen
at a uniform rate on each horizontal scan. However, the
circuit is not lossless and unless Cs is kept topped up the
dcvoltage on Cs, Vcc, willgradually decay. To prevent this
from happening a voltage supply can be added across Cs
but this introduces other problems.
Fig. 13. Deflection Circuit with Voltage Supply Added
The average voltage across Lc
must
be zero. Adc voltage
acrossLcwouldgenerateadccurrentwhichwouldproduce
apicture shift to theright. Applying Vccdirectly toCs would
result in a dc current component through the deflection
coils, so Cs must be charged to Vcc by some other way.
Applying Vcc to Cs via the deflection coils overcomes this
problem.
A large choke inductance, Lp, in series with the Vcc supply
isnecessarytopreventanenormousincreaseinthecurrent
through the power switch. Without it the Vcc supply would
be shorted out every time the transistor was turned on.
Typically the arrangement shown in Fig. 13 will result in a
20% increase in the current through the power switch and
the power diode.
East-West Correction
So to recap on the circuit so far: the series resistance of
the deflection coils is compensated for by the linearity
correction coil, Lsat, and the varying length of the electron
beam path, as the beam scans the screen from left to right,
is compensated for by the S-correction capacitor, Cs. This
capacitor modulatesthe voltage across the deflection coils
during each horizontal scan, modulating the magnetic field
ramp between them, and thus keeping the speed at which
the electron beam scans the screen constant.
However, as the picture information is written onto the
screen, by writing one line of information after another, a
furthervariationinthe lengthofthebeampath isintroduced
as the beam scans the screen from top to bottom. The
length of the beampath to the edge of the screen is shorter
when the central lines of picture information are being
written than it is when the lines at the top or the bottom of
the screen are being written.
This means that ahigher peak magnetic field is required to
deflect the beam to the screen edges when the beam is
writing the central lines of picture information, than that
required to deflect the beam to the screen edges when the
lines of picture information at the top and bottom of the
screen are being written.
Fig. 14. Modulation of the Peak Deflection Coil Current
This requires increasing the peak deflection coil current
gradually over the first half of each vertical scan, and then
reducing itgradually overthe laterhalf of eachvertical scan
(seeFig. 14). Thisisdonebymodulatingthevoltageacross
the deflection coils. This process is known as east-west
correction.
Thelinevoltage,Vcc,issuppliedbyawindingontheSMPS
transformer. This voltage is regulated by the SMPS and
during the operation of the TV set it is constant.
In order to achieve the required modulation of the voltage
across the deflection coils, a simple linear regulator could
be added in series with Lp. One disadvantage of this
solution is that it increases the circuit losses.
The Line Output Transformer (LOT)
Thehorizontaldeflectiontransistorservesanotherpurpose
as well as deflecting the beam: driving the line output
transformer (LOT). The LOT has a number of low voltage
outputs but its primary function is to generate the EHT
voltages to accelerate and focus the electron beam.
V
II=0
Centre Bottom
+4.5A
+3.5A
-4.5A
-3.5A
etc
Top
of screen
1 vertical
scan 2nd vertical
scan
deflection coil
current is
modulated
Cfb
+Vcc (150V)
Lsat
Vcc
(150V)
Lp (typ 5mH)
(typ 12nF) Lc (typ 1mH)
Cs
(typ 500nF)
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Fortunately, this function can be combined with a feature
previously described as a cure for Cs losses; the inductive
choke, Lp. The LOT has a large primary inductance that
serves the purpose of Lp so a separate choke is not
required, see Fig. 15 below.
Fig. 15. Position of Line Output Transformer
Alotof powermaybedrawnfromtheLOTbutthedeflection
must not be affected. In order to keep the secondary
windings of the LOT at fixed voltages, we need to keep the
voltage across the primary winding fixed. Therefore, the
requirements for the LOT and a regulated supply to Lp are
in conflict.
‘Real’ and ‘Dummy’ Deflection Circuits
As a way around this problem, consider a ‘dummy’
deflection circuit in series with the ‘real’ deflection circuit.
This enables one circuit to meet the requirements for
deflection, including east-west correction, and the dummy
circuit meets the requirements for the LOT, see Fig. 16.
Fig. 16. Position of Dummy Deflection Circuit.
The two deflection circuits operate in direct
synchronisation. Vmod is a voltage between zero and 30V
that controls the east-west correction. Thus we can vary
thevoltage acrossthe deflectioncoilsinthe‘real’ deflection
circuit without varyingthe voltage across the primary ofthe
LOT in the ‘dummy’ circuit.
Fig 17. Vmod Applied Directly to Cmod
For properflyback tuning, the‘real’ and ‘dummy’ deflection
circuits and the LOT must be tuned to the same flyback
frequency. The two deflection circuits are tuned through
careful selection of the flyback capacitors. In the case of
the LOT the capacitance of the windings provides the
necessary capacitance (typically 2nF) for correct tuning.
Since Lmod is only an inductor and not a real deflection
component, a net dc current through it is not a problem.
Therefore, we can apply Vmod directly to Cmod and this
way reduce the component count by removing Lpmod, see
Fig. 17.
Lmod is a quarter of the value of Lc. Cfbmod is four times
as big as Cfb. Cmod is not critical as long as it is large
enough to supply the required energy.
Suppose there is no voltage supplied externally to Cmod.
The supply voltage, Vcc, will split according to the ratio of
the impedances of the two circuits. In fact, the Vcc will split
according tothe ratio of thetwo flyback capacitors,Cfb and
Cfbmod, as shown in Fig. 18.
The average voltage across Cfbmod will automatically be
30V(for Vcc = 150V), ifno external voltages are applied to
the ‘dummy’ circuit. Consequently, Cmod will become
charged to 30V. The two deflection circuits are always
operating in direct synchronisation. Under the condition
where Vmod is 30V the currents in the two circuits would
also be equal.
Lc
Cfb
Cs
+Vcc (150V)
Lsat
Vcc
(150V)
E.H.T.
Line output
transformer
Lc
Cfb
Cs +Vcc-Vmod
Lsat
Vcc
E.H.T.
Line output
transformer
Lmod
Cmod
Cfbmod +Vmod
Lc
Cfb
Cs +Vcc-Vmod
Lsat
Vcc
E.H.T.
Line output
transformer
Lmod
Cmod
Cfbmod
Vmod +Vmod
Lpmod
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Fig. 18. Average Voltage Across the two
Flyback Capacitors.
The range of Vmod required is 0 to 30V. Vmod is reduced
below 30V as current is drawn from Cmod. An external
supply to Cmod is never needed. This is the arrangement
used in practice.
To draw current from Cmod a series linear regulator is
added across Cmod as shown in Fig. 19.
Fig. 19. Series Regulator Controlling Vmod
Diode Modulator Circuit
The circuit is now quite close to an actual TV horizontal
deflection circuit. As the two transistors are switching in
perfect synchronisation this circuit can be simplified further
by removing one transistor, as shown in Fig. 20. This
arrangement makes no difference to the operation of the
circuit.
The circuit in Fig. 20 now shows all the features of the
horizontal deflection diode modulator circuit. These
features should be distinguishable when studying actual
circuit diagrams.
Fig. 20. The diode modulator circuit
Diode Modulator: Upper Diode
Firstconsiderthevoltagerequirements. Inthisrespect, the
worst case conditions for the upper diode are when
Vmod = 0V. Under these conditions the upper diode must
supportthefullflybackvoltage. Therefore,thepeakvoltage
limiting value on the upper diode must match the VCES limit
of the transistor.
Now consider the current requirements. With no circuit
losses, the currents in the diode and the transistor are as
showninFig. 21whereIcisthetransistorcurrentandIdiode
is the diode current. Of this current, 80% flows in the
deflection coils and 20% flows in the LOT primary.
Fig. 21. With no load, Ic and Idiode are equal.
With circuit losses included, the transistor current will
exceed the diode current. Circuit losses add a dc
component tothe waveform shown in Fig. 21. The loading
on the LOT contributes a further dc component, increasing
the transistor current and reducing the diode current still
further, Fig. 22.
+Vcc
Cfb
Cfbmod (= 4*Cfb)Vmod
4*Vmod
Lc
Cfb
Cs +Vcc-Vmod = 120-150V
Lsat
Vcc
E.H.T.
Line output
transformer
Lmod
Cmod
Cfbmod +Vmod = 0-30V
Series
regulator
(150V)
Lc
Cfb
Cs +Vcc-Vmod = 120-150V
Lsat
Vcc
E.H.T.
Line output
transformer
Lmod
Cmod
Cfbmod +Vmod = 0-30V
Series
regulator
Ic Ic
Idiode
Idiode
Idiode Ic
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Fig. 22. With load and circuit losses, Ic > Idiode.
For example, for a current which is 12A peak to peak, 10A
of thiswill be deflection current and 2A will beLOT current.
With no load, the peak diode current would be equal to the
peak transistor current, ie both would equal 6A. However,
the LOT requires 1A dc in order to power the secondary
windings. This makes the peak diode current 5A and the
peak transistor current 7A. These are practical values for
a 32 kHz black line S (ie EHT = 30kV) TV set.
The diode must conduct the full current immediately after
theflyback period. Untilthe
forward
recoveryvoltageofthe
diode has been reached the diode cannot conduct. A high
forward recovery voltage device would impede the start of
the scan. If once the forward recovery voltage has been
reached the device takes a long time before falling to its VF
level then the voltage across the deflection coils would be
non-linear and, therefore, cause picture distortion. For a
32 kHz set the diode must recover to less than 5V in under
0.5µs.
Diode Modulator: Lower Diode
First consider the voltage requirements. At one extreme,
all the flyback voltage is across the top diode and at the
other extreme, the worst case condition for the lower diode
is when the flyback voltage is split between the two diodes
in the ratio 4:1 (ie when Vmod is at its maximum value of
30V). The voltage limiting value on the lower diode is,
therefore, usually given as one quarter of the rating of the
top diode. So, if the transistor is 1500V, the top diode is
also 1500V and the bottom diode is 400V.
However, it is not uncommon for fault conditions to occur
in TV circuits that cause large voltage spikes on the lower
diode. To accommodate such occurrences, a 600V device
is often used as the lower diode.
Now consider the current requirements. The lower diode
musttakethesame currentasthehorizontal deflectioncoil,
Fig. 23, and so its current requirement is the same as that
of the top diode.
Fig. 23. Situation when Vmod = 0V
As shown in Fig. 23, the lower diode is conducting its peak
currentimmediately before theflyback periodand switches
off as thetransistor. Therefore, the
reverse
recovery of the
lower diode must be very fast to minimise circuit losses.
Diode Modulator Circuit Example
Putting the above diode requirements into the circuit of
Fig. 20 enable a typical 16 kHz TV horizontal deflection
circuit to be constructed, see Fig. 24. This circuit is
representative of a modern 25" TV design. The deflection
transistor, BU2508A, will run with a peak ICof 4.5A at
16 kHz. The combined inductance and capacitance will
produce a flyback pulse of typically 1200V peak and 13µs
width. The upper diode, BY228, has the same current and
voltage capability as the deflection transistor. The lower
diode, BYW95C, has the same current capability but a
reduced voltage rating. More often than not, 600V devices
are used as the lower diode with 1500V upper diodes.
ThedcsupplycomesfromtheTVSMPScircuit. TheSMPS
will use a power switch also, typically BUT11AF in 16 kHz
TV. Atransformerwillprovideallthehighpowerdcsupplies
requiredforthe TV. Fora150Vsupply ahighvoltage diode
will be used in the output stage, typically BY229-600. The
LOT generates the EHT to accelerate the electron beam,
typically a voltage of 25kV is produced.
InsmallerTV’s(14-21")thiscircuitcouldbemuchsimplified.
Forsmaller screensizes EW correction isnot essential and
the diode modulator is not usually present. The circuit now
uses a single diode and capacitor. The diode can be
incorporated in the deflection transistor, for example, the
BU2508D. Also for the smaller screen sizes it is common
that the tube technology allows for lower flyback voltages.
In these applications the 1000V BUT11A and BUT12A are
often used.
In larger 16 kHz TV’s (28" and above) and all 32 kHz TV’s
the axial diodes will not normally be capable in terms of
current handling. These diodes are replaced by devices in
TO220type power outlines: BY359 forthe upper diode and
Ic
Idiode
Idiode
Ic
No losses
With
losses
Lc
Cfb
Cs
Lsat
ON
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BY229-600 or BYV29-500 for the lower diode. Also larger
deflection transistors are available: BU2520A and
BU2525A.
The same sort of scaling is also applicable to monitor
deflectioncircuits. Allmonitorstendtouse1500Vdeflection
transistorsandupper diodes. TheBU2508D (iewithdiode)
is often used so that the BY228 can be used as the upper
diode. Using a D-type deflection transistor takes some
current from the diode modulator but does not affect the
operatingprinciple. For the high frequency (up to 82 kHz),
multi-sync monitors BU2522A and BU2527A deflection
transistors are used. Above 64 kHz the BY459 is used as
an upper diode.
Fig. 24. The Diode Modulator for 16 kHz TV Deflection, Example
Linearity
E.H.T.
Line output
transformer
+Vmod
Series
regulator
BU2508A
SMPS
BUT11AF
Line scan
coils
0-30V
500nF
2uF
250uH
BY228
BYW95B
6.8nF
22nF
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Philips Semiconductors
4.1.2 The BU25XXA/D Range of Deflection Transistors
Introduction
The BU25XXA range forms the heart of Philips
Semiconductors 1500V power bipolar transistors. This
technology offers world class dissipation in its target
applicationof 16 kHz TV horizontal deflection circuits. The
range has been extended for state-of-the-art large screen
TV (8 A, 32 kHz) and all the volume monitor applications
(up to 6 A, 64 kHz). The successful application of the
BU25XXA range in all sectors of TV & monitor horizontal
deflection has proved it to be a global technology.
As a further improvement, the BU25XXD range of devices
have been introduced. Using BU25XXA technology,
horizontal deflection transistors incorporating base-emitter
resistive damping and a collector-emitter damper diode
have been produced. The devices in this range are
specifically aimed at the small-screen 16 kHz TV and
48 kHz monitor applications where the use of a D-type
devicecanofferasignificantcost-saving. TheD-typesoffer
the same performance as the A-type equivalent with only
slightly increased dissipation, at a similar cost.
Specification Notes
The ICsat value defines the peak current reached in a
horizontal deflection circuit during normal operation for
which optimum performance is obtained. Unlike other
specification points, it is not necessary to inset this value
inareal application. Operationeither muchaboveorbelow
the specified ICsat value will result in less than optimum
performance. For higher frequencies the ICsat should be
lowered to keep the dissipation down.
TheVCESM valuedefinesthepeakvoltage appliedunderany
condition. The BU25XXA/D range could operate under
continuousswitchingto1500Vinadeflectioncircuitwithout
any degradation to performance but exceeding 1500V is
neither recommended nor guaranteed. In normal running
the peak flyback voltage is typically 1150V but a 1500V
device is required for fault conditions.
The storage time, ts, and fall time, tf, limits are given for
operation at the ICsat value and the frequency of operation
given by the application limit.
The BU25XXA Range Selection Guide
Specification Application
Device ICsat VCESM tstfTV Monitor
BU2508A/AF/AX 4.5 A 1500 V 6.0 µs 600 ns ≤ 25", 16 kHz -
4.0 A 1500 V 5.5 µs 400 ns - 14", SVGA, 38 kHz
BU2520A/AF/AX 6.0 A 1500 V 5.5 µs 500 ns ≤ 29", 16 kHz 15", SVGA, 48 kHz
4.0 µs 350 ns ≤ 28", 32 kHz
BU2525A/AF/AX 8.0 A 1500 V 4.0 µs 350 ns ≤ 32", 32 kHz (17", 64 kHz)
BU2522A/AF/AX 6.0 A 1500 V 2.0 µs 250 ns - 15", 64 kHz
BU2527A/AF/AX 6.0 A 1500 V 2.0 µs 200 ns - 17", 64 kHz
The BU25XXD Range Selection Guide
Specification Application
Device ICsat VCESM tstfTV Monitor
BU2506DF/DX 3.0 A 1500 V 6.0 µs 500 ns ≤ 23", 16 kHz -
BU2508D/DF/DX 4.5 A 1500 V 6.0 µs 600 ns ≤ 25", 16 kHz 14", SVGA, 38 kHz
BU2520D/DF/DX 6.0 A 1500 V 5.5 µs 500 ns ≤ 29", 16 kHz 15", SVGA, 48 kHz
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Televisions and Monitors Power Semiconductor Applications
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Application Notes
Theapplicationsgivenintheselectionguideshouldbeseen
as an indication of the limits that successful designs have
been achieved for that device type. This should help in the
selection of a device for a given application at the design
concept stage. For example, a 15" monitor requiring
operation up to 6 A at 64 kHz could use either a BU2522A
or BU2527A. If the design has specific constraints on
switching and dissipation then the BU2527A is the best
option, but if cost is also a prime consideration then the
smaller chip BU2522A could be used with only slightly
degraded performance. For an optimised design the
BU2525A can be used in 17", 64 kHz applications but the
BU2527A is the recommended choice.
Outlines
Philips Semiconductors recognise both the varying design
criteria and the market availability of device outlines and
this is reflected in the range of outlines offered for the
BU25XXA/D range. Three different outlines are offered for
the devices available, one non-isolated (SOT93) and two
isolated/full-pack designs (SOT199, TOP3D). The outline
is defined by thelast letter in the type number, for example:
BU2508A SOT93 non-isolated
BU2508AF SOT199 isolated
BU2508AX TOP3D isolated
All three outlines are high quality packages manufactured
to Philips Total Quality Management standards.
The Benefits of the D-type
Fig.1. BU2508A vs. BU2508D
The BU2508D technology incorporates the damper diode
andabase-emitterdampingresistor,seeFig.1. Inthetarget
16 kHz applications the damper diode is usually an axial
type (eg. BY228), the D-type deflection transistor
incorporates this device in a monolithic structure. This
presents a significant cost-saving in the application. The
base-emitter resistor eliminates the need for external
damping at the transistor base-emitter. The only
consideration for replacing an A-type with a D-type is that
thebase current required for optimised switching is slightly
higher for the D-type.
Forhighercurrentsandfrequencieswherediodemodulator
circuits are used it appears at first that use of the D-types
is not possible. However, this is not so; D-type transistors
can be used WITH diode modulators in a beneficial way.
Forexample, a 15", 48 kHz SVGA monitor utilising a diode
modulator is at the borderline between an axial upper
damper diode and a TO220 type. The dissipation is such
that if an axial diode is used some sort of thermal
management may be necessary. By using a D-type
transistor some of the current is taken by the diode in the
D-type relieving the discrete upper damper device. Use of
a D-type in this way has allowed an axial diode to be used
in place of a TO220 type making a significant cost saving.
Causes of Dissipation
In the cycle of operation there are four distinct phases:
turn-on, on, turn-off, off. Each phase is a potential cause
ofdissipation. Ofcourse,forenhanced circuitperformance
dissipation in the deflection transistor must be minimised.
a) Turn-on. The primary function of a deflection transistor
is to assist in the sweep of the beam across the screen of
the display, ie to horizontally deflect the beam. As the
deflection transistor turns-on the beam is scanning from
just less than half way across. At mid-screen the beam is
un-deflected, ie the deflection current is zero. So, the
deflectiontransistorturns on witha smallnegativecollector
current, ICramping up through zero. At turn-on there are
no sudden severe load requirements that cause significant
dissipation. In horizontal deflection turn-on dissipation is
negligible.
b) On-state. As the beam is deflected from the centre of
the screen to the right - hand side the ICincreases as
determined by the voltage across the deflection coil. The
resulting voltage drop across the deflection transistor, VCE
depends not only on this ICbut also on the base drive: for
high IBthe VCE will be low; for low IBthe VCE will be high.
For high IBthe transistor is said to be overdriven giving low
on-state dissipation. For low IBthe transistor is said to be
underdriven giving higher on-state dissipation.
c) Turn-off. Turn-off starts whenthe forward IBisstopped.
Thisis followed 2 - 6 µs later (depending on the device and
application) by the ICpeaking. This delay is called the
storagetime, tsof the device. During this time the VCE rises
as the current rises causing increased dissipation: the
longer tsthe higher the dissipation. As the ICpeaks so
scanning ends and the process of flyback begins. Now, as
the ICfalls (in time tf, the fall-time) the VCE rises to the peak
flyback voltage; this a phase of high dissipation.
D2
Rbe
BU2508A, BE resistor, damper diode BU2508D
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Televisions and Monitors Power Semiconductor Applications
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Turn-off is the dominant loss phase for all deflection
transistors. The device characteristics in this phase are of
much interest to the TV & monitor design engineers.
d) Off-state. In an optimised drive circuit the device will
be off for VCE above 250V in flyback. For the rest of the
flyback the collector-emitter is reverse biased while the
base-emitterwillalso bereversebiased:between-1to -4V.
Any leakage through the device will be the cause of
dissipation. For the BU25XXA/D range, low-contaminant
processing ensuresthat the bulkleakageis very low. Also,
the long-established Philips glasspassivation has very low
leakage. The device characteristics coupled with the low
pulse width and duty cycle of the flyback mean that the
losses in the off-state are negligible.
InaD-type deflectiontransistorthereisan additionalcause
of dissipation:
e) Diode Conduction. At the end of flybackthe next scan
willstart. As theflyback voltage goesnegative sothe diode
conducts, this clamps the voltage on the flyback capacitor.
The fixed voltage provides a fixed ramp in current through
the deflectioncoil andthrough the diode;thebeam sweeps
from the left towards the centre of the screen. At, or near,
the centre this current approaches zero ending the diode
conduction phase. The dissipation here is dominated by
two characteristics of the diode: the forward recovery and
the on-state voltage drop. This can be a significant cause
of dissipation in a D-type transistor.
The effect of both underdrive and overdrive on the device
is increased device dissipation and hence increased
junction temperature. In general, the higher the junction
temperature the shorter the lifetime of the device in the
application. Optimised drive circuit design and good
thermal management can bring the device junction
temperature down to well below the limit Tjmax. Such
considerations enhance the reliability of the deflection
transistorinthe application. It isessential that care istaken
atthedesignstage tooptimisethe basedrive forthe device
product spread.
Dynamic Testing
The BU25XXA/D range is assessed in a deflection
switching test circuit designed to simulate the most
demanding running conditions of the application. The
horizontal deflection coils, which form themajor part of the
collector load, are represented by a variable inductance Lc
and the flyback and diode modulator circuits by a single
diode, BY228 and variable capacitance, Cfb. For
BU2508A/AF/AX TV applications the test circuit is shown
in Fig.2 below.
This circuit generates the characteristic deflection
waveforms,Fig.3,fromwhichthe storagetime,falltimeand
energylossatturn-offcanbemeasured. Theseparameters
define the device performance in the application.
Fig.2. BU2508A/AF/AX 16 kHz Deflection
Switching Test Circuit
Fig.3. Deflection Switching Waveforms
It is not valid to do a single-shot test for the switching
parameters as the characteristics of the nth pulse depend
on the previous (n-1)th pulse. To achieve this the tester
works in a double pulse mode.
‘Bathtub’ Curves
Fig.4. BU2508A Typical Turn-Off Losses
- ‘Bathtub’ curve
IBon
-VBB
LB
Lc
HVT
Cfb BY228
+150V nominal
adjust for Icm
Ib Ib
Ic Ic Ic Ic
ILc ILc
Tscan Tfb
Idiode
Vce Vce
ILc=Idiode
Idiode
Tfb
ILc=Ic
0.1 1 10
IB / A
Eoff / uJ BU2508A
1000
100
10
3.5A
IC = 4.5A
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Televisions and Monitors Power Semiconductor Applications
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A plot of base current, IB, against turn off dissipation, Eoff,
for one BU2508A measured in the switching test circuit at
a peak collector current of 4.5A gives the characteristic
‘bathtub’ shape shown in Fig.4 above. From this curve the
tolerance to base drivevariations can be assessed and the
optimum IBdetermined for a given IC.
The switching performance is also determined by the peak
reverse base current at switch off. For a typical hFE device,
of all types in the BU25XXA/D range, a peak reverse base
current, IBoff, equal to one half the typical peak ICis
recommended for optimum dissipation. This is largely
determined by the drive transformer and is usually difficult
tobefine-tuned. Inthetypicalnon-simultaneousbasedrive
circuit the level of forward base current, IB, is easily
controlled, hence, the presentation of the turn-off losses
versus IB.
The ‘bathtub’ curves are plotted for a reverse base voltage
at turn-off of -4V. This level of reverse base drive is
recommended for the BU25XXA/D range as it reduces the
risk of any noise, or ring, forward biasing the base-emitter
during flyback. However, in well-engineered designs the
BU25XXA/D can operate just as well with a reverse base
voltage at turn-off of only -1V. This tolerance to base drive
is very useful to design engineers.
On the far left of the curve, at low IBvalues, the device is
severelyunderdriven resulting in a high turnoff dissipation.
As the base drive is increased the degree of underdrive is
reduced and the device remains in saturation for a larger
proportion of its on time. This is the reason for the initial
decrease in Eoff with increasing IBseen in the ‘bathtub’
curve. Eventually, the optimum drive is reached and the
turnoffdissipation, Eoff,isatitsminimumvalue. Increasing
the base drive still further results in overdrive and the
appearance of an ICtail at turn off. The result of this, as
can be seen in the ‘bathtub’ curve, is increasing turn off
losses with increasing IB.
Typically, this curve has steep sides and a flatter central
portion; this gives it the shape of the cross-section through
a bathtub, hence the name ‘bathtub’ curve !
The BU25XXA/D technology gives a sharper looking curve
but a much lower level of Eoff/Poff than competitor types.
For optimised drive the BU25XXA/D technology offers
world class dissipation in 16 kHz TV deflection circuits.
Process Control
The success of the BU25XXA/D range has enabled
significant enhancements to be made to the benefit of both
our customers and ourselves. By utilising a continuous
cycle of quality improvement coupled with high volume
production, Philips Semiconductors can demonstrate their
excellent process control in specified hFE and dissipation
limits. This control is achieved bymanufacturing capability
rather than test selections. This process control improves
manufacturing throughput and yield and, hence, customer
deliveries. The improvements in manufacturing result in
higher process capability indices enabling the introduction
of tightened internal test specifications.
Critical Parameter Distribution Fact Sheets
Industry standard data sheets for all power semiconductor
devices offer an introduction to the device fundamentals
and can usually be used for a quick comparison between
competitortypes. Detaileduseofa specificdevicerequires
muchmore information thanis contained inany data sheet.
This is particularly relevant to high voltage bipolar
transistors,and especially horizontaldeflection transistors.
Ahorizontaldeflectiontransistorisonlyasgoodasthebase
circuit that drives it. The growth of power MOSFET’s is
mainly due to the difficulties in driving bipolar transistors.
However,MOSFETtechnologyisnotsuitableforhorizontal
deflection applications, Philips Semiconductors are
actively involved in supplying the support tools necessary
for the successful design-in of their BU25XXA/D range.
Recognising the designers requirements Philips
Semiconductorsnowprovidecritical parameterdistribution
factsheets for the BU25XXA/D range. This additional data
should be used in conjunction with the data sheets to give
a full picture of the device capabilities and characteristics
over the production spread.
The fact sheets give limit curves for the power dissipation
in the device caused by turn-off, Poff, at a given operating
frequencyand range ofload current, ICall at 85˚C(a typical
operating temperature for TV and monitor applications).
These curves provide limits to the typical ‘bathtub’ curves
given in data. It is important to recognise that these fact
sheet curves represent the LIMIT of production when
comparing the BU25XXA/D range with competitor types
which offer this information as TYPICAL only, if at all. This
information displays the technical performance of the
device and the measurement capability available.
Contained in the fact sheets is evidence of the world class
dissipation limits obtained by the BU25XXA/D range. As
an example,the BU2508A/AF/AX ‘bathtub’limit curves are
shown in Figs.5-7.
These fact sheetsalso containlimit hFE curves for VCE =1 V
and 5 V at three different temperatures: -40˚C, 25˚C, and
85˚C. Therangeoftemperatureschosenreflects therange
of customer requirements. These limit curves define the
device characteristics for all the important extremes of
operation. As an example the BU2508A/AF/AX limit hFE
curves for VCE = 1 V and 5 V at 25˚C are shown in Figs.8-9
below. The 100% test points are indicated by arrows.
334
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Fig.5. BU2508A/AF/AX Fig.6. BU2508A/AF/AX Fig.7. BU2508A/AF/AX
Max. Poff vs. IBMax. Poff vs. IBMax. Poff vs. IB
for IC = 3.5 A @ 85˚C for IC = 4.5 A @ 85˚C for IC = 5.5 A @ 85˚C
0.4 0.6 0.8
10
1
0.1
Poff (W)
Tj = 85C
f = 16kHz
IC = 3.5A
BU2508A/AF
1.0
IB (A)
0.5 1.5
10
1
0.1 1.0 2.0
Poff (W)
Tj = 85C
f = 16kHz
IC = 4.5A
BU2508A/AF
IB (A) 1 2 3
10
1
0.1 1.5 2.5
Poff (W) BU2508A/AF
IC = 5.5A
Tj = 85C
f = 16kHz
IB (A)
Fig.8. BU2508A/AF/AX hFE vs IC @ 1 V, 25˚C Fig.9. BU2508A/AF/AX hFE vs IC @ 5 V, 25˚C
100
10
1
hFE
0.01 IC (A)0.1 1.0 10
BU2508A/AF
VCE = 1V
Tj = 25C
100
10
1
hFE
0.01 IC (A)0.1 1.0 10
BU2508A/AF
VCE = 5V
Tj = 25C
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Televisions and Monitors Power Semiconductor Applications
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Drive Circuits
It was stated previously that a horizontal deflection
transistor is only as good as the base circuit that drives it.
Philips Semiconductors address this problem by providing
fact sheets with an example of a drive circuit for the target
applications. Thedrivecircuitspresentedareoftheindustry
standard non-simultaneous base drive type utilising
commercially available Philips components. An example
of these drive circuits is shown for the BU2508A in Fig.10
below.
This drive circuit is not an end in itself but a means to an
end: it producesthe waveforms at the base that enablethe
load set by Vcc, Lc and Cfb to be switched most efficiently.
For this reason the waveforms produced by this circuit are
also presented in the fact sheet. Again, for the BU2508A
example given above the waveforms are shown in
Figs.11-15 below.
Thedrive circuit employed in the application could be quite
different tothe one givenabove in Fig.10 but thebase drive
waveformsinFigs.11 & 13must be replicatedfor optimised
switching.
The fundamental concept of the non-simultaneous base
drive is well established in the TV and monitor industry for
driving the horizontal deflection transistor. Individual
designs, however, can differ significantly. A different
transformer design may enable the required base current
to be generated without the addition of Lb and D1, Rb.
Driving Rp from a low voltage supply could reduce thecost
by allowing a low voltage transistor, Q1 and capacitor, Cd
to be used.
The resistor Rbe is necessary to eliminate any overshoot
in the Vbe at the end of the base-emitter avalanche that
could turn the transistor on during flyback. Such an event
would lead to an early failure of the transistor by exceeding
theforwardbiasedsafe-operatingarea(FBSOA). Incircuits
operating at higher frequencies resistive damping alone is
usually not sufficient and RC damping is required.
In this application, the BU2508A could be replaced by the
BU2508D in the circuit of Fig.10. This would allow the
BY228 and Rbe resistor to be removed from the circuit.
Fig.10. BU2508A 4.5 A, 16 kHz Drive Circuit
Components and values: Rp = 1 k
Ω
, 2 W; Cr = 100 nF; Rd = 560
Ω
; Cd = 470 pF, 500 V; Q1 Philips BF819;
T1 Philips AT4043/87 Transformer; Lb = 0.5
µ
H; Rb = 0.5
Ω
, 0.5 W; D1 Philips BYV28-50; Rbe = 50
Ω
;
Lc = 1 mH; Cfb = 12 nF, 2 kV; D2 Philips BY228; Vcc = 115 V.
Lc
Cfb
Vcc
0V
D.U.T.
Rb
Cd
Rd
Rp
T1
Q1
Vp
Vce
Vbe
D1
Rbe D2
0V
Cr
0V
Lb
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Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Fig.11. BU2508A/AF IB vs. time Fig.12. BU2508A/AF IC vs. time
Fig.13. BU2508A/AF VBE vs. time Fig.14. BU2508A/AF VCE vs. time
IB(end) = 0.9 - 1.1 A; ICmax = 4.5
±
0.25 A;
|- IB(off)|
≥
2.0 A;
VCEmax = 1100 - 1200 V
Fig.15. VP vs. time
BU2508A/AF
IB (1A/div)
4
2
0
-2
-4 time (10us/div)
BU2508A/AF
time (10us/div)
IC (1A/div)
4
2
0
-2
BU2508A/AF
time (10us/div)
5
0
-5
-10
-15
VBE (5V/div)
BU2508A/AF
time (10us/div)
VCE (200V/div)
0
200
400
600
800
1000
1200
1400
time (10us/div)
VP (50V/div)
200
150
100
50
0
250 BU2508A/AF
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Televisions and Monitors Power Semiconductor Applications
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4.1.3 Philips HVT’s for TV & Monitor Applications
This section simplifies the selection of the power switches
required for the SMPS and horizontal deflection in TV and
monitor applications. Both high voltage bipolar transistors
and power MOSFET devices are included in this review.
As well as information specific to the Philips
Semiconductorsrangeofdevices,generalselectioncriteria
are established.
HVT’s for TV & Monitor SMPS
The vast majority of television and monitors have switch
mode power supplies that are required to generate an
90 - 170Vsupplyforthelinedeflectionstage,plusanumber
of lower voltage outputs for audio, small signal etc. By far
the easiest and most cost effective way of fulfilling these
requirements is to use a flyback topology. Discontinuous
mode operation is generally preferred because it offers
easier control and smaller transformer sizes than
continuous mode.
For the smaller screen size TV’s, where cost is a dominant
factor, bipolar HVT’s dominate. For large screen TV and
monitors power MOSFET’s are usually chosen.
Thepeakvoltageacrosstheswitchingtransistorinaflyback
converteristwicethepeakdclinkvoltage
plus
anovershoot
voltage which is dependent on the transformer leakage
inductanceand thesnubber capacitance. Thus,foragiven
mainsinputvoltagethereisaminimumvoltagerequirement
onthetransistor. Increasingthetransformerleakageand/or
reducing the snubber capacitance will increase the
minimum voltage requirement on the transistor.
a) Power MOSFET’s
For TV’s operating just with 110/120V mains applications
adevicewhichcan beused withpeakvoltages below400V
is required. For these applications the power MOSFET is
used almost exclusively. A wide variety of 400V power
MOSFET’s are available, leading to lower device costs,
which coupled with the easier drive requirements of the
MOSFET make this an attractive alternative to a bipolar
switch.
For 220/240V and, more recently, for universalinput mains
applications where an 800V device is generally required
the cost of power MOSFET was prohibitive. However,
improvements both in circuit design and device quality has
meantthata600Vdevicecanbeusedintheseapplications.
Philips Semiconductors have a comprehensive range of
powerMOS devices for these applications. The main
parameters of these devices most applicable to TV and
monitor SMPS applications are summarised in Table 1.
Part Number VDSS RDS(ON) @ ID
BUK454-400B 400 V 1.8 Ω1.5 A
BUK455-400B 400 V 1.0 Ω2.5 A
BUK457-400B 400 V 0.5 Ω6.5 A
BUK454-600B 600 V 4.5 Ω1.2 A
BUK455-600B 600 V 2.5 Ω2.5 A
BUK457-600B 600 V 1.2 Ω6.5 A
BUK454-800A 800 V 6.0 Ω1.0 A
BUK456-800A 800 V 3.0 Ω1.5 A
BUK438-800A 800 V 1.5 Ω4.0 A
Table 1. Philips PowerMOS HVT’s for TV & Monitor
SMPS Applications
The VDSS value is the maximum permissible drain-source
voltage of the powerMOS in accordance with the Absolute
Maximum System (IEC 134). The RDS(ON) value is the
maximum on-state resistance of the powerMOS at the
specified drain current, ID.
b) Bipolar HVT’s
Bipolar HVT’s still have an important role in TV SMPS
applications. Many new TV designs are slight
improvements on existing designs incorporating a new
controlorsignalfeature(egFastext,SCARTsockets)which
do not demand the re-design of the SMPS. If there has
been a good experience with an existing SMPS it is not
surprising that these designs should continue in new TV
models.
For 220/240V mains driven flyback converters, generally,
a 1000V bipolar HVT is used. The full voltage capability of
the transistor can be used as the limit under worst case
conditionsbut it mustnever be exceeded. Incircuits where
the transformer leakage inductance is high, and voltages
inexcessof1000Vcanoccur,adevicewithahighervoltage
handling capability is required.
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Televisions and Monitors Power Semiconductor Applications
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Philips Semiconductors have a comprehensive range of
bipolar HVT devices for these applications. The main
parameters of these devices most applicable to TV SMPS
applications are summarised in Table 1.
Part VCESM VCEO ICsat VCEsat
Number
BUX85 1000 V 450 V 1 A 1.0 V
BUT11A 1000 V 450 V 2.5 A 1.5 V
BUT18A 1000 V 450 V 4 A 1.5 V
BUT12A 1000 V 450 V 5 A 1.5 V
BUW13A 1000 V 450 V 8 A 1.5 V
BU506 1500 V 700 V 3 A 1.0 V
BU508A 1500 V 700 V 4.5 A 1.0 V
Table 2. Philips Bipolar HVT’s for TV SMPS Applications
The VCESM value is the maximum permissible
collector-emitter voltage of the transistor when the base is
shorted to the emitter or is at a potential lower than the
emittercontact. TheVCEO valueisthemaximumpermissible
collector-emitter voltage of the transistor when the base is
open circuit. Both voltage limits are in accordance with the
Absolute Maximum System (IEC 134). The VCEsat value is
the maximum collector emitter saturation voltage of the
transistor, measured at a collector current of ICsat and the
recommended base current.
c) Selection procedures
Some simple calculations can be made to establish the
devicerequirements. Thefirstrequirementtobemetisthat
the peak voltage and current values are within the
capabilities of the device. For a flyback converter the peak
voltageandcurrentvaluesexperiencedbythepowerswitch
are given by the equations in Table 3.
Peak voltage across
the device
Peak device current
Table 3. Peak Voltage and Current in a Flyback
Converter.
where:
Vs(max) = maximum dc link voltage
σ= voltage overshoot due to transformer leakage
Vs(min) = minimum dc link voltage
Pth = throughput power of SMPS
δm= maximum duty cycle of SMPS
Note: in this example, the throughput power is equal to the
input power less the circuit losses up to the power switch.
MOSFET or bipolar?
The main factors influencing this decision are ease of drive
and cost, given the limitation on percentage of throughput
power which can be dissipated in the power switch.
MOSFETsrequirelower driveenergyand lesscomplicated
drive circuitry. They also have negligible switching losses
below 50 kHz. However, large chip sizes are required in
order to keepon state losses low (especiallyas breakdown
voltage is increased). Thus the larger chip size of the
MOSFET is traded off against its capacity for cheaper and
easier drive circuitry and higher switching frequencies.
For 110/120V mains driven TV power supplies the 400V
MOSFET dominates. At 220/240V there is a split between
bipolar and power MOSFET
Which MOSFET?
The optimum MOSFET for a given circuit can be chosen
on the basis that the device dissipation must not exceed a
certain percentage of throughput power. Using this as a
selection criterion, and assuming negligible switching
losses, the maximum throughput power which a given
MOSFET is capable of switching is calculated using the
equation;
where:
Pth(max) = maximum throughput power
δmax = maximum duty cycle
τ= required transistor loss
(expressed as a fraction of the output power)
Rds(125C) =R
DS(ON) at 125˚C
Vs(min) = minimum dc link voltage
A transistor loss of 5% of output power gives a good
compromise between device cost, circuit efficiency and
heatsink size (ie τ= 0.05)
Note that the RDS(ON) value to be used in the calculation is
at 125˚C (a practical value for junction temperature during
normal running). The RDS(ON) specified in the device data
is measured at 25˚C. As junction temperature is increased
the RDS(ON) increases, increasing the on state losses of the
MOSFET. The extent of the increase depends on the
device voltage, see Fig. 1.
For 400V MOSFET’s Rds(125C) = 1.98 x RDS(ON) @ 25˚C,
where RDS(ON) @ 25˚C is the value given in device data.
For 800V MOSFET’s Rds(125C) = 2.11 x RDS(ON) @ 25˚C.
Pth(max)=3×τ×Vs(min)
2×δmax
4×Rds(125C)
(2×Vs(max))+σ
2×Pth
δm×Vs(min)
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Televisions and Monitors Power Semiconductor Applications
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Fig. 1. Change in RDS(ON) vs. VDSS.
Which bipolar?
For maximum utilisation of a bipolar transistor it should be
runat its data ICsat. Thisgives agood compromisebetween
cost, drive requirements and switching losses. Using this
as a selection criterion the maximum output power which
a given bipolar transistor is capable of switching is
calculated using the equation;
where: Pth(max) = maximum throughput power
δmax = maximum duty cycle
Vs(min) = minimum dc link voltage
ICsat =I
Csat in transistor data
d) Selection table
Using the selection procedures just discussed, and the
device data given previously, the following selection table
of suitable devices for flyback converters of various output
powers has been constructed.
Output 110/120V (ac) 220/240V (ac)
Power mains mains
50 W BUK454-400B BUK454-600B
BUK454-800A
BUX85
100 W BUK455-400B BUK455-600B
BUK456-800A
BUT11A/BU506
150 W BUK457-400B BUK457-600B
BUK438-800B
BUT12A/BU506
200 W BUK457-400B BUK438-800A
BUW13A/BU508A
Table 4. Power Switch Selection Table
HVT’s for TV & Monitor Horizontal
Deflection
This application is one of the few remaining applications
which is entirely serviced by bipolar devices. The
technology is not yet commercially available to provide
MOSFET or IGBT devices for this application. A horizontal
deflection transistor is required for each TV and monitor
employing a standard cathode ray tube display.
The deflection transistor is required to conduct a current
ramp as the electron beam sweeps across the screen and
then withstand a high voltage peak as the beam flies back
before the next scan starts. The peak current and voltage
in the application define the device required. In addition to
this, the deflection transistor is required to switch between
the peak current and peak voltage states as quickly and
efficiently aspossible. In this application the switchingand
dissipation requirements are equally as important as the
voltage and current requirements.
Standard TV switches at a frequency of 16 kHz, rising to
32 kHz for improved definition TV (IDTV). In the future,
high definition TV (HDTV) will switch between 48 and
64 kHz.
Standard VGA monitors switch at 31 kHz, rising to 48 kHz
for SVGA. However, many other (as yet unnamed) modes
exist for PC monitors and work stations, extending up to
100 kHz switching frequencies.
Verticaldeflection is muchlower in frequency (50to 70 Hz)
and will not be discussed as this uses lower power devices
(typically 150V / 0.5A).
a) Voltage and Current Requirements
For a given scan frequency the voltage and current
requirements of the horizontal deflection transistor are not
fixed. However,thesuitabletransistors arealllinked bythe
relationship;
The derivation of this law is as follows:
The horizontal deflection angle (typically 110o) covered in
a given time is proportional to the magnetic field sweep
between the horizontal deflection coils. This is in turn
proportional to the product of the number of turns on the
deflection coils and the peak to peak current. The average
current in the deflection coils is zero and hence the peak
positive current in the coils is half the peak to peak current.
These relationships yield the following equation;
0 200 400 600 800 1,000 1,200
100, 1.74
200, 1.91
400, 1.98
500, 2.01
800, 2.11 1,000, 2.15
1.7
1.8
1.9
2
2.1
2.2
Device voltage rating (Volts)
Rdson(125C)/Rdson(25C)
Pth(max)=δ
max ×Vs(min)×ICsat
2
ICsat ×VCESM =constant
B∝n×I
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where:
B = magnetic field sweep between the horizontal
deflection coils
n = number of turns on the horizontal deflection coils
I = peak positive current in the horizontal deflection coils
The inductance of the horizontal deflection coils, L, is
proportional to the square of the number of turns, ie
Combining these two equations gives
and so for a given deflection angle and horizontal scan
frequency, and therefore a given B, L x I2is a constant.
For a given deflection frequency the flyback time is also
fixed. Flyback time is related to the deflection coil
inductance, L, and the flyback capacitance, C, by the
equation
During the flyback period the energy in the deflection coils
(1/2.LI2) is transferred to the flyback capacitor and so the
voltage across the flyback capacitor rises. Assuming all
the energy is conserved during this transfer, the increase
in voltage across the flyback capacitor, δV, is given by
So, if LI2is a constant then CδV2is a constant also.
Therefore, as LC is a constant so is (IδV)2. So we have:
δV is the voltage rise across the flyback capacitor due to
the energy transferred from the deflection coils during the
flyback period. The peak voltage across the flyback
capacitor, Vpeak, is given by
where: VCC = line voltage (typically +150 V)
The flyback capacitor is positioned across the collector
emitter of the horizontal deflection transistor. Therefore,
the peak voltage across the flyback capacitor is also the
peak voltage across the collector - emitter of the deflection
transistor.
Inordertoprotectthetransistoragainstoverloadconditions
(eg picture tube flash) a good design practice is to allow
VCEpeak tobe80%oftheVCESM rating. VCC isgenerallyaround
10%oftheVCEpeak (inordertoobtainthecorrectratioofscan
time to flyback time). This gives
All the positive current in the horizontal deflection coils is
conductedbythehorizontaldeflectiontransistor. However,
this is not the peak current in the transistor. The transistor
is normally also required to conduct the current in the
primaryof the line output transformer (LOT). Typically, this
will increase the peak current in the deflection transistor by
40%. Foroptimumdeflectioncircuitdesignthepeakcurrent
in the transistor will be its ICsat rating, ie
Therefore, for a given deflection angle and a given
horizontal scan frequency the horizontal deflection circuit
can be designed around any one of a number of devices.
However,the suitabledevicesare alllinked bythe equation
Summary
For a given horizontal deflection angle and horizontal scan
frequency
where:
VCESM = maximum voltage rating of the horizontal
deflection transistor
ICsat =I
Csat rating of the horizontal deflection transistor
δV = voltage rise on the flyback capacitor due to the
energytransferfromthehorizontaldeflectioncoils
I = peak positive current in the horizontal deflection
coils
L = inductance of the horizontal deflection coils
C = value of flyback capacitance
These relationships apply only for the assumptions
declared previously.
b) Switching and Dissipation
Requirements
In TV, for a given scan frequency the minimum on time of
the transistor is well defined. For 16 kHz systems the
transistor on time is not less than 26 µs and for 32 kHz
systemsit isnot less than13 µs. This enables the required
storagetime ofthetransistortobe welldefined. For16 kHz
systems a maximum storage time of 6.5 µs is the typical
requirement. For 32 kHz systems the required maximum
storagetime is typically 4.0 µs. For higher frequencies the
required maximum storage time is reduced still further.
L∝n2ICsat =1.4×I
B2∝L×I2
ICsat ×VCESM =constant
tfb∝√
L×CVCESM ≥1.25×δV+190
ICsat =1.4×I
ICsat ×VCESM =constant
L×I2=C×δV2=constant
1
2×L×I2=1
2×C×δV2
δV×I=constant
Vpeak =δV+VCC
VCESM ≥1.25×δV+190
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In monitorapplications, especially multi frequency models,
the on time is not well defined. There are many different
frequency modes and several control ic’s giving different
duty cycles. However, it can be said that the higher the
frequency, the shorter the storage time required.
Storagetimeinthecircuitcanalwaysbereducedbyturning
the transistor off harder. However, this eventually leads to
a collector current tail at turn off and as a consequence the
turnoffdissipation increases. Turnoff dissipationaccounts
for the bulk of the losses in a deflection transistor and it is
crucial that this is kept to a minimum. The deflection
transistor must be tolerant to drive and load variations if it
is to achieve a low turn off dissipation because the east -
west correction onlarger screen television sets means that
circuit conditions are not constant. Turn off can be
optimised during the design phase by ensuring that the
peak reverse base current is roughly half of the peak
collector current and the negative base drive voltage is
between 2 and 5V.
Turn on performance is not a critical issue in deflection
circuits. At turn on of the deflection transistor the ICis low,
the VCE is low and, therefore, the dissipation is low. The
actualturnonperformanceofthetransistorhasa negligible
effect.
c) HVT’s for Horizontal Deflection
The deflection circuit must satisfy any specified cost,
efficiency and EMC requirements before it can be called
acceptable. A very highvoltage deflection transistorwould
allow a lower deflection coil current to be used, reducing
the level of EM radiation from the deflection coils, but it
would require a higher line voltage and it would also result
in higher switching losses in the transistor. A very high
deflectioncoilcurrentwouldallowalowervoltagedeflection
transistor to be used and a lower line voltage. This would
alsoyieldlowerswitchinglosses inthe deflectiontransistor.
However, high currents in the deflection coils could lead to
EMC problems, and the need to keep the resistive coil
losses low would mean that thicker wire would have to be
used for the windings. Above a certain point the skin depth
effect makes it necessary to use litz wire.
For 16 kHz and 32 kHz applications the 1500V bipolar
transistor has become the designers first choice, although
many 16 kHz systems could work well using 1000V
devices. However, concern over fault conditions that can
cause odd high voltage pulses has seen 1500V adopted
as the ‘standard’. The collector currents involved range
from2.5Apeak to8Apeak forTVand3.5A peakto 7Apeak
formonitors. Thetransistorsforthese applicationsarenow
considered.
16 kHz applications
Table 5lists the1500V transistors for16 kHz TV deflection
systems and a summary of their main characteristics.
Part VCESM ICsat Application
Number
BU505/D 1500V 2A Monochrome sets
BU506/D 1500V 3A 90˚ Colour; ≤ 23"
BU2506DF
BU508A/D 1500V 4.5A 110˚ Colour; 21-25"
BU2508A/D
BU2520A/D 1500V 6A 110˚ Colour; 25-29"
Table 5. Transistors for 16 kHz TV deflection
Alloftheabovetypesareavailableinbothnon-isolatedand
isolated outlines (F-pack), except the BU2506DF - F-pack
only. Isolated outlines remove the need for an insulating
spacer to be used between device and heatsink. Devices
are available both with and without a damper diode (eg
without: BU505, BU2520A and with: BU505D, BU2520D).
The BU2506DF is only available with a damper diode.
The BU25XX family is a recent addition to the range of
Philips deflection transistors. Far from being just another
1500V transistor, the BU25XX has been specifically
designed for horizontal deflection. By targeting the device
for this very specialised application it has been possible to
achieve a dissipation performance in deflection circuits
which is exceptional.
TheBU2520A usesthe superiortechnologyofthe BU25XX
family applied to a large chip area. The BU2508A has an
hFE of 5 at 5V VCE and 4A IC. The BU2520A has an hFE of
5at5VV
CE and 6A IC. This gives designers working on
large colour television sets a high hFE deflection transistor
with a high current capability. The high hFE reduces the
forward base drive energy requirements. The high current
capability enables the energy drawn from the line output
transformer to be increased. Using a BU2520A device
allows the EHT energy to be increased for brighter pictures
(a feature of new ‘black line’ tubes) without having to
increase the forward base drive energy to the deflection
transistor.
32 kHz applications
Table 6 gives the 1500V transistors for 32 kHz deflection
systems and a summary of their main characteristics.
Part VCESM ICsat Application
Number
BU2520A 1500V 6A 110˚ Colour; ≤ 28"
BU2525A 1500V 8A 110˚ Colour; ≤ 32"
Table 6. Transistors for 32 kHz deflection
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For the foreseeable future 32 kHz TV will be concentrated
at the large screen sector ( ≥ 25"). These TV’s will employ
diode modulator circuits lessening the need for D-type
transistors. With a switching frequency twice that of
conventional TV the dissipation in these devices will be
higher. Forthisreasonthenon-isolatedversions,withlower
thermal resistance, will be prevalent in these applications.
Monitor applications
The applications given in Table 7 should be seen as an
indication of the limits that successful designs have been
achieved for that device type. This should help in the
selection of a device for a given application at the design
concept stage. For example, a 15" monitor requiring
operation up to 6A at 64 kHz could use eithera BU2522A,
a BU2525A or a BU2527A. If the design has specific
constraintson switching and dissipation then theBU2525A
and BU2527A would be better. If, as well, a guaranteed
RBSOA is required then the BU2527A is the best choice.
Table 7 gives the 1500V transistors for monitor deflection
systems, concentrating on the common pc and industry
standard work station modes.
Part VCESM ICsat Application
Number
BU2508A/D 1500V 4.5A 14", SVGA, 38 kHz
BU2520A 1500V 6A 15", SVGA, 48 kHz
BU2522A 1500V 6A 15", 64 kHz
BU2525A 1500V 8A (17", 64 kHz)
BU2527A 1500V 6A 17", 64 kHz
Table 7. Transistors for Monitor deflection
All devices are available in non-isolated and isolated
outlines. The excellent dissipation of this range of devices
meansthat,evenatmonitorswitchingfrequencies,devices
in an isolated package can be used
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4.1.4 TV and Monitor Damper Diodes
Introduction
Philips Semiconductors supply a complete range of diodes
for the horizontal deflection stage of all volume TV and
monitor applications. This note describes the range of
Philips parts for the damper (also called efficiency) diode
in horizontal deflection. The damper diode has some
unusual application specific requirements that are
explained in this section.
Damper diodes form an essential part of the horizontal
deflection circuit. The choice of diode has an effect on the
total circuit dissipation and the display integrity. A poor
selection can lead to unnecessary power loss and a visible
picture distortion.
As well as a full range of discrete devices for the damper
diode application Philips offfer a range of horizontal
deflection transistors with integrated damper diodes.
These devices offer a cost and space saving, especially
beneficial for high volume TV production.
Discrete Damper Diode Selection Guide
IFWM,I
F(AV). The quoted IF(AV) values do not correspond to
any particular current in the application. The values are
standard data format for selection purposes and
comparisonwithcompetitortypes.Ingeneral,thelargerthe
IF(AV) the higher the deflection coil current and/or frequency
in the application. A more meaningful specification is IFWM,
this refers to the peak operating current in a 16 kHz TV
application given a standard current characteristic. The
applicationcolumns in table 1define thelimit fitness for use
of each diode.
VRSM.The damper diode should have a voltage capability
equal to the deflection transistor. In most applications this
will be 1500V. The VRSM value equates to the peak flyback
voltage. The diode data should not be viewed as that for
other diodes where it is quite common to use devices with
VRSM 5 or 10 times greater than the peak circuit voltage.
Damper diodes will operate in horizontal deflection circuits
with peak flyback voltages up to the specified limit.
However, the limit VRSM should not be exceeded in any
circumstance. In practice, a device with VRSM of 1500V will
befoundinapplicationswithpeakflybackvoltagesof1300V
in normal running; fault conditions do notusually see more
than a 200V rise in flyback voltage.
Outline. The Philips range spans the available outlines for
this application from axial to TO220 type. The SOD57 and
SOD64 are hermetically sealed axial - leaded glass
envelopes. These outlines combine the ability to house
large chips with proven reliability and low cost. For high
ambient temperatures with severe switching requirements
the addtion of cooling fins may be necessary to achieve
successful operation at the application limit.
For higher currents and frequencies there are devices in
TO220 type outlines. TO220AC is a two-legged
non-isolated outline. The pin-out is such that the tab is
alwaysthecathode.Foranisolatedequivalentoutlinethere
are SOD100 and the newer SOD113. The SOD100 is the
traditional isolated TO220 outline allowing the device to be
attached to a common heatsink without any separate
isolation. The SOD113 is an enhanced version of SOD100
offering an improved isolation specification. Philips offer a
complete range of mounting accessories for all these
outlines.
Specification Application
Device Type IFWM,IF(AV) VRSM Outline TV Monitor
BY448 4 A 1650 V SOD57 ≤ 21", 16 kHz -
BY228 5 A 1650 V SOD64 25", 16 kHz -
BY328 6 A 1500 V SOD64 28", 16 kHz 14", SVGA, 38 kHz
BY428 4 A 1500 V SOD64 21", 32 kHz 14", 64 kHz
BY359 10 A 1500 V TO220AC 36", 32 kHz 17", 64 kHz
BY359F(X) SOD100 (SOD113)
BY459 10 A 1500 V TO220AC HDTV 19", 1280x1024, 82 kHz
BY459F SOD100
Table.1 Philips Semiconductors Damper Diode Selection Guide
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Horizontal Deflection Transistors with
Integrated Damper Diode
Fig.1. BU508A/2508A vs. BU508D vs. BU2508D
The range of devices available covers most high volume
TV&Monitorapplicationswheredesignersrequireachoice
of devices to meet their requirements. The differences are
shown in Fig. 1 above. These devices are all monolithic
structures. The process of integrating the diode does not
reduce the performance of the deflection transistor.
For traditional horizontal deflection circuits with a single
damper diode it is easy to see the benefits of integrating
the deflection transistor and damper diode. The additional
dissipation in the integrated damper diode should be taken
into account in the thermal management considerations.
The use of the deflection transistor with integrated diode
allows a simpler layout with lower component count and
cost.
For circuit designs that employ a diode modulator circuit it
is still quite common to employ a deflection transistor with
an integrated damper diode. In these circuits the current
is shared between the integrated diode and the discrete
modulator damper diode. This technique allows smaller
discretediodestobeusedorreducedthermalmanagement
for the discrete device. For example, this could allow the
circuitdesignertoremoveanycoolingfinsonanaxialdiode;
orreplaceaTO220typewithacheaperaxialtypeofdiscrete
diode in the modulator.
Table2belowshowsaselectionofPhilipsSemiconductors’
horizontal deflection transistors with an integrated damper
diode.
ICsat.Thisvalueisanindicationof thepeakcollector current
in a 16 kHz TV horizontal deflection circuit for which
optimum dissipation and switching can be obtained. For
the diode the ICsat value should also be taken as the peak
current (ignoring any instantaneous spikes at the start of
scan). For higher frequency applications in monitors the
ICsat value reduces slightly.
VCESM.The voltage capability of the deflection transistor
and damper diode are the same. As for discrete devices,
there is no need for excessive insets. The VCESM value
equates to the peakflyback voltageand, asfor the VRSM of
a discrete damper diode, should not be exceeded under
any circumstance.
Outlines. Devices are available in three different outlines,
one non-isolated (SOT93) and two isolated/full-pack
designs (SOT199, TOP3D). The outline is defined by the
last letter in the type number, for example:
BU2508D SOT93 non-isolated
BU2508DF SOT199 isolated
BU2508DX TOP3D isolated
All three outlines are high quality packages manufactured
to Philips Total Quality Management standards.
Rbe
BU508A, BU2508A
BU2508D
BU508D
Rbe
BE resistor, damper diode
Specification Application
Device Type ICsat VCESM Outline TV Monitor
BU2506DF 3.5 A 1500 V SOT199 21", 16 kHz -
BU2506DX TOP3D (SOT399)
BU508D 4.5 A 1500 V SOT93 21-25", 16 kHz -
BU508DF SOT199
BU2508D SOT93
BU2508DF 4.5 A 1500 V SOT199 21-25", 16 kHz 14", 38 kHz, VGA
BU2508DX TOP3D (SOT399)
BU2520D SOT93
BU2520DF 6 A 1500 V SOT199 25-29" 16 kHz 14", 48 kHz, SVGA
BU2520DX TOP3D (SOT399)
Table 2 Philips Semiconductors Deflection Transistors with Integrated Damper Diode Selection Guide
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Operating Cycle
The waveforms in Fig. 2 below show the deflection coil
current and flyback voltage in a simplified horizontal
deflection circuit. In Fig. 2 the damper diode current is
highlighted. All the flyback voltage is applied across the
damper diode. These waveforms are valid for both the
traditional type of deflection circuit (Fig. 3) and the diode
modualtor deflection circuit (Fig. 4).
Fig. 2. Horizontal Deflection Damper Diode
Operating Cycle.
During flyback the energy in the deflection coil, Lcis
transferred to the flyback capacitor, Cfb. With the transfer
of energy the voltage on Cfb, hence the voltage across the
diode, rises sinusoidally until all the energy is transferred.
Now the current in Lcis zero and the diode and Cfb are at
the peak flyback voltage. The energy now transfers back
to Lc. As the energy is transferred the voltage decreases
until all the energy is back in Lcwhen there is no voltage
across Cfb, and maximum current through Lc. If there was
no diode present, this operation would continue with the
energy transferring back to Cfb with the voltage continuing
to decrease until all the energy in Lchad been transferred;
the peak voltagenow reversed. But with the damperdiode
in place across Cfb (see Figs. 3 & 4), as the voltage falls
negative the diode will be forward biased and tend to
conduct.
Consider now the application requirement which is to
establish a peak negative current in Lc before the start of
the next scan. As the decreasing voltage on Cfb tends to
zero sothe currentin Lcreaches apeak negative valueand
the next scan can start. The transfer of energy into the
capacitor has to be stopped during the scan, hence the
addition of the damper diode.
Most TV & monitor display circuits will employ an element
of over-scanning; this means at the start of diode
conduction the beam will be off-screen. Over-scanning is
introduced to reduce the effect of any spurious switching
characteristics as the diode switches.
Power Dissipation
There are two significant factors contributing to power
dissipation in a damper diode: forward recovery and
on-state forward bias. Reverse recovery and reverse bias
losses are negligible in this application. As a general rule,
thetotaldissipationishalfforwardrecoveryandhalfforward
bias. To explain this further we have to consider the
operating cycle of the diode in detail.
scan
Idiode
ILc=Idiode
Idiode
ILc=Ic
flyback flyback
flyback
voltage
flyback
voltage
deflection
coil current
Fig. 3. Traditional Deflection Circuit. Fig. 4. Diode Modulator Circuit Example.
Lc
Cfb
Cs
Lsat
Vcc
E.H.T.
Line Output
Transformer Horizontal
Deflection
Coil
Smoothing
Capacitor
Damper
Diode
+
-
Deflection
Transistor
Linearity Coil
Lc
Cfb
Cs
Lsat
Vcc
E.H.T.
Line Output
Transformer Horizontal
Deflection
Coil
Smoothing
Capacitor
Damper
Diode
+
-Deflection
Transistor
Linearity Coil
Cmod
Lmod
Vmod
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Forward Recovery. As the voltage goes negative the
electricfield builds up acrossthe diode. The device design
and process technology determine the point at which
conduction starts. At the start of conduction the voltage is
a maximum: the forward recovery voltage, Vfr. A detailed
view of the damper diode voltage and current at the start
of diodeconduction is given in Fig. 5 below. As the current
flows the voltage acrossthe diode drops to its steady-state
VFvalue; the time this takes is called the forward recovery
time, tfr.
Fig.5. Damper Diode V & I Waveforms.
The values for Vfr and tfr are application dependent. In
general, Vfr is ≤ 20V; tfr is ≤ 500ns for VFto fall to 2V; and
the rate of rise in diode current, dIF/dt, will be between 25
- 90 A/µs. A ‘good’ damper diode will not only have low Vfr
and low tfr but as a result, it will also allow the current to
switch to the diode faster, giving a higher dIF/dt.
Forward Bias. As the beam scans from the left towards
the centre the voltage drop across the diode is determined
by the device VFcharacteristic for a given Lccurrent. As
the beam scans from left to right the diode current
decreases.
Measurement of the VFis not possible in the application.
The best indication of thelosses comes from themaximum
hot VFinformation contained in the datasheets.
Reverse Recovery& Reverse Bias. To theright of centre
screenLccurrent becomes positive and, hence, current no
longerflowsthroughthedamperdiode. Inreverserecovery
the damper diode does not experience any high current -
high voltage characteristic that would be a cause of
significant dissipation.
Duringflybackthediodeisreversebiased,anotherpossible
cause of dissipation. The combined effects of reverse
recovery and reverse bias are negligible in comparison to
forward recovery and forward bias.
Picture Distortion
The diode does not start to conduct until the forward
recovery voltage is approached: a device with a high
forward recovery voltage, Vfr, will take longer to start
conduction than a device with a low Vfr. A delay in the start
of diode conduction means that the deflection coil current
is dominated by the flyback capacitor, Cfb at the start of the
scan. This can cause a visible distortion to the left-hand
side of the display.
Thevoltage acrossthe diodemodulatesthe voltageacross
the coil. For a device with a long forward recovery time, tfr,
the diode forward recovery characteristics will affect the
voltage across the deflection coil at the start of the scan.
This can also cause a visible distortion to the left-hand side
of the display.
For display circuit optimisation it is essential that the
requirements for the damper diode are understood and
taken into account in device selection.
time
time
Vfr
VF
IF dIF
dt
tfr
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TV Deflection Circuit Examples
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4.2.1 Application Information for the 16 kHz Black Line
Picture Tubes
With the introduction of the black line picture tubes new
drive circuits are required. To have full benefits, the EHT
voltageand beam current mustbe increased. This section
describes the horizontal deflection and EHT generation.
Some hints for vertical deflection and video amplifiers are
given as well.
Summary
This section describes the horizontal deflection circuitry of
the black line picture tube A66EAK022X11 and
A59EAK022X11. To take full advantage of this new tube
it must be driven at 27.5kV @ 1.3mA. This implies that in
an ordinary combined EHT and deflection stage in the no
load condition, zero beam current, the high voltage
increases toabout 29.5kV. The mainchange to thiscircuit,
compared with existing circuits, is the line output
transformer (AT2077/34) uses four layer DSB technology.
For the vertical deflection a minor modification on PCALE
report ETV8831 is given. The video output stage suited to
this tube is described in PCALE report ETV8811.
1. Introduction
One step in improving picture quality is the introduction of
the black line picture tube. With this tube day-time TV
viewing with a bright high contrast picture becomes
possible. To achieve this the picture tube is provided with
a dark screen and increased EHT power capability by
means of an invar shadow mask.
When the 45AX black line picture tube is compared with
the existing 45AX tubes the following modifications in the
application must be made.
-Increase of the EHT power to 27.5kV @ 1.8mA beam
current.
-Increase of the cut off voltage to 160V.
TocopewiththishighEHTpowerdemand,anewlineoutput
transformer has been developed (AT2077/34). The main
part of this section is dedicated to the horizontal deflection.
For supply, vertical deflection and video amplifier details
reference will be made to separate reports.
Special care is taken to suppress certain geometrical
picture distortions which otherwise would become
noticeable at the increased levels of dynamic EHT load
variations. These distortions are the result of oscillations
in the line output stage.
All measurements and circuitsare based and tested onthe
66FS picture tube. Since the 59FS tube is electrically
identical to the 66FS tube this circuit is also suited for the
59FS picture tube. With some minor circuit modifications
(component values) this circuit is also suited for 33" picture
tubes.
2. Circuit Description
The horizontal deflection board is built up of four major
parts, see Fig. 1.
Fig. 1. Block Diagram of Horizontal Deflection Board.
In high end TV sets a lot of low voltage supply current is
required. In this design a separate transformer in parallel
to the line outputtransformer is foreseen for the generation
of these auxiliary supplies. In applications where this
auxiliary power is not required this part can be omitted by
simply leaving out the additional transformer. The other
parts of this horizontal deflection are more or less classic.
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2.1 Drive Circuit
Fig. 2. Drive Circuit
The horizontal drive circuit is a classical transformer
coupled inverting driver stage. When driver transistor T1
is conducting energy is stored in the transformer. When
T1 is turned off the magnetising current continues to flow
in the secondaryside of the transformer thus turning on the
deflection transistor. At this time the voltage on the
secondary side of the driver transformer is positive
(VBE+IBxR6). When the driver transistor turns on again this
secondary voltage reverses and will start to turn off the
deflection transistor. At the same time energy is stored in
the transformer again.
During this turn off action the forward base drive current
decreases with a controlled dI/dt, thereby removing the
stored charge from the deflection transistor. The dI/dt
depends on the negative secondary voltage and the
leakage inductance. As a rule of thumb, the deflection
transistor stops conducting when its negative base current
is about half the collector peak current.
To prevent the deflection transistor from turning on during
flybackduetoparasitic ringingon thesecondaryside ofthe
drivertransformeradamperresistorisconnectedinparallel
with the base emitter junction of the deflection transistor.
Also at the primary side of the driver transformer a damper
network is added (R5 & C10) to limit the peak voltage on
the driver transistor.
D5 is added for those applications where in the standby
mode the deflection stage is turned off by means of
continuous conduction of the driver transistor. The
explanation is as follows:
When T1 is suddenly made to conduct continuously, a low
frequency oscillation willoccur in C9 and the primary of L3.
As soon as the voltage at pin 4 of L3 becomes negative T2
starts conducting until the driver transformer is
demagnetised. This will cause an extremely high collector
current surge. D5 prevents pin 4 of L3 going negative and
so this fault condition is avoided. For those applications
where this condition cannot occur, D5 can be omitted.
2.2 Deflection Circuit
The horizontal deflection stage contains the diode
modulator which not only provides east-west raster
correction but also inner pincushion correction, picture
width adjustment and EHT compensation. It is not easy to
achieve optimum scan linearity over the whole screen.
Either the linearity inside the PAL test circle is good and
outside the circle the performance is poor, or the average
performance over the whole screen is good but inside the
test circle deviation is visible. In this application the
S-correction capacitors C15 and C16 are balanced in such
a way that a good compromise for the scan linearity is
achieved.
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Fig. 3. Deflection Circuit
As already stated in the introduction, due to high beam
current in combined EHT and deflection stages picture
distortions will occur.
One of these effects is the so called cross-hatch "noses",
visible as horizontal phase ringing just below each
horizontalwhiteline. DuringabrighthorizontallinetheEHT
atthepicturetubedecreases. Inthenextflybackthepicture
tube capacitor is recharged by the line output transformer.
Thisenergy is taken from theS-correction capacitor, which
mustberechargedvia theprimarywindingof thelineoutput
transformer. This action hasa resonance of a few kHz and
thus oscillation is visible at the screen.
To avoid this a dip rectifier is connected in parallel with the
S-correction capacitor (D9, C14, R8). The energy taken
from the S-correction capacitor can now be recharged by
C14.
Another geometry distortionis the "Krückstockeffekt". Due
to trapezium correction the EW-drive signal applied to L6
can be discontinuous. This will cause amplitude ringing at
the topof the screen. An effective way to damp this ringing
is a resistor in series with the EW-injection coil.
Theconsequenceoftheabovementioned measuresisthat
the drive reserve of the diode modulator has decreased.
Tocompensateforthis aseparatewindingof thelineoutput
transformer is connected in series with the deflection coils.
2.3 EHT Generation
For an optimum performance the black line picture tube
must be driven at an increased EHT of 27.5kV @ 1.3mAav.
This implies that the EHT in the no load condition, zero
beam current, will be about 29.5kV. To generate this
increasedEHT a newlydesigned lineoutput transformer is
used with a 4-layer diode split EHT section. From an
integrated potentiometer the adjustable focus and grid 2
voltages are taken.
Also the frame supply voltage (26V) and video supply
voltage (180V) are taken from the line output transformer.
Theauxiliarywindings ofthistransformercanbeconnected
rather freely so that a diverse range of auxiliary supplies
canbe obtained. The only restriction is that the RMSvalue
of the current in a given winding may not exceed 2A.
Furthermore the supply current for the heater is obtained
from this transformer just like the Φ2 feedback pulse
(positive) anda (negative) pulse to synchronise the SMPS.
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2.4 Auxiliary Supply
Fig. 4 Auxiliary Supply
Whenmore auxiliarypower than can behandled by theline
output transformer is required, an auxiliary supply
transformer,asshowninFig. 4, isa goodalternative. Such
an extra transformer in parallel with the primary winding of
the line output transformer is an efficient way to generate
low voltage high current supplies. As the transformer is
optimised for this purpose no additional stabilisation is
required.
Due to the high inductance of the primary winding no
influence on the collector current is noticeable. Output
voltagesareveryclosetothe targetvalues(fineadjustment
with primary taps) and have low Ri (HT line is stabilised).
2.5 Additional Circuit Information
12V supply:
The SMPS used in this concept delivers 16V unstabilised.
This needs to be regulated to supply the sync processing
IC which operates at 12V. This regulation can be done on
thesyncprocessingboardorthehorizontaldeflectionboard
since this also acts as a power distribution board.
Tuning voltage:
The tuning voltage is created simply by means of a series
resistor R1and a 30Vreference diode located atthe tuning
board.
EHT compensation:
For proper picture performance it is essential that EHT
information is available to compensate picture width and
height for EHT variations. For this reason the aquadag is
connected to the foot point of the line output transformer.
This point is connected to ground by C18 and to the 26V
byanonlinearresistornetwork(R12,D11,R13,R14). This
network is designed in such a way that it matches with the
non linear impedance of the line output transformer and
C18 matches with the picture tube capacitance. Thus the
voltage available at the foot point of the line output
transformer is a good representation of the EHT variation.
This EHT information is sent to the geometry processor
TDA8433. This information can also be used for beam
current limiting. It must then be fed to the video processor
for contrast/brightness reduction.
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Fig. 5. Circuit Diagram (continued on next page)
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Fig. 6 Circuit Diagram (continued from previous page)
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3. Oscillograms
Oscillogram 1:
In this oscillogram the lower trace is the voltage across the
deflection transistor (200V/div). The middle trace is the
current in the horizontal deflection coil (1A/div). The upper
trace is the collector current in the deflection transistor
(2A/div). The time base is 10µs/div.
Remarks:
At the end of flyback there is a negative overshoot at the
collectorvoltage. Thisiscausedbytherelativeslowforward
recovery of the damper diode. A part of this current is
reverseconducted by thedeflection transistor. About 12µs
after the start of the scan the deflection transistor is turned
on and starts reverse conducting and takes over a part of
the current in the damper diodes. See also oscillogram 3.
Oscillogram 2:
The upper trace is the voltage across the deflection
transistor (200V/div). The lower two traces are the
minimum and maximum voltage in the diode modulator
(cathode D8) with nominal EW and amplitude settings
(50V/div). The time base is 10µs/div.
Oscillogram 3:
In the upper trace the current in the upper diode is shown
(1A/div). The middle trace is the current in the lower diode
(1A/div). The time base is 10µs/div.
Remarks:
12 µs after the start of the scan the deflection transistor is
turned on. Current in the diode modulator is then taken
over by the deflection transistor. See also oscillogram 1.
Oscillogram 4:
In the upper trace the collector voltage of the driver
transistorisshown(100V/div). Themiddletraceisthebase
drivecurrent of thedeflection transistor (1A/div). The lower
trace is the base emitter voltage of the deflection transistor
(5V/div). The time base is 10µs/div.
Remarks:
Theovershootatthecollectorvoltageofthedrivertransistor
is damped by R5 and C10. The current spike in the base
drive current (marked with *) is the reverse conduction of
the deflection transistor during the forward recovery of the
damper diode. See also oscillogram 1.
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4. Vertical Deflection, Synchronisation
and Geometry Control
The vertical deflection, synchronisation and geometry
control circuits are based on an existing PCALE report (ref
3). Becauseforthisapplicationanothertubeandlineoutput
transformer are used some minor modifications are
required (component values), see Fig. 7.
- Due to the increased deflection current, the vertical
feedback resistor must be decreased: one of the 2.2Ω
resistors becomes 1Ω.
- Due to the increased beam current, the EHT
compensation network at pin 24 of the TDA8433 needs to
be modified: 120kΩ → 100kΩ, 82kΩ → 150kΩ,
27kΩ → 33kΩ, 68 nF → 10 nF.
-For additional phaseshift pin 14of the TDA2579 isbiased
withacurrentfromanegativevoltagesource(rectified from
theΦ2feedbackpulse). Whilethisfeedbackpulseissmaller
than in the original circuit, the 240kΩresistor must be
increased to 1.2MΩ.
Orientation values for the TDA8433 register settings are:
reg hex
00 25
01 5A
02 07
03 2C
04 2B
05 16
06 16
07 15
08 22
09 21
0A 21
0B 0F
0F 04
Oscillogram 5:
Vertical deflection current (1A/div) and the output voltage
(pin 5) of TDA3654 (10V/div). The time base is 5ms/div.
Remarks:
The noise on the output voltage is cross talk from the line
deflection coils.
5. Video Amplifiers
The gun of this picture tube is a new design and has its
optimumperformanceatacutoffvoltage,VCO = 160V. This
implies thatthe video supply voltageshould be atleast 20V
higher, so Vvideo ≥ 180V. A video amplifier very well suited
for this purpose is the TDA6100.
The RMS voltage of the heater winding of the line output
transformer is V10 = 7.35VRMS, so a series resistor must be
used (3.9Ω; 400mW).
6. References
Information from this section was extracted from
"Application information for the 16 kHz black line picture
tubeA66EAK022X11and A59EAK022X11";ETV89010 by
Han Misdom.
1. "Some aspects of the diode modulator"; EDS7805 by
C.H.J. Bergmans.
2. "A synchronous 200W Switched Mode Power Supply
intended for 32kHz TV"; ETV89009 by Henk Simons.
3. "Deflection processor TDA8433 with I2C-bus control";
ETV8831 by D.J.A. Teuling
4. "Application of the TDA6100 video output stage";
ETV8811 by D.J.A. Teuling
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Fig. 7. Vertical Deflection, Synchronisation and Geometry Control
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4.2.2 32 kHz / 100 Hz Deflection Circuits for the 66FS Black Line
Picture Tube
This report contains a description of deflection circuitry
(horizontal32 kHz,vertical50-120 Hz)for the66FSpicture
tubeA66EAK22X42. Thisdesignisintendedfor flickerfree
TV applications. Provision is made to supply the power for
the frequency conversion box.
Summary
The 66FS picture tube is compatible but not identical with
the types of the 45AX range. To obtain the typical Black
line high contrast and high brightness, the beam current
and EHT must be increased at nominal operating
conditions. ThishigherEHTalso improvesthespotquality.
The deflection current is increased because of the higher
EHT and reduced sensitivity of the deflection unit.
In comparison with laboratory report ETV8713, describing
deflection circuits for 45AX, most modifications are found
in the horizontal deflection stage. To generate the
increased EHT power a new line output transformer (LOT)
with a four layer EHT coil is used. To handle the higher
deflection currents two transistors are used in parallel and
also two flyback capacitors are used. We have also taken
the opportunity to introduce the TDA8433. This deflection
processor -in BiMos technology- is the successor of the
TDA8432.
The vertical deflection stage is redesigned in such a way
that vertical shift signals can be inserted without bouncing
effects. The insertion of vertical shift signals is necessary
in 100 Hz operation for a proper interlace.
1. Introduction
In this report a description is givenof double line and frame
frequency(32 kHz; 100 Hz) deflection circuits forthe 66FS
picture tube A66EAK22X42. The report is based on report
ETV8906, describing these circuits for the 78FS picture
tube 1. By changing some component values the pcb for
the 78FS can also be applied to drive the picture tube
A66EAK22X42. In the line output stage the output
transistor BU2508 is used.
2. General description
2.1 Block diagram
The block diagram is given in Fig. 1. The main
interconnections are given as well. The separate blocks
can be recognised in the circuit diagram.
The separate H and V sync and V shift are available from
the frequency conversion box.
2.2 Circuit architecture
A key component in this set up is the deflection processor
TDA8433. Thehorizontalandverticalpicturegeometrycan
be controlled by means of I2C bus commands. Because
thisdeflectionprocessor hasno verticaloscillator,therewill
be no vertical deflection when there are no vertical sync
pulsesappliedtotheset. Forlaboratorypurposeaseparate
vertical oscillator is added to make the monitor part a self
containedunit. Whenincorporatedinareceiverthisvertical
oscillator can be omitted. When there are no vertical sync
pulses, the guard circuit of the vertical output stage will
blank the video information. This prevents spot burn-in of
the tube.
Fig. 1 Block diagram
HORIZONTAL
OSCILLATOR HORIZONTAL
DRIVER
AT4043/87
HORIZONTAL
DEFLECTION
& EHT
AT2077/33TDA2595
VERTICAL
OSCILLATOR DEFLECTION
PROCESSOR
TDA8433
EW DRIVER
AUXILIARY
SUPPLY
AT4042/32B
VERTICAL
DEFLECTION
& DC SHIFT
TDA 3654
H SYNC
V SYNC
I2C BUS V SHIFT
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The vertical deflection stage consists of the well known
TDA3654verticaloutput IC. Tomakeverticalshift insertion
possible, the output stage is slightly redesigned. This ac
coupled output stage now has a quasi dc coupled
behaviour.
The EW driver is a voltage amplifier, acting as a buffer
betweenthe deflection processorand thediode modulator.
The block "horizontal oscillator" consists of the TDA2595
with its Φ1 and Φ2 loop, the horizontal oscillator itself and
sandcastle generation. The other features of this ic are not
used.
Coupling between the TDA2595 and the deflection stage
is made by the horizontal driver stage. This stage is a
transformer coupled inverting driver stage.
The horizontal deflection stage is a classic concept. It
consists of a combined deflection and EHT generation. It
also comprises linearity correction, S-correction, inner pin
cushion correction and a dc shift circuit. The LOT
(AT2077/33) belongs to the transformer family DSB (diode
splitbox) and has four EHTlayers. It delivers the following
voltages:
* EHT = 27.5 kV @ 1.3mA
* Focus = 0.22 - 0.30 x EHT
*Vg
2= 0.011 - 0.033 x EHT
* Heater ≈10.4VRMS
* Video supply = 192V
* Frame supply = 28V
*Φ2 ref. pulse = +40Vpp
Furthermore the LOT has some taps which can be useful
when the application is modified.
In parallel to the primary winding of the LOT the auxiliary
supply transformer(AT4043/32B) is located. This auxiliary
supply delivers the following voltages:
* +5V @ 5A
* +15V @ 1A
* -12V @ 1A
These supply voltages are intended for the digital and
analog signal processing circuits.
The philosophy behind this circuit needs some further
explanation.
It is very difficult to generate exactly 5V at the output of an
SMPS or LOT. Due to an optimum winding design of this
kind of transformer, the voltage ratio per turn is high (2 - 5V
per turn). This implies that a stabilizer is required. A
switching post regulator adds to the circuit complexity and
cost. A dissipative series stabiliser needs at least 2V, so
for a 5A supply the losses are already 10W.
The auxiliary transformer used in this concept is optimised
for generating these low voltages at high currents. The
winding design is such that no stabilizer is required after
the rectifier. Due to the high primary inductance of this
transformer the collector current increase of the deflection
transistor is negligible.
If the auxiliary loads are low, this auxiliary supply
transformercan be omitted andthe unused tapsof the LOT
can be used to generate these voltages.
3. Circuit description
Using circuit diagram blocks the total concept will be
explained. This will be done with reference to the function
blocks of Fig. 1. The complete circuit diagram is given in
Figs. 11-13.
3.1 Vertical oscillator
As already stated in section 2.2 this vertical oscillator can
be omitted in a final design. The circuit is shown in Fig. 2.
This oscillator is the well known astable multivibrator built
uparoundT8andT9. Thisoscillatorisfreerunningat45 Hz
and can be synchronised up to at least 120 Hz. T7 is an
additional sync transistor and is ac coupled to the vertical
sync signal (TTL level).
Fig. 2 Vertical oscillator
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3.2 Deflection processor TDA8433
The TDA8433 is an analog, I2C bus controlled, deflection
processor. It generates the vertical deflection current
waveform and the EW (East-West) waveform. The
necessary corrections on these waveforms are I2C bus
controlled. This ic also includes some DACs and ADCs.
They can be used for control functions of other circuitry 2.
The resistor at pin 4 determines the reference current for
this ic. Pin 2 is the vertical sync input. At the capacitor at
pin 5 (C-flyback) a triangle waveform is generated which is
used for internal timing. This signal is used to generate the
vertical sawtooth at pin 22 (C-saw). At pin 23 (C-amp1) a
storagecapacitoroftheamplitude stablisationloopisfound
whose voltage determines the amplitude of the sawtooth.
The V-sync input can only handle unequal spacings of the
pulsesifthereisa2-sequence(e.g.Teletext312-313lines).
A 4-sequence from the 100 Hz box cannot be handled by
the amplitude loop.
The vertical sawtooth is internally connected to the
"geometry control" section. In this section S-correction,
vertical shift and linearity correction are added to the
sawtooth by I2C commands. The amplitude is controlled
by pin 24 (EHT-comp) to compensate for EHT variations.
From here the signal goes to one input of the internal error
amplifier. The other input is connected to pin 21 and the
output topin 20. By meansof anI2Ccommand theexternal
input pin can be selected as an inverting or non-inverting
input. This provision is made to handle both non-inverting
and inverting vertical output stages.
The block "geometry control" also generates the EW
parabola. TheI2C bus controllablefunctions are: parabola,
corner,trapeziumandpicturewidth. Bymeansofthesignal
at pin 24 this signal is corrected for EHT variations. The
EW drive output is available on pin 19.
Onthe digital side of this IC we find the following functions:
Pin 15 (SCL) is the Serial CLock and pin 14 (SDA) is the
Serial DAta. Pin 1 is the address pin and can either be
connected to ground or +12V. The three external DA
converters can be controlled by the bus: DACA, DACB &
DACC. With DACA the horizontal free funning frequency
of the TDA2595 can be adjusted. The other two DACs (pin
7 & 6) are not used. They can be used for H-shift and
H-phase control. See appendix A.
Pins 9 and 10 are output switch functions: not used in this
application. When pin 10 is programmed high, it can be
used as an input pin. Together with pin 17 it forms a
comparator. Pin 10 is connected to the Φ1 voltage of the
TDA2595 and pin 17 to the reference voltage. In this way
an I2C bus signal is available whether the horizontal
oscillator is in centre, locked and mute or coincidence so
information can be sent to the IN-input. This also makes
automatic fOadjustment possible.
The supply part of this IC contains 4 pins. Pin 18 is ground
for thegeometry and sawtooth part: pin 13 is groundfor the
output stages and I2C bus. Pin 12 is the +12V input and at
pin 16 an external capacitor is required for filtering the +5V
(internally generated).
Fig. 3 Deflection processor TDA8433.
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3.3 Vertical output stage
The vertical output stage is controlled by the well known
TDA3654. To make this stage suited for 100 Hz TV some
modifications of the ordinary solution are required. The
circuit is shown in Fig. 4.
A damper network is located in parallel to the vertical
deflection coil. The values depend on the characteristics
of the deflection unit. The line ripple that is injected from
the horizontal deflection coil is damped by the series
connectionof R55 and C48. R55 reduces the line ripple to
an acceptable value. C48 is added to block the relatively
lowverticaldeflectionvoltageinordertolimitthedissipation
in R55. A resonant circuit is created by C48 and the
inductance of the deflection coil. R54 is a critical damper
for this circuit to minimise excessive oscillations after the
vertical flyback.
The deflection current is sensed by two 1.5 Ωresistors in
parallel and fed back to the deflection processor. The
network C36, R45 and C35 is added for a stable loop
transfer because of the non resistive load at the output of
the TDA3654.
The output stage is ac coupled. The dc bias point is fixed
by the resistors R60 and R61. By the V-shift setting of the
TDA8433 vertical shift of the picture is possible.
An additional shift circuit is connected in parallel to the dc
shift circuit to make an alternating frame shift possible. It
consists of T11 and its series elements. When T11 is
conductingasmalldccurrentwillflowthroughthedeflection
coil. DuetotheS-correctionoftheverticaldeflectioncurrent
a smaller current is required at the top and bottom than in
the middle of the tube to guarantee proper interlacing
across the whole screen. Therefore, the waveform of the
shift current is derived from the parabola voltage of C49. A
potentiometer is provided because this interlace setting is
critical.
The drive signal required for this alternating frame shift is
generated by the 100 Hz conversion box.
If two independent shift signals are needed, the whole
circuit must be duplicated.
3.4 Horizontal oscillator
The horizontal oscillator used is the TDA2595. The
horizontal syncsignal (TTL level) is dividedand ac coupled
to the input pin 11. At pin 14 the reference current is set
by a 13 kΩresistor. The sawtooth capacitor for the
oscillatorisconnectedtopin16. Thefreerunningfrequency
is 31.25 kHz and determined by the value of its capacitor
andthe reference current. By varyingthe referencecurrent
the free running frequency can be adjusted. This is done
using the DAC-A output (pin 8) of the TDA8433 via resistor
R25, see section 3.2.
Fig. 4 Vertical output stage
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This oscillator is locked to the incoming sync signal by a
PLL (Phase Locked Loop). The starting point of the
horizontal sawtooth is compared with the horizontal sync.
If this starting point is not in the middle of the horizontal
sync pulse, an error signal will appear at pin 17. Via R27
the current of pin 14 is affected and thus the horizontal
phase can be locked. The loop filter consists of C25, R28
and C26.
The output of the oscillator is internally connected to a
second PLL Φ2 and to a phase shifter. The phase shifted
signal is available via an output stage at pin 4 (horizontal
output). Thissignaldrivesthedeflectionstage. Afeedback
signal of the deflection stage is applied to the other input
of the Φ2phase detector (pin 2). In this way the horizontal
flybackofthe deflectionstageis lockedtothe oscillatorand
thus to the sync as well. The loop filter of Φ2 consists of
one capacitor at pin 3. This second PLL has a much larger
bandwidth to compensate for the storage time variations in
the deflection transistors.
Atpin 6atwo levelsandcastlepulse isavailable. Itis mixed
with the vertical blanking signal of the TDA8433 or the
flyback of the TDA3654 to generate a three level
sandcastle. See also the description of TDA8433 section
3.2 and TDA3654 section 3.3.
Pin7 isthemute output. Thissignal issent tothe TDA8433
so that"oscillator locked" information is available at the I2C
bus.
As these kinds of ic are sensitive to supply pollution
provisionismadeforalocalsupplyfilterR21,C18andC19.
For layout recommendations see section 4.
3.5 Horizontal drive circuit
The horizontal drive circuit is a classical transformer
coupled inverting driver stage. When driver transistor T1
is conducting, energy is stored in the transformer. When
T1 is turned off the magnetising current continues to flow
in the secondaryside of the transformer thus turning on the
deflection transistor. At this time the voltage on the
secondary side of the driver transformer is positive
(VBE+IB*R6). When the driver transistor turns on again this
secondary voltage reverses. At the same time energy is
stored in the transformer again.
During this turn off action the forward base drive current
decreases with a controlled dIB/dt, thereby removing the
stored charge from the deflection transistor. The dIB/dt
depends on the negative secondary voltage and the
leakage inductance. When the drive circuit is designed
properly, the deflection transistor stops conducting when
its negative base current is about half the collector peak
current.
Fig. 5 Horizontal oscillator
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Fig. 6 Horizontal drive circuits
To prevent the deflection transistor from turning on during
flybackduetoparasitic ringingon thesecondaryside ofthe
driver transformer a damp resistor is connected in parallel
with the base emitter junction of the deflection transistor.
Also at the primary side of the driver transformer a damp
network is added (R4 & C5) to limit the peak voltage on the
driver transistor.
Fig. 7 Horizontal deflection
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3.6 Horizontal deflection
The horizontal deflection is the classical deflection stage
with the diode modulator which not only provides the EW
raster correction but alsoinner pincushion correction. Due
to the high frequency in combination with large currents
some problems do appear here. The horizontal deflection
coil needs 10.4A peak-to-peak. This results in a collector
peak current of 6-7A, too much to handle with one
BU2508A. So, two transistors are used in parallel. If the
print layout is made in a proper way no special precautions
are required to use this type of transistor in a parallel
configuration. (NB the circuit was constructed before the
BU2525A became available.)
Fortheflybackcapacitor thecurrent istoohigh aswell. So,
here, also, two devices are used in parallel. The
S-correction capacitors do not have problems in handling
the current.
There are two possible solutions for the damper diodes.
The BY359 is a high current damper diode available in
isolated and non-isolated TO220 packages. This device
has been re-designed for operation as a damper diode
specifically for 32 kHz deflection systems. An alternative
solution is to place a third diode in direct parallel with the
collector-emitter’s of the deflection transistors. This option
allowstwocheaperaxialdiodestobeusedinthemodulator,
eg BY328. This option is shown in Fig. 7.
For full performance of scan linearity a horizontal dc shift
circuit is incorporated. In an ordinary TV set the horizontal
off centre of the picture tube is compensated by the phase
shift of the horizontal oscillator. This, however, introduces
a linearity error in the deflection. In many cases this error
is acceptable; if not, it can be compensated by means of
an adjustable linearity corrector.
Fig. 8 East - West amplifier
The most proper way of picture alignment is the following:
the linearity corrector is only used for compensating the
linearity errorcaused bythe resistive partof the impedance
of the horizontal deflection yoke. The off centre of the tube
iscompensatedbyashiftcircuit. Therefore,adcshiftcircuit
is incorporated. This circuit has been built up around L5.
With P1 the amount of shift current can be adjusted and
with S1 the polarity can be selected. The horizontal shift
can be made bus controlled by using DAC-B or DAC-C of
theTDA8433. A suggestionfora suitableinterfaceis given
in Appendix 2.
3.7 East - West correction
Fig. 9 EHT generator
TheEW waveformis generatedbythedeflectionprocessor
TDA8433. An externalamplifier feeds this correction to the
horizontal deflection stage. It is injected in deflection via
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L2. The possible corrections are: picture width, EW
parabola, corner correction, trapezium and EHT
compensation.
The EW amplifier is shown in Fig. 8. It consists of a
Darlington power transistor, T6, a differential amplifier, T4
& T5, and a feedback network R13 R14. R12 is added for
proper dc bias. Because this amplifier has a non real load,
special attention is paid to loop stability. Across T6 there
isamillercapacitor, C14. Thelineripple currentofL2 flows
mainly through C15 and R16.
3.8 EHT generation
The darker glass requires a higher EHT power for an equal
light output. An increase of only the beam current has two
disadvantages: a larger spot size and a higher drive from
the video amplifiers. An increase of only the high voltage
wouldcomeinconflictwiththelegislationonX-rayradiation.
As a compromise an EHT of 27.5kV @ 1.3mA is chosen.
A new design of LOT is used, see Fig. 9. This LOT is a
four layer diode split box (DSB) design with extra high
voltage capability. From an integrated potentiometer the
adjustable focus and grid 2 voltages are taken.
3.9 Auxiliary supply
Someoftheauxiliary suppliesare takenfrom theLOTsuch
as heater, video and frame supply. The other auxiliary
supplies are taken from a separate transformer. The
philosophy behind this concept is explained in section 2.2.
The auxiliary transformer is connected in parallel to the
LOT. Onthesecondarysideofthistransformertheauxiliary
voltages are taken. These outputs supply the signal
processing circuitry (5V @ 5A, 12V @ 1A, -12V @ 1A).
Theprimaryinductanceofthistransformerisrelativelyhigh,
so the increase of collector current in the deflection
transistor is low. To adjust the output voltage the primary
winding has some taps. Due to the relative high ESR of
the 5V smoothing capacitors a π-filter is required.
With moderate current levels all the auxiliary supplies can
be taken from the LOT.
4. PCB design considerations
For general information see reference 3.
4.1 TDA2595
The following tracks and pins of the TDA2595 are critical
and need special attention:
* The track length at pin 14 - reference pin - to its
peripherals should be kept as short as possible.
Fig. 10 Auxiliary supply
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* The peripheral components connected to pins 14, 16,
3,17and15shouldbeconnecteddirectlytotheground
of this ic.
* The ground track of this ic may not carry current from
other parts of the set.
* As this ic is sensitive to high frequency ripple on the
supply rail, local decoupling is essential.
4.2 TDA8433
The following tracks and pins of the TDA8433 are critical
and need special attention:
* The components connected to pins 4, 5, 12, 16, 19, 22
and 23 should be connected tothe analog ground (pin
18) and as close as possible.
* The track length of the pins 4, 5, 22 and 23 should be
as short as possible.
* The ground track of this ic may not carry current from
other parts of the set.
* Local decoupling is essential because this ic is
sensitive to high frequency ripple on the supply rail.
4.3 Horizontal deflection and supplies
ThiskindofcircuitcarriescurrentswithhighdI/dt. Theloops
that contain these currents should have an area as small
as possible to limit magnetic radiation. Examples are the
loop ofdeflection coil withthe deflection transistors, diodes
and flyback capacitors. Also the loop formed by smoothing
capacitor C43, primary of LOT and deflection transistor
should be kept small.
In case of rectifiers the ground track between transformer
winding and smoothing capacitor may not be a part of any
other ground track.
4.4 Drive circuit
To ensure current balance in the deflection transistors, the
base andemitter tracks of the two transistors should be as
similar as possible to create thesame impedances for both
transistors.
5. Oscillograms
All oscillograms were taken under nominal load conditions
of the auxiliary supply and 1mA beam current.
Oscillogram 1:
In this oscillogram the upper two traces show the average
deflection current (2A/div) and the voltage across the
deflection transistors (200V/div). The peak VCE is 1244V.
Thelowertwotracesaretheminimumandmaximumvalues
at the mid-point of the diode modulator (100V/div) due to
EW modulation.
Remarks:
At the end of flyback there is a negative over shoot at the
collector voltage. This is caused by the forward recovery of
the damper diode.
Oscillogram 2:
The lower trace is the current in the deflection transistors.
The upper trace is the current in damper diode D4 and the
middle trace is the current in the upper flyback capacitors
(C7 + C8). All current settings 2A/div.
Remarks:
At the end of flyback the current in the flyback capacitor is
taken over by the damper diodes. Due to parasitic
capacitanceandinductance ringingoccurs. 5µs later there
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is a negative current in the deflection transistor. This is
reverse conducting of the base-collector of the transistor
caused by the fact that the base drive is already turned on.
Oscillogram 3:
Inthisoscillogramtheupper traceisthe currentinthe lower
flyback capacitors C9 + C10. The second trace is the
current in the diode D5. In the bottom part the current in
the bridge coil is given and the deflection current is shown
once more as a reference. All settings 2A/div.
Oscillogram 4:
In this oscillogram the deflection transistor ICand VCE are
given as a reference. The upper trace is the current in the
thirddiodeD3. As soonasthedeflectiontransistoristurned
on the current of D3 is taken over by the transistors. All
current settings 2A/div.
Oscillogram 5:
The upper trace is the VBE of the deflection transistors
(5V/div). The middle trace is the IBof the deflection
transistors (1A/div). The lower trace is the VCE of the drive
transistor T1 (50V/div).
Remarks:
The over shoot at the rising edge of the driver transistor is
causedbytheleakageinductanceofthedrivertransformer.
By means of the damping network R4, C5 this over shoot
is limited. This network is chosen in such a way that the
ringing is critically damped.
The base drive circuit is designed in such a way that the
peak of the negative base drive current, IBoff,is
approximately half the collector current, IC. During turn off
the VBE of the deflection transistor should remain negative.
To achieve this the ringing is damped by R7 and R8.
Oscillogram 6:
This split screen oscillogram was made with two different
time base settings. In the upper grid the minimum and
maximum values of the flyback pulses across C9 + C10 of
the diode modulator are given under nominal conditions.
The lower grid shows the amplified EW drive signal
(collector T65V/div) and the output of the TDA8433 (pin19
2V/div).
Remarks:
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At the collector of T6 some line ripple is visible.
Oscillogram 7:
The upper trace gives the vertical sync signal (1V/div,
100µs/div). The middle trace is the sandcastle (5V/div,
100µs/div). Thelowertraceisthesandcastleduringvertical
scan (5V/div, 10µs/div).
Remark:
Thisthree level sandcastlepulse is the sum ofthe two level
sandcastle of the TDA2595 and the vertical blanking of the
TDA8433.
Oscillogram 8:
The upper trace is the generated sawtooth at pin 22 of the
TDA8433 (5V/div). The second trace is the output signal
of the error amplifier of the TDA8433 pin 20 (5V/div). The
third trace is the sawtooth current in the vertical deflection
coil (1A/div). The lower trace is the output signal of the
vertical output amplifier TDA3654 pin 5 (20V/div).
Remark:
Due to the L/R of the vertical deflection coil the current in
the coil can not follow the fast retrace time of the sawtooth
generator. The output amplifier clamps after the flyback to
2xVb. Whenthecontrol looplocksafterthe flyback,a slight
voltage overshoot can be found at the output of the
TDA3654. This is damped by C48, R55 and R54.
Oscillogram 9:
The lower trace is the voltage at the foot point of the line
outputtransformer(10V/div). Thissignalisarepresentation
of the EHT variations needed by the anti breathing.
The upper trace is the EW waveform at T6 (5V/div). On
the EW waveform a correction signal is added to prevent
the picture from breathing.
6. References
Information for this section was extracted from
"32kHz/100Hz deflection circuits for the 66FS Black Line
picture tube A66EAK22X42"; ETV89012 by J.v.d.Hooff.
1. P.C.A.L.E. report ETV8906. "32 kHz / 100 Hz
deflection circuits for the 78FS picture tube" by Mr.
J.A.C. Misdom.
2. C.A.B. report ETV8612. "Computer controlled TV; the
deflectionprocessorTDA8432" by Messrs.E.M. Ponte
and S.J. van Raalte.
3. C.A.B. report ETV8702: "EMC in TV receivers and
monitors" by Mr. D.J.A. Teuling.
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7. Circuit diagrams
Fig. 11 Horizontal Deflection and EW Amplifier
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Fig. 12 Deflection Processor and Horizontal & Vertical Oscillators
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Fig. 13 Auxiliary supply, EHT Generator and Vertical Output Stage
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Appendix A
Fig. A1 Bus controlled dc shift
The deflection processor TDA8433 has three DAC’s. In
this application only one DAC (DAC-A) is used. In this
section some ideas are given to use the other two DAC’s.
DAC-B is a 6-bit DAC, like DAC-A, and is controlled by the
H-PHASE register. Its output voltages can be controlled
from 0.5V to 10.5V typically. The output resistance is less
than 1kΩ.
DAC-C is a 2-bit DAC and is controlled by the VTRA and
VTRC bits. Its typical output characteristic is:
VTRA VTRC Output voltage Output resistance
0 0 12V 7.5kΩ
0 1 5.3V 3.3Ω
1 0 1.7V 1.0kΩ
1 1 0.3V <1kΩ
All settings in the set that are now manual controls can be
madeI2C bus controlled by using one of these DAC’s. The
only restriction is that the alignment is controlled with a dc
voltage. Otherwise an interface circuit is needed.
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Appendix B
As a degaussing circuit the following suggestion is given.
Fig. B1 Degaussing Circuit
Parts list:
Dual degaussing PTC : 2322 662 96116
Degaussing coil : 3111 268 20301
Oscillogram 10:
Current in degaussing coil 2A/div. 376
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SMPS Circuit Examples
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4.3.1 A 70W Full Performance TV SMPS Using The TDA8380
The following report describes the operation of a 70W full
performance switched mode power supply for use in
television.
The TDA8380 SMPS control ic is used in a mains isolated,
asynchronous flyback converter configuration.
The power supply incorporates the following features:
· Full mains range (110 - 265Vrms)
· AT3010/110LL SMPS transformer
· BUT11A switching transistor
· Standby (suppression of output voltages by 50%)
· Standby supply (5V, 100mA)
· Facility for synchronisation (using apulse transformer)
· Short start-up time (less than 0.3 sec at 220Vrms)
· Provision for anti-breathing circuit
· Output voltages 147V/57W, 25V/5W, 16V/7.5W
A full description of circuit operation, a circuit diagram and
circuit performance figures are given.
A further additional circuit diagram is included in which the
above power supply incorporates a power MOS switching
transistor for a mains range of 90 - 135Vrms. Also details
are given on an extension to the power capability of the
supply, up to 120W output, for European mains using the
bipolar switching transistor.
1. Introduction
The TDA8380 control ichas been designed to enable safe,
reliable and efficient SMPS to be realised at minimum cost
for TV and monitor applications. For further information on
the ic, reference should be made to ‘Integrated SMPS
Control Circuit TDA8380’ (ref. 1).
The 70W design employs a currently available
AT3010/110LL foil wound transformer and the BUT11A
bipolar switching transistor.
Feedbackistakenfromthesecondarysidetogivelessthan
1% line and load regulation over the whole range. The
output voltageis suitably divideddown and compared inan
error amplifier with a fixed reference voltage. The error
amplifierthendrivesanoptocoupler,whichpassestheerror
signal to the primary side directly into the TDA8380. A
secondary side error amplifier is used to reduce the
importance of the optocoupler characteristics.
Standby is achieved by injecting asignal into the feedback
looponthesecondaryside,suppressingall outputvoltages
by50%. A5V standbysupplyisavailablerelievingtheneed
for a separate standby supply. During standby conditions
the line output oscillator is halted to disconnect the main
B+ load.
Synchronisation of the power supply to external control
circuits is possible through a loosely coupled pulse
transformer.
Appendix A gives a circuit diagram and a short description
ofthe useofthepowerMOSswitchingtransistorinthe70W
supply. The mains range is 90 - 135Vrms.
Appendix B gives notes on how to extend the power
capability of the 70W power supply to 105W, 32 kHz and
120W, 30 kHz. Both these power supplies have a mains
range of 180 - 265Vrms.
2. TV SMPS design
Flyback versus Forward Converter.
Although this ic can be applied in any type of SMPS, for
example in FORWARD (or Buck) or FLYBACK (or
Buck-boost) converters, the preferred SMPS type for TV
applications is the Flyback converter. This is mainly
because it allows for mains isolation of the TV chassis.
Other advantages it affords in comparison with a forward
converter are:
(a) It doesnot need ‘crow-bar’ protection against the input
voltage appearing across the chassis in the event of
short-circuit failure of the power switching transistor.
(b) The load permissibleon auxiliary output supplies is not
related (and, therefore, not limited) by the main line
timebase supply (B+ voltage) load. Thus the auxiliary
supply is available when there is no B+ voltage load,
and this is important for servicing and fault finding on a
TV chassis.
With the Flyback converter, however, mains pollution and
visible interference require to be minimised by careful PCB
layout and customarily a mains input filter is used.
Discontinuous versus Continuous Current Mode
Operation:
Operationcanbeinthe‘continuous’orinthe‘discontinuous’
current mode.
In the discontinuous mode the power switching transistor
is not allowed to switch onuntil the SMPS transformer core
is demagnetised. This has the advantages:
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(a) It is inherently a safer mode of operation, since for all
operating conditions, other than a dead-short of the
output or a very severe overload transient occurring
when the power transistor is conducting near peak
current, it is not possible for the core to saturate. To
protect for these two exceptions, the very fast (second
level) current protection is included.
(b) Steep current pulse edges at switch-on are eliminated
(important from point of view of Radio Frequency
Interference problems).
Satisfactory performance, in terms of voltage regulation
and input mains voltage range, can be obtained using the
discontinuous mode which is, therefore, preferred for TV
applications.
For this type of converter, thevoltage transfer function can
be deduced from the Volt-second equilibrium condition for
a unity turns ratio transformer:
(1)
where:
Taking into consideration the transformer turns ration n =
Np/Ns, then:
(2)
where: = oscillator period
= transformer primary turns
= transformer secondary turns
= on time of output transistor
= conduction time of output rectifier
= output B+ voltage
= input DC voltage
The limit condition for ‘discontinuous’ current mode
operation occurs at minimum input voltage and maximum
load.
Thus, for this condition
so that
(3)
The power output (including losses supplied via the
transformer) is:
(4)
where: = primary inductance of transformer
= frequency of operation
from which general expressions for d, Lp and f can be
obtained in terms of the power.
Thus, for a given transformer(n,Lp) the value of dmax can
be calculated from (3) and the required frequency of
operation from:
(5)
The peak current in the power switching transistor is given
by:
(6)
The peak voltage across the power switching transistor
(excluding ringing) is:
Since most transformers produce ringing, a clamp circuit
may be necessary and in order to slow the rising edge of
the voltage a snubber circuit is usually required.
3. General circuit description
This section gives an overall general description of the
power supply.
Fig. 1 shows a block diagram of the circuit functions.
Descriptionsof specificcircuitswillbecarriedoutinthenext
section.
3.1 Mains filter
This is positioned at the ac mains input. Its function is to
minimise mains pollution resulting from RFI generated
within the SMPS due to fast transients of voltage and
current. It is designed to meetthe required mains pollution
regulations (C.I.S.P.R. Special Committee on Radio
Frequency Perturbation).
3.2 Rectification and Smoothing
The mains voltage is rectified and smoothed to provide a
dcsupplywhichisswitchedthroughtheSMPStransformer.
3.3 SMPS Controller
This drives the power switching transistor regulating the
frequency and the amount of current pulsed through the
transformer primary. Thus, the controller regulates the
energy transferred to the secondary windings. A voltage
feedback signal which is representative of the output
voltage is fed back to the controller in order to regulate the
output voltage. At start up the supply voltage for the
controllericisderivedfromtherectifiedmains. A‘take-over’
winding on the transformer supplies the ic once normal
operation is established.
3.4 Transformer secondary circuits
The dc output voltage is obtained by simple rectification
and smoothing of the transformer secondary voltage.
f=dmax2×Vimin2
2×P×Lp
Ip =Vi ×d
Lp ×f
Vi +(n×Vo)
Vi ×d×T=Vo ×d×T
Vo
Vi =d
d
d=(1−d)
n×Vo
Vi =d
d
T
Np
Ns
d
d
Vo
Vi
d=(1−d)
n×Vo
Vimin =dmax
(1−dmax)
P=Vi2×d2
2×Lp ×f
Lp
f
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3.5 Feedback Attenuator
The output voltage to be regulated is fed back via an
attenuator to the error amplifier.
3.6 Error Amplifier
The error amplifier compares the feedback signal with a
fixed reference voltage to give an error signal which is
passed to the primary side via an optocoupler.
Mains isolation is provided within the optocoupler and the
power transformer, between the input primary and
take-over, and the output secondary windings.
4. Detailed circuit description
This section gives a detailed description of each of the
functions of the power supply circuit (Fig. 2).
4.1 Mains Input and Rectification
Diodes V1 to V4 rectify the ac mains voltage and, together
with a smoothing capacitor C13, provide a dc input HT
voltage for the SMPS. R1 is placed in series with the input
to limit the initial peak inrush current whenever the power
supply is switched on when C13 is fully discharged.
C1 and C3 together with L1 form a mains filter to minimise
the feedback of RFI into the mains supply.
C6 to C9 suppress RFI signals generated by the rectifier
diodes.
Asymmetrical mainspollution is reduced bythe insertion of
R26 and C18 between primary ground (‘hot side’) and
secondary earth (‘cold side’) of the power supply. These
components are required to satisfy the mains isolation
requirements.
4.2 Control ic TDA8380
This section describes the function of each pin of the
TDA8380 and its associated components.
Pin 1 - Emitter of Forward Drive Transistor:
The TDA8380 incorporates a direct drive output
stage consisting of two NPN transistors. The
collector and emitter of each are connected to
separatepinsoftheic(pins1,2,15,16). Theforward
base drive current for the switching transistor is
limitedby R15. C16 actsas avoltage sourceequal
to the zener voltage of V7 and is used for the
negative base drive.
When the reverse drive transistor is turned on the
zener voltage appears across L2, causing stored
chargetoberemovedfromtheswitchingtransistor,
thereby ensuring correct storage time and
minimum transistor dissipation during turn-off.
Pin 2 - Collector of the Forward Drive Transistor:
Connected through a resistor to the ic reservoir
capacitor.
Pin 3 - Demagnetisation Sensing
Demagnetisation protects the core of the
transformer against saturation by sensing the
voltage across a transformer winding. In this
applicationoperationisinthediscontinuouscurrent
mode. Sensing is achieved by resistor R10 from
the take-over winding of the transformer to pin 3 of
theic. Fig. 3 illustrates demagnetisation operation
at low mains where the turn-on pulse is delayed
until demagnetisation of the transformer is
complete.
Pin 4 - Low Supply Trip:
Connected to the icground (pin 14), the low supply
protection level is 8.4V.
Pin 5 - IC supply:
On power-up the ic supply is first drawn from C15.
This capacitor is charged up directly from the
rectified mains through bleed resistors R21 and
R24.
Once the SMPS is running, the supply for the ic is
takenoverbytheSMPStransformer. R12prevents
peak rectificationof spikes. V8 rectifies the flyback
signal which is smoothed by C15 to give a dc level.
R16 limits the current drawn by the forward drive
transistor. R9 and C5 provide a filtered dc supply
to pin 5.
Pin 6 - Reference Current:
Thispinallowsthe externalsetting oftheIC current
source. This is set by R11.
Pin 7 - Voltage Feedback:
This is the input to the internal error amplifier for
primary side feedback. Feedback in this case is
taken fromthe secondary side and passedthrough
a separate secondary side error amplifier where it
is compared with a reference voltage. The error
signal is then passed directly into the duty pin (pin
9) via an optocoupler.
To ensure that the Transfer Characteristic
Generator (TCG) in the ic remains optional a
‘pseudo’ feedback voltage from the ‘take-over’
winding of the SMPS transformer is applied to pin
7. R3 and R4 provide a nominal 2.5V level at pin
7 during normal operation of the power supply.
Pin 8 - Stability:
This is the output of the error amplifier which is left
open circuit.
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Pin 9 - Duty:
This is the input to the pulse width modulator and
is directly driven by the optocoupler transistor. R2,
C2 and C27 form a frequency compensation
network.
Pin10- Oscillator:
The frequency of the internal oscillator is set here
by C4 and R11 on pin 6 (nominally 25 kHz).
Pin11- Synchronisation:
This is achieved by a loosely coupled pulse
transformer passing sync pulses from the
secondary to the primary side of the power supply
(see later section).
Pin12- Slow Start:
The slow start option is selected here by the use of
capacitor C11. Fig. 4 shows a typical slow-start.
Pin13- Over-Current Protection:
To keep the collector current of V10 within safe
operating limits over-current protection is
incorporated into the power supply. R27 is the
collector current monitoring resistor providing a
negative going signal. This voltage is then shifted
to a positive level with respect to ground potential
by a reference current from the ic flowing through
R14. An extra voltage shift is provided by R34
which varies with the ic supply voltage. This is
particularly useful in output short circuit conditions.
If the main regulated output is progressively short
circuited, then all SMPS transformer flyback
voltageswilldecrease,respectively,and hencethe
shift level of the current protection function leading
to lower short circuit output currents (current
foldback). The signal at pin 13 is then compared
with two internal voltage levels to provide the two
forms of current protection.
(The addition of R34 may not work in other power
supplies using the TDA8380 because careful
attention has to be given to the ratio of current
through R34 to current output at pin 13 and to the
start-up sequence of the power supply at different
mains and loads. Conventional current protection
canbeachievedbyomittingR34andchangingR14
to 13 kΩand V13 to BYW95C).
Fig. 5 illustrates the current protection waveform.
Pin14- Ground
Pin15- Emitter of Reverse Drive Transistor:
Grounded to the emitter of the switching transistor.
Pin16- Collector of reverse drive transistor.
4.3 Error Amplifier
Theexternalerroramplifierconsistsof twoPNPtransistors,
V15 and V16, connected to form a high gain comparator.
The stabilized reference voltage for the comparator is
derived from a series-connected resistor R28 and zener
diodes V5 and V6 at the SMPS output. The voltage to be
compared with the reference voltage is a sample of the
147Voutput derived froma potential divider (R29,R31 and
R5). The optocoupler is directlydriven with the error signal
from the comparator. The level of the 147V output can be
adjusted by R5.
4.4 Standby
In standby mode the power supply output voltages are
suppressed to 50% of their normal level. Standby is
achieved by reducing the reference voltage used in the
comparator circuit and thus the power supply regulates at
a lower output voltage level. A +5V dc level is applied to
the standby input, which turns transistor V14 on. The
voltage reference level is halved from 12.4V to 6.2V and
the main 147V output is reduced to 75V. In this condition
the power supply still maintains a 5V standby supply. In
the television receiver during standby the line output
oscillator should be halted to disconnect the main 147V
load.
To return the power supply to its normal operating levels,
the standby input is removed.
The speed of transition to and from standby is controlled
by the time constant of R13, R32 and C23.
4.5 Synchronisation
Synchronisation of the power supply is achieved by a
loosely coupled mains isolated pulse transformer. Sync
pulses of +5V are applied to the sync input at a frequency
slightly lower than the free running frequency of the power
supply. R6 limits the current in the primary winding of the
pulse transformer and R8 loads the secondary winding.
The pulse transformer differentiates thesync pulse input to
create negative and positive going transitions of the sync
input. Theac coupling(C14)shifts theentiresignalpositive
and theinternalcircuitry of the icclamps the negativegoing
excursions to 0.85V. The positive going spikes are
removed by a transistor in the ic and the negative going
spikesare used tosynchronise the oscillator. Fig. 6 shows
plots of how the power supply is synchronised to a lower
frequency.
A series RC network (C28, R35) is connected from pin 11
to ground to filter out high frequency noise which may
interfere with synchronisation.
If the synchronisation option is not to be used, the sync
input may be left open circuit. Another alternative is to
short-circuit C14 and remove T2.
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4.6 Beam Current Limiting (BCL)
Anti-breathing technique, whereby the 147V voltage is
reduced for increasing beam current in such a manner as
to compensate for the increase in picture size due to the
fall in EHT. The components concerned are R30, C24, R7.
4.7 Power Switching Transistor
Pulsing of the transformer is carried out by the BUT11A
bipolar power transistor under the control of the TDA8380.
Fig. 7showsplotsofthecurrentthroughandvoltageacross
the BUT11A. The base drive waveforms are shown in
Figs. 8(a)-(b) during standby conditions.
Fig. 9is a plotof the instantaneous powerdissipated in the
transistor during turn-off.
4.8 Snubber Network
A snubber network has been added across the switching
transistor to protect it from excessive switching dissipation
and to suppress ringing on the SMPS transformer.
The dV/dt limiter consists of V9, C17, R22 and R23. When
V10 isswitched off, part of the energy stored in theleakage
inductanceoftheSMPStransformerwillchargeC17. When
V10 is switched on again this energy is dissipated in R22
andR23. Whensuchanetworkisomitted,thisenergymust
be dissipated in the switching transistor itself.
R22 and R23 are calculated in such a way that they also
actasanetwork,dampingtheresidualenergyinthewinding
capacitance of the transformer when the secondary
rectifiers have stopped conducting.
4.9 Outputs
There are three secondary rectifiers; the 147V (scan
voltage for deflection stage), 25V (audio supply) and 16V
(small signal supply). The 5V standby supply is derived
from a regulator connected to the 16V output.
R25 and C25 form a damping network to dissipate the
energy in the high frequency ringing on the B+ secondary
winding. Fig. 10 shows the current through and voltage
across the B+ winding.
A short circuit or overload of these outputs will cause the
power supply to repeatedly go through the slow start
procedure.
5. Performance specification
Mains input: 110 - 265V ac 50 - 60 Hz
Outputs: B+ 147 V 57 W
Audio 25 V 5 W
L.T. 16 V 7.5 W
Standby 5 V 0.5 W
Switching frequency: 25 kHz
Efficiency (normal operation): 72 %
Line and load regulation: 0.1 %
Start-up time (220V rms, 300 msec (B+)
full load): 225 msec (+5 V standby)
Max. collector current: 2.3 A
Max collector voltage: 870 V
Forward base current: Normal operation 0.30 A min.
0.39 A max.
Standby (*) 0.20 A min.
0.24 A max.
ic supply voltage: Normal operation 18.5 V min.
21.0 V max.
Standby (*) 8.8 V min.
9.7 V max.
Ripple output voltage (110V rms, 50 Hz, full load):
B+ L.T. Audio Standby
Frequency (mV) (mV) (mV) (mV)
25 kHz 600 230 145 20
199 Hz 230 50 60 -
* The only load in this condition is the standby load.
6. Output short-circuit foldback
The SMPS incorporates duty factor foldback protection for
short circuits on the 147V (B+) output. Fig. 11 shows the
plot of the foldback characteristic for increasing load onthe
147V output using conventional protection and current
foldback techniques.
8. References
Ref. 1. "Integrated SMPS Control Circuit TDA8380".
Philips ComponentsPublication Number 9398 358
40011
December 1988.
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Appendix A
70W FULL PERFORMANCE USING POWER MOS (BUK456-800A)
A power MOS switching transistor was incorporated into
the 70W power supply design. This new power supply has
a mains range of90 - 135V rms. A circuit diagram is given
in Fig. 12. Oscillograms of the power MOS gate and drain
switching waveforms are given in Figs. 13 and 14.
Alterations to Existing 70W Bipolar Transistor Design
(i) The value of C17 in the snubber is smaller, hence less
dissipation in the snubber resistors. The dV/dt at the drain
isnowhigher,butthepowerMOStransistorhasmuchlower
switching losses than the bipolar transistor.
The smaller value of C17 causes the 100 kHz ringing on
the primary winding of the SMPS transformer after flyback
to be more prevalent. This ringing has an effect on the
demagnetisationfunctioncausingprematureoperation. To
overcome this a resistive divider network has been used
on pin 3 to minimise the effect of ringing.
(ii) The value of R14, the current protection shift register,
is increased. This is to compensate for the fact that the
power MOS transistor does not suffer from storage effects
at turn-off.
(iii) The filtering on the take-over winding for the ic supply
is increased. This is because the average current
demandedbythe gatedriveof thepowerMOSis muchless
thaninthecaseofthebipolartransistor. Energyinswitching
spikes onthe flybackvoltage cannot be channelledintothe
gate of the power MOS and so has to be dissipated in
increased filtering.
A smaller value for the gate-source resistor is used to
provide extra loading on the transformer winding.
(iv) C13 is increased to filter the higher current ripple at
low mains voltages.
(v) A larger heatsink for the switching transistor is
necessary due to the higher on-resistance of the power
MOS transistor facilitating the need for higher heat
dissipation.
Performance Specification
Mains supply: 90 - 135V rms
Switching frequency: 20.8 kHz
Outputs: B+ 147 V 57 W
Audio 25 V 5 W
L.T. 16 V 7.5 W
Standby 5 V 0.5 W
Regulation: 0.1 %
Peak drain voltage: 650 V
Peak drain current: 2.3 A
Start-up time 300 msec
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Addendum
Alternative Cheap BUT11A Base Drive Design Eliminating 5 W Zener Diode
Fig. A1 - Alternative Cheap BUT11A Base Drive.
An alternative base drive for the power switching transistor
(BUT11A) has been designed to eliminate the 5W zener
diode 1N5339B (V7) to reduce cost.
Alternative Low Cost Base Drive
This design has not been implemented into a PCB design
yet, but the existing PCB design requires little alteration to
accommodate the changes.
When the forward drive resistor is turned on at the start of
the duty cycle, a current defined by R15 passes through
C16 and into the base of the BUT11A. The 1 kΩresistor
in parallel with C16 discharges the capacitor when the
SMPSisofftohelpstarting atlow mains. Whenthereverse
drive transistor is turned on, the 5.1V zener diode appears
across C16 clamping the voltage across it, thus a reverse
current flows from the base of the BUT11A through C16
and L2 turning off the power switching transistor. Some
forwardcurrentdoes flowthroughthe5.1Vzenerdiode,but
not enough to warrant a power zener. The BAX12A diode
acrosstheinductoristopreventlargenegativegoingspikes
appearing at pin 1 of the ic; this can also be used in the
previous base drive.
Base Measurements
Forward base current: 250mA min
400mA max
Standby mode 190mA min
(standby load only) 250mA max
BUT11A storage time: 1.4µsec
Fig. 1 - Block Diagram of SMPS with Secondary Side Feedback via an Optocoupler
MAINS MAINS
RECTIFICATION
SMPS
TRANSFORMER
SMPS
CONTROLLER
SECONDARY
RECTIFICATION
CIRCUITS
FEEDBACK
ATTENUATOR
ERROR
AMPLIFIER
FILTER
ISOLATION
START-UP
SUPPLY
DRIVE
TAKE-OVER SUPPLY
FEEDBACK
OUTPUTS
Vref
MAINS SUPPLY
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Fig. 2 - 70W Full Performance TV SMPS (Bipolar switch)
Fig. 3 - 70W Full Performance TV SMPS (Power MOS switch)
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Fig. 4 - Demagnetisation operation. Oscillogram of the
Oscillator Waveform and Transformer Primary Current
Fig. 5 - Slow Start. Oscillogram of the Voltage at the Slow
Start Pin (TDA8380) and Current through the Switching
Transistor.
Fig. 6 - Current protection. Oscillogram of voltage at Pin
13 (TDA8380) and the Current through the Switching
Transistor
Fig. 7 - Synchronisation. Oscillogram of Oscillator
Voltage, Voltage at Pin 11 (TDA8380) and Sync Input
Voltage
Fig. 8 - Switching waveforms. Oscillograms of the current
through and voltage across BUT11A.
Fig. 9 - BUT11A Base Waveforms. Oscillograms of
Base-Emitter Voltage and Base Current Waveforms for
BUT11A
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Fig. 10 - BUT11A Base Waveforms During Standby.
Oscillograms of Base-Emitter and Base Current
Waveforms for BUT11A in Standby Conditions
Fig. 11 - Turn off dissipation in BUT11A. Oscillogram of
Collector-Emitter Voltage and Collector Current for
BUT11A
Fig. 12 - B+ Winding. Oscillogram of Voltage Across and
Current Through the B+ Winding
Fig. 13 - Plot of Duty Factor Foldback using Current
Feedback and Conventional Foldback Techniques
Fig. 14 - Oscillogram of Drain-Source Voltage and Drain
Current for Power MOS Transistor (BUK456-800A) at
Full Load, 110V rms
Fig. 15 - Oscillogram of Gate Current and Gate Source
Voltage for Power MOS Transistor (BU456-800A) at Full
Load, 110V rms
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4.3.2 A Synchronous 200W SMPS for 16 and 32 kHz TV
Description of 200W Switched Mode Power supply
incorporating the AT3020/01A transformer, the TDA8380
control IC, one opto-coupler for feedback, synchronisation
and remote on/off. The SMPS is intended for TV and can
be synchronised to 32 kHz by flyback pulses of either 32
or 16 kHz. A 5V standby supply is also provided.
1. Introduction
In this report a description is given of a 200W SMPS circuit
and evaluation board, incorporating the TDA8380 control
IC, the new SMPS transformer AT3020/01A and the
BUW13power switching transistor. The SMPS is a flyback
converter that has been designed to handle a maximum
average outputpower of 200W and a peak powerof 250W.
The free running frequency of the SMPS is 34 kHz, while
it can also be synchronised down to 32 kHz by either 16 or
32 kHz line flyback pulses. For testing purposes no
pre-loading is required. The circuit operates at a mains
input voltage of 185-265VRMS, 50-60Hz. The output
voltages are 150V, 32V and 16V. The 150V output is short
circuit proof, while the 32V and 16V can be made
short-circuit proof.
A new wire wound SMPS transformer has been designed.
Thistransformer,theAT3020/01AwithanEE46/46/30core
(grade 3C85), has a new winding technique which makes
the RFI screens superfluous. Thanks to its low leakage
inductance, the efficiency of the system is high (88%).
The control IC TDA8380 receives its start-up supply from
the rectified mains voltage. The takeover supply is derived
from a flyback and forward auxiliary winding on the
transformer. This IC offers many attractive operating
features:itdirectlydrivesthepowerswitchingtransistorand
incorporates several overload protections.
Due to its high current hFE, the BUW13 was chosen as the
power switching transistor. For an output power of up to
150W, the BUT12 can be used. A CNG82 opto-coupler is
used for feedback. If synchronisation is not required, the
cheaper CNX82A can be used.
For the supply of some digital IC’s in the standby mode, a
small self-oscillating supply is used: the so called µSOPS
(5V, 300mA). In the standby mode the output voltages will
be fully suppressed.
A printed circuit board (no 3634) is available incorporating
the 200W SMPS and µSOPS but without mains filter.
2. Circuit description
2.1 Block Diagram
Fig.1 shows the block diagram of the 200W mains isolated
flyback converter.
Fig.1. Block Diagram
Mains
Input
Circuit
u-SOPS
SMPS
Transformer
Opto
Coupler
Control
Circuit
Power
Switching
Transistor
Secondary
Rectifiers
Error
Amplifier
5V SB
150V
32V
16V
SYNC IN
STAND BY
220V
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Fig.2. Basic Circuit Diagram
The200WSMPS evaluationboarddoesnot containanRFI
filter, fuses or a degaussing circuit. These components
should be located on the inletof the mains cord into the TV
set. The mainsinput voltage isrectified by bridge rectifying
diodes and the dc supply to the SMPS transformer
(AT3020/01A) is smoothed by a 220µF buffer capacitor.
The control IC TDA8380 derives its start-up supply from
this dc voltage and as soon as the IC supply voltage
exceeds a certain limit, the IC is initialised. Hereafter, the
duty factor of the SMPS power switching transistor
(BUW13) increases slowly from zero upwards and its rate
of increase is controlled until the SMPS output voltage
reaches its nominal level. The take over supply is derived
fromaflybackandforwardrectifierconnectedtoanauxiliary
winding of the SMPS transformer. The SMPS is a flyback
converter that operates in the discontinuous mode. At the
secondary side the flyback voltage is rectified. One of the
output voltages is fed back via an attenuator circuit to the
error amplifier. The error signal is sent back via the
opto-coupler circuit to the duty cycle control input of the IC
TDA8380.
For standby purposes the µSOPS delivers a 5V supply. In
the standby mode the output voltages will be fully
suppressed. TheSMPSrunsatafixedfrequencyof34 kHz,
however, it can also be synchronised down to 32 kHz by
either 16 or 32 kHz line flyback pulses.
2.2 Basic Operation
Fig.2 shows the basic circuit of the mains isolated flyback
converter.
The control IC TDA8380 directly drives the power output
transistor. Whenthetransistorconducts,alinearincreasing
current flows through the primary winding of the
transformer. As a consequence energy is stored in the
transformer. After switching off the transistor, the stored
energy is transferred into the load via diode D. The
attenuated output voltage Vo is compared with the
reference voltage, REF, in the error amplifier. The error
signal is fed back via the opto-coupler tothe control IC. By
controlling the duty cycle of the drive pulses the output
voltage Vo is kept constant.
Theflyback converter under discussionhas beendesigned
for the discontinuous current mode. The principle of this
circuit has already been described in chapter 2 "Switch
Mode Power Supplies". For a nominal output voltage of
150V,185VRMS mains,amaximumloadof250Wandafixed
free running frequency of 34 kHz, the primary inductance
ofthe transformer can be calculated. The required primary
inductance is Lp = 420µH ± 10%.
An attractive feature of this SMPS is that it can be
synchronised down to 32 kHz by either 16 or 32 kHz line
flyback pulses.
3. Circuit diagram
The circuit diagram is given in section 8, Fig.3; detailed
information about several parts of the supply follows.
3.1 Mains input
The diode bridge D1 to D4 rectifies the mains input voltage
and the dc supply to the SMPS is smoothed by C5.
Capacitors C1 to C4 suppress the RFI generated by the
diodesinthemainsbridgerectifier. IfC5isfullydischarged,
the inrush current has to be limited by R1 to protect the
bridge rectifier diodes. During continuous operation of the
SMPS this resistor is for efficiency reasons short circuited
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by a thyristor, THY1. After the soft start of the SMPS,
thyristor THY1 is fired continuously by the peak voltage
clamp of the SMPS via R3 and C6.
3.2 Start-up supply
The control IC TDA8380 receives its start-up supply from
the mains rectified voltage by the low wattage resistor R4.
The IC is initialised as soon as the voltage on the supply
pin 5 reaches 17V. This takes approximately 1.5s
(Oscillogram6). Shortertimesarepossible byloweringthe
value of R4. During the timeleading up to theinitialisation
of the IC, the base coupling capacitor C10/C11 is
pre-charged. So, the power switching transistor T1 is
switchedoffcorrectlyduringthestartupperiod. Withaduty
cycle from zero onwards, the SMPS starts up. The take
over supply is derived from a forward and flyback auxiliary
winding on the transformer (AT3020/01A). The forward
rectifying diode D7 ensures that a temporary decrease of
the supply voltage of the IC is restricted. After a while the
flyback rectifying diode D6 directly provides all the current
neededbytheIC. DuringcontinuousoperationoftheSMPS
the supply voltage for the IC is about 17V.
3.3 Control IC
The integrated SMPS control circuit TDA8380 offers many
attractive operating features. It controls the SMPS power
throughput and regulation by pulse-width modulation. It
can directly drive the power switching transistor and it can
operate at a fixed frequency or a line locked frequency. A
detailed description is given in Reference [1]. The function
of each pin is described below.
Pin 1 Emitter of the forward drive transistor. It directly
drives the power transistor with a source current of
about 0.7A.
Pin 2 Collector of the forward transistor. This pin is
connectedviaR14tothe supply. ResistorR14 and
R15 mainly determine the source current of the
power switching transistor.
Pin 3 Demagnetisation sensing. For this flyback
converter, operating in discontinuous mode, the
voltageacrosstheSMPStransformerissensedvia
R12 and R13.
Pin 4 Low supply-voltage protection level. This pin is
connected to ground,so the min.Vcc of the ICis set
at 8.4V.
Pin 5 IC supply. When the mains input is applied to the
SMPS, the IC supply reservoir capacitor C9 is
charged by a current determined by resistor R4.
When the voltage at pin 5 reaches 17V, the IC
initialises and diode D6 rectifies the flyback signal
from winding 10/11 of the SMPS transformer to
supply the IC with 17V.
Pin 6 Masterreferencecurrentsetting. ResistorR11sets
the master reference current for the TDA8380 to
600µA.
Pin 7 Voltage feedbackand overvoltage protection. The
flyback signal from winding 10/11 of the SMPS
transformer is smoothed by D6/R7/C9, to give a dc
level that varies in proportion to variations in the
150V output. This level is reduced by the divider
R9/R10 and fed to pin 7.
Pin 8 In this application the feedback amplifier of the
TDA8380 isnot used. However, an overvoltage on
pin 7 will still activate a protection and slow start
sequence.
Pin 9 Output of the error amplifier. Not used.
Pin 10 Oscillator. A 680pF capacitor C15 is connected to
this pin; together with resistor R11 (4k3) the
oscillator frequency is set to 34 kHz.
Pin 11 Synchronisation. The trailing edge of the positive
sync-pulses,which are superimposedon thelinear
feedback signal, synchronise the oscillator.
Pin 12 Slow-start (capacitor C17) and maximum duty
cycle (R20).
Pin 13 Over current protection. The over current
protection safeguards the power switching
transistor for being overloaded with a too high
collector peak current. For that reason resistors
R22 to R26 in the emitter circuit of the power
switching transistor sense the collector current.
This negative going signal is dc shifted into a
positive signal with respect to ground by a dc
current from pin 13 flowing through R21, while C18
removes the spikes.
Pin 14 Ground
Pin 15 Emitter of the reverse drive transistor, connected
to ground.
Pin 16 Collector of the reverse drive transistor. See drive
of the BUW13 SMPS power transistor.
3.4 SMPS Transformer
As already mentioned before, the transformer
(AT3020/01A) has been designed to handle a maximum
output power of 200W and a peak power of 250W. The
nominalprimaryinductanceis420µH. Tokeeptheleakage
inductance (~2%) as small as possible, a turns ratio of 1:1
was chosen. The magnetic circuit of the transformer
comprises two Ferroxcube E46/23/30 cores, grade 3C85.
The coil is built-upin layers of copper wire, separated from
each other by insulation foil. Thanks to a clever winding
design no screens had to be applied, and as a result, the
size of this transformer could be reduced significantly with
respect to the transformer described in Reference [2].
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The energy stored in the leakage inductance will be
dissipated in the dV/dT limiter (D8/C13/R19) and peak
voltage clamp (D5/C7/R5); the energy stored in the
parasitic winding capacitance in the power switching
transistorT1anddampingnetworks(R6/C8and R37/C28).
3.5 Power switching transistor
By fixing the primary inductance of the transformer and its
operatingfrequency,thecollectorpeakcurrentofthepower
switching transistor is fixed. At a peak output power of
250WtheIcpeakisapproximately6.5A. Ontheotherhand,
the maximum base drive is determined by the control IC
TDA8380: Isource max = 0.75A. The BUW13 has a sufficient
highcurrentgainand,moreover,theIboff couldbekeptwithin
the limit of the control IC: Isink max = 2.5A. For slightly lower
maximum power (150W) the BUT12 can be used as the
power switching transistor. Note that, in that case, current
sensing resistor R21 has to be reduced.
The correct forward drive of the transistor is provided by
the supply voltage of the IC and R14/R15, resulting in an
Ibon of 0.7A. To obtain a correct negative base drive, the
bias voltage across C10/C11 is kept constant by three
BAW62 diodes (D9 to D11). During turn off, inductor L1
(1.7µH) in combination with the bias voltage, determines
the negative base current (-dI/dt) of the power switching
transistor. R17 and C12 damp the ringing of the
base-emitter and prevents parasitic switch on of T1 during
the flyback.
3.6 Secondary rectifiers
Thethreesecondaryflybackrectifiersdeliverthe150V(line
deflectionsupply),32V(audiosupply)and16V(smallsignal
supply). A 12V stabiliser is not provided on the PC board.
The load determines the dissipation in the rectifyingdiodes
and hence the size of the heatsink (D15) and copper area.
The number of electrolytic capacitors is determined by the
load (ripple current) and the ESR of the capacitors.
To prevent interference between the SMPS switching
frequency and the line frequency an L-C filter has been
added. The inductor is L2 while the capacitor is located on
the line deflection board. If the SMPS is running in the
synchronous mode, the filter action is not required and L2
can be replaced by a 1 Ωresistor. The feedback voltage
for the control circuit is taken in front of this L-C filter.
To prevent cross-talk, the audio supply is brought out
floating. The negative of the 32V supply is connected to
ground via R38 to prevent static charge of this transformer
winding if kept unused. If there is an overload on the 32V
or 16V supply, the currents in the secondary transformer
windings can be excessive before the over-current
protectionofthe ICisactivated. The useofafuseor fusible
resistor (1Ω / 4W) at position J3 and a fusible resistor
(1Ω / 1W)at J4 will make the SMPS short circuit proof also
on these two outputs.
3.7 Error amplifier
The error-amplifier consists of a single transistor T5. The
base of this transistor receives the filtered and divided
outputsignalwhiletheemitterisconnectedtothereference
voltage (ZD3). The output current of this error-amplifier is
fedtotheopto-coupler.AsthecurrentthroughT5andhence
its gain will settle at a value inversely proportional to the
current gain of the opto-coupler, the SMPS loop gain will
be independent of tolerance or ageing of the opto-coupler.
Thecurrent throughZD3isfixedbyR42at2mA.Bykeeping
it constant, the temperature dependence of the Vbe of T5
is compensated by that of ZD3 [3].
3.8 Opto-coupler
Forfeedbackandmainsisolationanopto-coupler(CNG82)
is used. As already mentioned before, the opto-coupler is
driven in such a way, that the large variation of IC/IF of the
opto-coupler is filtered by means of R27 and C19. The
emitter ofthe transistor drives the output-amplifierT2. This
transistor is used for keeping the operating voltage of the
phototransistor constant; this keeps the bandwidth high.
Except for feedback, the opto-coupler is also used for
synchronisation. For this purpose, the sync pulses are
superimposed on the linear feedback signal. The slower
CNX82A can also be used at the expense of the
wave-shape and delay of the sync pulse. Then at 16 kHz
the maximum power will be restricted somewhat due to
unequal duty factors of the odd and even SMPS pulses.
For standby operation, the opto-coupler diode is driven in
forward (IF = 2mA) either by switch-on transistor T6 or by
externally driving resistor R43. In this mode the output
voltages will be switched off.
3.9 Standby supply
For the supply of some digital IC’s in the standby mode, a
small self-oscillating, current-mode controlled flyback
converterdelivers5V/300mA. Itusesthesamemainsinput
filter, bridge rectifier and RFI/safety capacitor C22 as the
main SMPS. For mains isolation and power conversion a
small transformer (AT3006/300) is used. The power
switching transistor BUX87 requires a small heatsink [4].
4. Synchronisation
The SMPS can be synchronised down to 32 kHz by either
16 or 32 kHz (±4%) negative-going pulses on pin 5 of J17,
with a pulse width of 18% of the line time (Eg. line flyback
pulses). The amplitude of the sync-pulse measured at the
sync-input, should lie between 2 and 6V. The sync-pulses
are superimposed on the linear feed back signal. This can
only be done, if they do not affect the voltage stabilisation.
To obtain short rise and fall times of the sync-pulse at the
sync-inputof the TDA8380, theCNG82(A) shouldbe used.
Capacitor C16 ac couples the sync-pulses to pin 11 of the
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TDA8380.Theoscillatorsawtoothistriggeredbythetrailing
edgeofthepositivesync-pulseatpin11andallsubsequent
sync-pulses are ignored until the oscillator sawtooth is
completed. The oscillator is then inhibited until the end of
the next positive sync-pulse. The free-running oscillator
frequency is determined by R11 (4k3) and C15 (680pF).
Both components should be 1% tolerance types.
If synchronisation of the SMPSis not needed,the following
components can be deleted: C19, C35; R29, R27, R49,
R50, R51; D18, R19; T2. The opto-coupler CNG82 canbe
replaced by the CNX82A. Jumpers J1 and J2 should be in
place.
5. Mains interference
RFImeasurementsaremadeoftheSMPS(200W)together
with the µSOPS and a mains input filter, consisting of the
AT4043/93andtwoX-capacitors(220nF). Theresultsmust
stay below the limits of EN55013, which are drawn in the
graphshown later. Themeasurements justmeetthelimits.
If the SMPS is used with more than 165W input power,
measuresmust beincludedto meetthe IEC552-2standard
on mains pollution by higher harmonics.
6. Performance
INPUT 185-265V RMS 50/60Hz
OUTPUTS 150V 1.0A LINE SCAN
STABILISED
32V 1.5A AUDIO
UNSTABILISED
16V 0.2A SMALL SIGNAL
UNSTABILISED
RIPPLE 150V ≤10mV SWITCHING
FREQUENCY
peak to peak ≤20mV 100Hz
32V ≤150mV SWITCHING
FREQUENCY
≤10mV 100Hz
16V ≤50mV SWITCHING
FREQUENCY
≤10mV 100Hz
EFFICIENCY 88% 200W LOAD
SWITCHING 34kHz
FREQ.
7. Oscillograms
The oscillograms have been made at the following
conditions, unless otherwise indicated.
Vinput = 220V RMS Load =200W not synchronised.
Oscillogram 1. Collector Current and Collector Voltage of
the BUW13.
Oscillogram 2. Collector Current and Collector Voltage of
the BUW13.
Oscillogram 3. Base Emitter Voltage and Base Current of
the BUW13.
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Oscillogram 4. Voltage and Current at pin 13 of the
transformer.
Oscillogram 5. Voltage and Current at pin 19 of the
transformer.
Oscillogram 6. SMPS switch on behaviour.
Oscillogram 7. BUW13 Collector Current and Voltage at
short circuit 150V output.
Oscillogram 8. BUW13 Collector Current and 15.625kHz
sync pulses.
References
Information for this section was extracted from
"Synchronous 200W Switched Mode Power Supply for 16
and 32 kHz TV"; ETV89009 by H.Simons.
[1] Integrated SMPS control circuit TDA8380.
Philips Semiconductors Publication Number 9398 358
40011 Date: 12/88.
[2] ETV8711 A 200W switched mode power supply
for 32kHz TV. Author: H.Misdom.
Date: 01/09/87.
[3] ETV89003 Novel optocoupler circuit for the
TDA8380. Author: H.Verhees.
Date: 2/89.
[4] ETV8834 A dual output miniature stand by power
supply. Author: H.Buthker.
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8. Circuit diagram.
Fig.3
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9. RFI measurement
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Monitor Deflection Circuit Example
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4.4.1 A Versatile 30 - 64 kHz Autosync Monitor
Thissection describes thesupply and deflection circuits for
driving the 15" FS M36EDR colour monitor tube. Key
features of this monitor design are auto synchronisation,
full mains range, dc control, good picture stability and high
contrast. The circuit uses a low number of components
without making compromises to the performance.
1. Introduction
Theincreasingnumberofsuppliersofvideointerfacecards,
createsa variety ofvideo standards. The most widelyused
standard at this moment is the VGA mode. Already this
standard gives a choice of three different modes, giving
resolutions between 720x350 to 640x480 pixels. The
horizontal frequency is fixed at 31.5 kHz, while the vertical
frequency varies between 60 and 70 Hz.
New standards, with resolutions up to 800x600 and even
1024x768 pixels, are becoming popular very rapidly,
although the old standards are still present on these new
interface boards.
This increase in resolution calls for larger screens,
compared to the nowadays widely spread standard 14"
tube, dueto the minimum discernible detail at a convenient
viewingdistance. Furthermore, the electronic drivecircuits
of the picture tube have to be able to adapt to the various
standards.
As a successor of the standard 14" high resolution colour
picturetube,the15"FlatSquareM36EDRseriestubesoffer
a noticeable increase in useful screen area
(14": 190x262 mm2; 15": 210x280 mm2), while the total
cabinet size hardly increases, and offering a resolution up
to 1024x768 pixels (pitch is .28mm.). The M36EDR series
offers a wide range of deflection impedances and an
excellent performance with respect to convergence and
geometry distortion, resulting in simple deflection
electronics. Fortheenduser,thetubeoffersaverypleasant
flat and square screen, with hardly any disturbing visible
effects.
To display the new, as well as the old video standards, on
this new tube, the electronic circuits are becoming more
complex. For example, to display 768 lines, without
disturbing screen flicker, the line frequency must be
increased to 57 kHz. The horizontal deflection circuit
described in this report is able to synchronise over a
continuous frequency range from 30 to 64 kHz, while the
vertical frequency may vary between 50 and 110 Hz. The
circuit is built around the advanced monitor deflection
controller TDA4851, significantly reducing the component
count of the total circuit, while providing a high standard of
performance. The vertical deflection output stage is the
TDA4861. This power operational amplifier offers the
designer great flexibility with respect to input signals and
supply voltages.
The increase in resolution also demands that the video
channels are able to drive the picture tube with ever higher
frequencies, due to the increasing amount of pixels on one
video line in a decreasing period of time (increase of line
frequency). Forexample,to display1024pixelsononeline
at 57 kHz line rate, the video amplifiers must be capable of
handling dot frequencies up to 65 MHz.
This report covers the electronic circuits for driving the
horizontal and vertical deflection coils of a 15"FS tube, for
generating all the grid voltages necessary for this tube (the
cathodes are driven by the video amplifiers), and a full
mains range supply, generating the supply voltages.
2. Supply
This autosync monitor is equipped with a full mains range
switchedmodepowersupply(90-265Vac)withamaximum
output power of about 90W1. This mains isolated power
supply is running asynchronous, because of the large
frequency range of the horizontal deflection stage.
To handle the required output power a new wire wound
SMPS transformer, the CE422v, was designed. The
controller IC is the TDA8380 2directly driving the power
switch BUT11A. Feedback is obtained through an
opto-coupler circuit that senses the +155V output, the line
supply voltage. The output voltages of the SMPS are:
+ 155 V @ 350 mA Horizontal deflection and EHT
generation
+ 10.5 V @ 450 mA Vertical deflection
- 10.5 V @ 650 mA Vertical deflection and CRT heater
+ 30 V @ 30 mA Vertical deflection (flyback)
+ 12 V @ 400 mA Video, IC and small signal supply
The +12V supply is derived from the +17V supply rail by
means of a separate voltage stabiliser.
An extrawinding on thetransformer, not used in thiscircuit,
delivers a -17V supply. The rectifier and smoothing
capacitor are not implemented in this design.
As this switched mode power supply is running
asynchronous, additional measures are taken to prevent
interference. All output lines are equipped with LC or RC
filters.
399
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
The +155V rail is protected against short circuit by means
of the TDA8380, while the +12V output is protected by the
IC voltage stabiliser. To make the low voltage outputs
protected against short circuit, fuses or non flammable
resistors are used in the output lines.
2.1 Degaussing
The demo monitor is equipped with a conventional
automatic degaussing circuit, making use of one duo PTC.
Forfull mains range applicationthe inrush andsteady state
current are just on the limit. If support is needed please
contact the local or regional sales/application office.
2.2 Provisions for full-mains range
TomaketheSMPSfullmainsrangethefollowingprovisions
are made:
TheslowstartcapacitorC15isreducedto0.47µFtoshorten
the start up time.
The overcurrent protection is extended in order to ensure
proper working of the IC with respect to the first trip level.
The IC supply capacitor C13 is increased to 220µFto
prevent excessive voltage drop during start up.
3. Deflection
The deflection circuit is greatly simplified by making use of
the advanced monitor deflection controller TDA4851. This
isan upgradedversion of the TDA4850mainly with respect
to horizontal line jitter. To accommodate a horizontal
frequency range from 30 to 64 kHz, the input frequency is
continuously monitored by an F/V converter, in order to
adaptthe centralfrequencyoftheTDA4851. Theminimum
and maximum frequencies of this circuit are limited by an
upper and lower clamp.
Fig. 1 The TDA4851 controller IC.
+12V
+12V
P60
R84*)
P55
R96
R95
R94
R92
R88
R89
P53
R87
R86
R85
R83
C69
C68C67
C66
C65
C63
C62
T51
R82R81
R80
R79
R78
IC52
CURRENT
OSCILLATOR
HORIZONTAL
FLYBACK
HORIZONTAL
DRIVE
HORIZONTAL
OUTPUT
MODE OUT
VERTICALEW
OUT
EHT
COMP.
V-SYNC
H-SYNC
CLAMP PULSE
HEIGHT
HSHIFT
EW adj.
1M*)
33u
220n10n
1n5
8K2
SMD
SMDSMD
SMD
SMD
SMD
220n
120K
100n
BC 548
220n
1K8
1K8
180K
750K
3K9
10K
10E
10K
10K
16V
TDA4851
120K
120K
68K
68K
82K
82K
22K
400
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
The TDA4851 drives a line driver stage and a, so-called,
T-on driver stage. The line driver is of the well known
transformer coupled non-simultaneous type, giving the
designerafreechoicewheretoputthelineoutputtransistor
with respect to the supply voltage. The T-on driver stage
drives a switching transistor, synchronised with the
horizontal frequency to control the horizontal scan voltage.
The conducting period of this switch is kept constant over
the whole frequency range, resulting in a constant picture
width, independent of the scan frequency.
The TDA4851 also drives the vertical output stage
TDA4861. DC coupling of the deflection coil to this
amplifier,together withthehigh linearityofthe drivesignals
from the TDA4851, offer an excellent linear vertical
deflection, without bouncing effects after a mode change.
East-westcorrectionofthehorizontaldeflectionandpicture
width control is performed by the width control stage.
HorizontalS-correctionisperformedbyonefixedcapacitor,
plus a selection of three additional capacitors. This
minimum set-up is chosen to keep the circuit simple, while
giving the minimum amount of distortion at four selected
frequencies. The S-correction capacitor selection circuit
drives floating FET switches.
Vertical S-correction is achieved by modulation of the
vertical amplitude control current with a parabola voltage.
The line output transformer AT2090/01 generates the
anode, focus and grid 2 voltages for the picture tube. The
built-in bleeder with smoothing capacitor provides an
excellent source for retrieving EHT information. This
informationis usedtostabilise thepicture widthandheight.
The primary winding of the line output transformer is also
used as the choke of the switch mode scan control stage.
Thisarchitecture has the advantage ofusing only four wire
wound components: LOT, bridge coil, base drive
transformer and horizontal linearity coil.
Theauxiliarywindingsonthelineoutputtransformerdeliver
supplyvoltagesforthegrid-1circuitandvideooutputstages
and provide information for the protection circuit.
The various circuits will now be discussed in detail in the
next sections.
3.1 Advanced monitor deflection controller
TDA4851
The heart of the deflection circuit is the advanced monitor
deflection controller, the TDA4851, see Fig. 1. This
deflection controller is driven by separate horizontal and
verticalsynchronisationpulses. AlthoughtheTDA4851can
process a sync-on-green signal, this is not implemented in
this monitor. The polarity of these pulses can be chosen
freely, except for VGA modes,their amplitude must be TTL
level. With these pulses, the horizontal and vertical
oscillators are synchronized. The horizontal output drives
the line driver, the vertical drive signals are connected to
theverticaloutputstageIC51. Aparabolavoltagefordriving
the east-west correction stage and a clamping signal for
the video stages are also generated by the TDA4851.
3.1.2 Horizontal part
The horizontal oscillator is synchronized with the pulse on
pin 9: TTL amplitude, positive or negative polarity and
accepting composite sync (sync-on-green is not
implemented in this design). The catching range is limited
to ±6.5%. The oscillator frequency is set by C67 and the
dc current in pin 18, which in this application is set by a dc
current source, driven by the frequency to voltage
converter. Compared to the TDA4850, the current in pin
18 of the TDA4851 is approximately 10 times higher to
achieve a lower phase jitter.
The low-pass filter in the first phase locked loop is
connected to pin 17. In a single frequency application, the
values of the filter components are fixed. For a frequency
rangefrom30to64 kHz,thevaluesofthefilter components
are set by the lowest frequency, resulting in a less than
optimum responsefor the higher frequencies. For reasons
of simplicity, the filter is fixed here, but should be adapted
to the actual frequency for best response.
Fig. 2 Horizontal flyback pulse coupling to the
TDA4851.
The synchronised horizontal oscillator drives the second
phase locked loop, in which the horizontal flyback pulse is
used as feedback to position the horizontal drive pulse in
relation to the horizontal sync pulse. Low-pass filtering is
performed by capacitor C68 connected to pin 20. Phase
shift can be accomplished with a dc current in pin 20.
R105
D57
T54
C74
C73
C72 R104
R103
R102
R101
C61 D54
D53
R77
C60
CLAMPED FLYBACK PULSE OUT
NEGATIVE FLYBACK PULSE IN (1100Vpp)
+12V
680E
33u
100n
10E
16V
PH2369
18P
100K
18P
100K
100K
47P 5K6 C3V3
BZX79
BAW62
BAW62
401
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Nominal phase shift is set with resistor R96. User
adjustable phase control is available through P55,
controlling the dc current in pin 20.
The horizontal flyback pulse is connected to pin 2. The
TDA4851 expectsa positive flyback pulse, see Fig. 2. The
negativeflybackpulsefromthe lineoutputstage isinverted
and ac coupled to pin 2 of the TDA4851.
3.1.3 Vertical part
The vertical oscillator can be synchronized with a pulse on
pin 10 (or combined sync to pin 9) over a range of 50 to
110 Hz without adjustment and with constant amplitude
output signal. Frequency determining elements R89/C63
and the amplitude stabilisation loop capacitor C62, should
not be changed.
The output signals of the vertical part are two balanced
currents, available on pins 5 and 6. Both currents consist
of an equal dc part and an adjustable sawtooth part. The
adjustment is achieved by means of controlling the dc
current in pin 13. There are five signals which determine
the current in pin 13:
1. R87 sets the nominal dc current;
2. P53 allows user control of the picture height via R83;
3. By means of R82, the amplitude control current is
modulatedwithaparabolacurrent. Inthisway,vertical
S-correction is achieved. The disadvantage of this
method, is that the amount of S-correction is
dependent on the setting of the east-west control
potmeter P60.
4. R84 compensates for a change of the east-west
correction voltage (explanation: when the east-west
voltage on pin 11 is changed by means of the ’EW
par.’ potmeter P60, the mean dc voltage on pin 11
changes, influencing the vertical amplitude via R82).
The influence of the east-west setting on the vertical
amplitude is small,therefore, R84 is marked optional,
it’s not absolutely necessary.
5. Changes in the EHT voltage compensate the vertical
amplitude via R80.
3.1.4 East-west parabola
A parabola voltage is available on pin 11, for driving the
pincushion correction stage. The bottom of this parabola
voltage, equal to the middle of the screen, is set internally
on 1.2V, independent of the amplitude setting. In this way,
adjusting the parabola amplitude changes the horizontal
width in the corners only, while theamplitude in the middle
of the screen remains constant.
Amplitude adjustment of the parabola voltage is achieved
by a dc current in pin 14. No user control is available,
adjustment is only possible with P60 on the main
deflection/supply printed circuit board. The amplitude of
theparabola voltage is corrected for changesin the vertical
amplitude setting by means of R85. The amplitude of the
parabola voltage is independent of the vertical scan
frequency.
The parabola voltage from pin 11 is connected to the base
of T51. The collector current is then transferred to the
picturewidthdriver. Inthatstagethe amplitudeismultiplied
with the horizontal frequency, to achieve a correction
independent of the horizontal frequency.
3.5 Miscellaneous I
A clamping signal for the video pre-amplifier, ic TDA4881,
is generated in the TDA4851. This clamping signal is
availableon pin8,and isonlypresentwhenhorizontal sync
pulses are present on pin 9.
Fig. 3 Frequency to Voltage Converter and Current Source.
P59
T52
P54
IC53A
R93
R91
R90
C64
T50
C53
C52C51
C50
IC50
R66
R65
R64
R63
R62
R61
C54
HORIZONTAL
OSCILLATOR
CURRENT
CLAMP
PULSE
F/V adj.
PHI-1 adj.
14
13
12
VFRQ
LM324
1/4
82K
10E
220E
820E
470n
33u
Ser. 82n
100n
683
180K
SMD
PH2369
33K
10K
24K
220P 33K
12p
BC549C
8K2
1K8
NE555
16V
402
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
The mode input/output pin 7 is connected to a switching
transistor T76. This transistor is driven by comparator
IC56B. This comparator detects whether or not the
incoming horizontal frequency is below 33 kHz. In this
case, the TDA4851 assumes a VGA signal is present,
resulting in an automatic adjustment of the vertical
amplitude,dependingon thesync. polarity. Above 33 kHz,
the internal mode detector of the TDA4851 is switched off
with T76, to prevent automatic vertical amplitude
adjustment.
3.2 Additions to the TDA4851 for
auto-sync operation
Since the catching range of the horizontal section of the
TDA4851 is only ±6.5%, the frequency range from 30 to
64 kHz cannot be covered without extra circuits. To
accommodate the specified range, the horizontal oscillator
current (pin 18) is constantly adapted to the incoming
horizontalsyncpulsefrequency. Thisisachievedbymeans
ofafrequencytovoltage converterdrivingacurrentsource.
To protect the power output stages from too low and too
highfrequencies,voltage clampsonthedrivevoltageofthe
current source keep the frequency of the horizontal
oscillator of the TDA4851 within the specified limits of 30
to 64 kHz.
3.2.1 F/V Converter
Thefrequency to voltage converteris built around one-shot
IC50, see Fig. 3. R65/66 attenuate the signal from pin 8 of
the TDA4851 in such a way that only the horizontal clamp
pulses (5.5Vpeak) and the vertical blanking pulses
(1.9Vpeak) do not drive T50. This has the advantage that
the incoming horizontal sync pulse is not disturbed in any
way, and the sync pulse polarity and amplitude variations
of the incomingsync pulse are of noinfluence to thecircuit.
The output pulses of IC50 pin 3 are attenuated and filtered
with R90/91 and C64. The dc voltage VFRQ is used to
drive the current source and the S- correction capacitor
selection circuit. The width of the output pulse of the
one-shotcanbeadjustedwithP59(F/Vadj.). Thispotmeter
should be adjusted in such a way, that the switching of the
S-correction capacitors is performed at the desired
frequencies.
The conversion factor S of this F/V converter is:
S = {tc + 1.1 x (R62+P59) x C51} x {R91 / (R90+R91)} x Uo
where:
tcwidth of the TDA4851 clamp pulse: 1µs
1.1 x (R62 + P59) x C51 width of the output pulse of the
one-shot IC50: 5.81 - 8.23µs
R91 / (R90 + R91) attenuation of the output pulse: 0.313
Uoamplitude of the output pulse: 10.8V
The duration of the output pulse of the one-shot is affected
by the trigger pulse. During time tcthe output voltage is
already high, while the charging of C51 is halted. This
results in:
S = 23.0 to 31.2 mV / kHz
The voltageVFRQ can be foundwith the followingformula:
VFRQ = FH x S
where FHis the horizontal scan frequency. With S =
27.1mV/kHz, this results in:
VFRQ(31.47 kHz) = 31470 x 27.1 x 10-3
VFRQ(31.47 kHz) = 853 mV
and:
VFRQ(63.69 kHz) = 63690 x 27.1 x 10-3
VFRQ(63.69 kHz) = 1726 mV
Because the resistor divider for the s-correction capacitors
is fixed, the conversion factor S is adjusted to the voltage
VREF, feeding the divider.
3.2.2 Current source
The current source for driving pin 18 of the TDA4851 is
build around op. amp. IC53A and T52.
The collector current of T52 can be adjusted with P54 to
adapttotheactualconversionfactorSandforthetolerance
onthe oscillatorcapacitorC67 on pin19 of IC52. Thismust
be done after the conversion factor S is adjusted with P59
to the correct s-correction switching frequency.
3.2.3 Voltage clamps
Fig. 4 Voltage clamps.
D83
IC53C
R179
R178
R164
IC53D
C106
R154
D82
VREF
VFRQ
5
67
1
3
2
LM324
1/4
LM324
1/4
910E
91E
220E
1K1
100n
BAW62
BAW62
403
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
To prevent the horizontal power output stage from running
at either too low, or too high, a frequency,the drive voltage
for the current source is limited. IC53C limits the lower
voltage and IC53D the upper voltage, see Fig.4. The
frequency range of the TDA4851 is now limited to 30 kHz
minimum and 66 kHz maximum.
Referenceforthe comparatorsIC53C/Disa voltagedivider
network, driven by IC53B. The input voltage for IC53B is
the temperature compensated reference voltage at pin 15
of the TDA4851.
3.3 Horizontal scan control driver
In order to keepthe picture width constant, independent of
the horizontal scan frequency, the scan voltage is
continuously adapted. The relation:
Vscan = L x I x fH
is valid. From this equation, it can be seen that with
increasing scan frequency, the supply voltage must also
increase proportionally.
Realisation of this demand is in fact quite simple. The
horizontal drive pulse from the TDA4851 triggers one-shot
IC54. The output pulse width of this one-shot is constant,
independent of the trigger frequency. Therefore, the
duty-cycle increases with increasing frequency. Via buffer
stage T62/63, FET switch T64 is controlled. In this way,
the scan voltage for the horizontal deflection output stage
is controlled, according to the previously stated relation.
Fig. 5 Horizontal Scan Control Driver.
IC54 is a retriggerable one-shot, see Fig. 5, which is
important at higher scan frequencies (above 66 kHz),
where the duty-cycle becomes 1.0. In case of a
non-retriggerableone-shot, theduty-cyclewould suddenly
drop to 0.5. This would not only result in half the deflection
amplitude, but also in a drop of the EHT.
One other important item is the phase relation between the
drive pulses of T53 and T64. When the horizontal output
stage is in the flyback part, T64 must always conduct (in
order to keep the EHT constant). This is realised by
choosingthe appropriate trigger edge (positiveedge of the
horizontaldrivepulse)andbyalowerlimitoftheadjustment
range of the pulse width, to ensure T64 is conducting all
throughtheflybackperiodofthehorizontaldeflectionstage.
3.4 S-correction capacitor selection
The necessary value of the S-correction capacitor varies
withthe horizontal frequency,given a certaindeflection coil
impedance and screenradius. The correct capacitor value
ischosenfromeightdifferent valuesthroughacombination
of one fixed and a choice of three capacitors.
The voltage VFRQ, representing the horizontal scan
frequency, is connected to the inverting inputs of seven
comparators IC56/57. Each non-inverting input of these
comparators is connected to a different output of a resistor
ladder R154/164-170/178/179. This resistor ladder is fed
by a voltage VREF.
At pin 15 of IC52 a temperature compensated voltage of,
typically, 3.0V is available for setting the vertical oscillator
current (R89). To avoid extra loading of this pin, a voltage
follower with high input impedance IC53b is used. The
output of op. amp. IC53b is a stable dc voltage with very
low output impedance: VREF.
Resistors R171 to R177 provide each comparator with
some hysteresis to prevent parasitic oscillations on the
switch-over points.
Thecomparatoroutputsareconnectedtoan8-3multiplexer
IC58. The outputs of IC58 drive transistors T73/74/75.
These transistors can withstand the possible high voltages
(max. 150V) that drive the S-correction capacitor switches.
D79/80/81 are added to prevent T73/74/75 from break
down during line flyback. In this way, the selection of the
resistors sets the frequencies when the circuit switches to
another S-correction capacitor value.
3.5 Picture width driver
Since pin-cushion distortion is a fixed percentage of the
scanvoltage, the peakto peakparabola voltage,correcting
the pin-cushion distortion, must be adapted to the actual
scanfrequency. Thismultiplication isachievedinthe same
manner as the scan voltage for horizontal deflection is
adapted.
One-shot IC55, see Fig. 7, is triggered with pulses having
the same frequency as the horizontal scan frequency. But
now, also the parabola voltage modulates the width of the
output pulse. This output pulse is integrated through
R138/C98; the voltage drives a common base stage T69.
+12V
C87
R123
T63
T62
R122
R121
C86
C85
P56
R120
IC54
PULSE
H-DRIVE
T64
TO GATE
EHT adj.
+10.5V
C15
47K
1K
100n
220p
Ser
683
4538
HEF
33u
22k
10E
BC558
BC548
22E
16V
404
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Fig. 6 S-Correction Capacitor Selection Circuit.
+12V
+12V
R179
R178
IC58
IC56D
IC56A
IC57B
IC56C
IC57D
IC57C
IC57A
R177
R170
R176
R175
R174
R169
R168
R167
R166
R173
R172
R171
R165
R164
R157 up to R163
C108
C107
R156
R155
C106
R154
R153R152
R151
T75T74T73
D79 D80 D81
TO S-CORR. CAPACITOR SWITCHES
IC57 PIN12
IC56 PIN12
IC57 PIN3
IC56 PIN3
IC53 PIN11
IC53 PIN4
VREF
VFRQ
13
11
10
14
9
8
1
7
6
2
5
4
13
11
10
14
9
8
1
7
6
910E
91E
39E
470K
470K
470K
39E
240E
150E 470K
470K
1/4
1/4
1/4
1/4
1/4
1/4
1/4
LM339
LM339
LM339
LM339
LM339
LM339
LM339
300E
200E
220E
33u
33u
470K
1K1
100n
22K
HEF4532
22K 22K
BAW62BAW62
BAW62
MPSA42
MPSA42MPSA42
470K
7X 10K
16V
10E
10E
16V
405
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
In this way, the modulation is converted to the collector
current of T69 (IEW), simplifying the interface to the
east-west power stage, which is connected to the +155V
supply.
Two other functions are also incorporated in this part of the
circuit:
- User control of the picture width via P58; and
- Compensation for variation of the EHT; achieved
through R136 - the information is supplied by the EHT
compensation circuit.
Fig. 7 Picture Width Driver.
3.6 Horizontal deflection output stage
To allow a frequency range of 30 to 64 kHz, additional
measures have to be taken to keep the deflection current
and the EHT constant. This is realised by continuously
adapting the scan voltage to the horizontal frequency by
means of the horizontal scan control. The scan control
switch T64 is connected to ground, resulting in simple gate
drive. The horizontal deflection output transistor T55 is
connected to thesupply rail because the drivertransformer
L50 already provides isolation. The primary winding of the
Line Output Transformer L54 is used as a choke.
Thisset-up of horizontal deflection has twodisadvantages:
1. The pincushion correction stage is either floating or
connected to the same supply rail as the line output
transistor,T55. Inthisconcept,connectiontothesupply
rail is chosen, and driving it with a current IEW from the
small-signal circuit.
2. The S-correction capacitor switches are floating, which
makes it less simple to drive them.
The biggest advantage of this deflection stage is that a
minimum of wire wound components has been used.
3.6.1 Line driver stage
The circuit is shown in Fig. 8. As driver device T53 a small
MOSFET (BSN274) is chosen. Its advantages over a
bipolar device onthis particular application are: less power
dissipation (enabling smaller encapsulation), less drive
power required, better switching behaviour and no storage
time (so no additional stress on the Φ2loop). To protect
the gate the standard precautions (zener diode and series
resistor) are taken.
The driver transformer is equipped with a damper network
at the primary side (C71/R98) to damp excessive ringing.
On the secondary side a damper (C76/R107) is present
between the base and emitter of the deflection transistor
T55.
In order to achieve sufficient negative drive voltage during
flyback, resistor R106 and diode D58 are added. Proper
-dIb/dT is achieved by the leakage inductance of L50.
When the X-ray protection is activated, the inverting driver
stage will be turned on continuously. This will switch off
T55, but also cause a low frequency swing in the driver
transformer. To prevent voltage inversion across the
primarywindingofdrivertransformer L50,whichwouldturn
on the deflection transistor T55 for a relatively long time,
D55 is added. A second measure, that must be
implemented when X-ray protection is installed, is
increasing the power rating of the current source resistor
R97 to 16W (!) or usingan electronic resistor with a current
fold back characteristic.
Fig. 8 Line Driver Stage.
3.6.2 Horizontal scan control output stage
At double the line frequency, the scan voltage must be
doubled as well to have the same picture width on the
screen. Furthermore, the supply voltage to the line output
transformer must be proportional to the horizontal line
frequency to have constant EHT over the whole frequency
range. To achieve this, a synchronous switching series
regulator is added. This series regulator operates with a
constant T-on time.
+12V
+12V
C94
IC55
T69
R141
R140
C98
P58
C97
C96
C95
R139
R138
R137
R136
R135
R134
IC50 PIN 3TRIG. INPUT
EHT COMP.EAST-WEST IEW
+155V
WIDTH
22n
220n
Ser.
683
220p 33u 82n
NE555
SMD
390K
330E
470E
10K
5K6
16V
10E
MPSA42
56K
1K
10K
21
34
C76
L50 D58
D56
D55 T55
T53
C75
C71
C70
R107
R106
R100
R99
R98
R97
TO DEFLECTION CIRCUIT
LINE DRIVE PULSES FROM TDA4851 PIN 3+12V
+155V
C15
1K5 PR03
1K5
BU2520A
2.7E AC04
68nBAS11
250V
150u
1K
BSN274
BYD33D
22E
BZX79
250V
2E2
220n
AT4043/87
150p
406
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
InFig. 9thebasic circuitdiagramandinFig. 10therelevant
waveforms are given.
Fig. 9 T-on Switch.
Transistor T64 controls the horizontal supply voltage and,
hence, the peak value of the flyback pulse which is directly
related to the horizontal amplitude and EHT (flyback time
is fixed, independent of frequency).
Fig. 10 Horizontal Scan Control Waveforms.
The average voltage across a coil must be equal to zero.
With T-on = 100% the area under the flyback pulse must
be equal to the scan amount. At half frequency, for equal
EHT with fixed flyback time, half scan amount will be
sufficient. This is indicated in Fig. 10. During T-off the
current will flywheel in D73.
63kHz
31.5kHz
Vlot
Vlot
Tperiod
Ton
Tfb
Ilot
L52 R124
C89
C88
D73
D72
R123
T64
C15
BUK455-200B
120E
BYV99 250V
22n
2n7
22E
BZX79
12uH
TO LOT+155 V
BUFFER
FROM
Fig. 11 Power Output Stage.
C99
R125
L53
C80C79
L51
C78
C77
C76
D61
D60
D59
T57
T56
T55
C75 R109
R108
R107
BASE DRIVE
+155 V
+155 V
1200 V
IEW
+HDEFL
-HDEFL
5n6
220n
1K PR02
15n
BU2520A BYW96D
BYW96D
5u6
BC547C
250V
150u
AT4042/33A
10K
BD648
1600V
100V
400V
BYW95C
2KV
1E 25V
47u
2E2
220n
AT4043/13
407
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
3.6.3 Power output stage
Thepower output stage, seeFig. 11, isa conventional one
with a diode modulator.
Because this stage is connected to the supply rail, the
flyback pulse has negative polarity. This makes the use of
T54, see Fig. 2, necessary because the deflection
controller IC52 expects a positive flyback pulse.
As lower damper diodes two low voltage diodes BYW96D
in series are chosen because they switch faster than one
single high voltage device.
As deflection transistor, the BU2520A is chosen. This
device performs remarkably well over the frequency range
of 30 to 64 kHz.
The value of flyback capacitor C78 depends on the
impedance of the line deflection coils and the desired
flyback time. With 180µH deflection coil impedance, C78
shouldbe 5n6/2kV,with 220µH impedance, C78 should be
4n7/2kV.
3.6.4 East-west power stage
To drive the diode modulator, the drive current "IEW" must
be converted to a voltage by means of R109. A power
buffer T56/57, see Fig. 12, drives the diode modulator. To
prevent high ac currents from flowing through T56, an
additional filter R108/C80 is added.
Fig. 12 EW Power Stage.
3.6.5 S-correction capacitor switches
The curvature of the screen determines the percentage
S-correction. Thispercentageisconstantandindependent
of frequency. Since the scan voltage is adapted according
to the horizontal frequency, the S-correction voltage also
has to be adapted, according to the frequency. The value
of the S-correction capacitor is determined with the
following equation:Cs = Tp2 / (8 x σ x Lh)
where:
Tp= the visible line period time;
σ= the percentage of S-correction;
Lh= the impedance of the deflection coil.
With a constant flyback time of 3µs, this gives the results
asshowninFig. 14. Thesevaluesarerealisedinthe circuit
shown in Fig. 13.
C80C79
L51
T57
T56 R109
R108
+155 V
TO DEFLECTION CIRCUIT
IEW
5u6
BC547C
10K
BD648
100V
1E 25V
47u
AT4043/13
Fig. 13 S-correction Capacitor Switches.
C100
C99
C105
C104
C103
C102
C101 D78
T72 R150
R149
R148
T71
D77
R147
R146
R145
D76
T70 R144
R143
R142
C80C79
L51
T57
T56 R109
R108
TO DEFLECTION COIL
+155 V
DRIVERS
TO SWITCH
IEW
C15C15C15
BUK455-
150K
47K
100K
120n
BUK455-
150K
47K
100K
330n
BUK455-
150K
47K
100K
820n
220n
5u6
BC547C
630V
250V
10K
BD648
100V
250V
400V
22n 22n 22n
200B
200B 200B
1E 25V
47u
BZX79BZX79BZX79
AT4043/13
408
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Fig. 14 S-correction Capacitor in Relation to the
Horizontal Frequency.
Each switch contains a MOSFET with built-in anti parallel
diode, see Fig. 14. When, for instance, T75, see Fig. 5, is
notconducting,C101willbechargedviaR143/144andT70
willconduct. D76 prevents the gate fromtoo high voltages.
When T75 conducts, the voltage at C101 will be zero and
T70willblock. Suchaswitchcanalsobebuildwithabipolar
device, however that would require a higher drive current,
resulting in high losses because of the ac and dc voltage
difference between drive and switch.
The switch itself functions as follows. During the first part
of scan the current is conducted by the MOSFET; the
S-correction capacitor will be charged. During the second
partofscanthe currentwillbeconductedbytheantiparallel
diode. In case the MOSFET is not conducting, the
S-correction capacitor will not be charged during first part
of scan, except from a very small current through the
resistors parallel to the MOSFET switches. So, during the
secondpart of the scanthe VDS will remain positive and the
anti-parallel diode will not conduct.
3.6.6 EHT, Focus and Vg2
The EHT, focus and Vg2 are generated by the Line Output
Transformer (LOT) AT2090/01. This transformer can be
used up to 85 kHz and has a built-in bleeder (with focus
and Vg2 potentiometers) and an EHT smoothing capacitor
of 3nF. Not only the flyback but also the scan voltages are
frequency independent. So, auxiliary voltages can be
extracted from the LOT in the ordinary way. There is one
exception: often the heater voltage for the CRT is taken
from an unrectified winding of the line output transformer.
Since due to T-on the RMS value is not frequency
independent, the CRT heater must be supplied from a
rectified winding. For practical reasons in this design an
SMPS voltage was more suited (-10.5V).
Fig. 15 EHT vs. Frequency.
3.7 Vertical deflection
The vertical deflection output stage used in this design is
the TDA4861(IC51), see Fig.16. This vertical output stage
can be considered as a power operational amplifier with an
extra flyback generator and guard circuit.
The inputs are driven by the balanced outputs of the
TDA4851. The TDA4851 supplies complementary drive
currents, which can be directly connected to the input pins
of the output stage.
To determine the values of resistors R71/75, conventional
operational amplifier theory is applicable. This theory says
that, in practical cases, the differential input voltage of an
operational amplifier always equals zero:
V2 = V3
Iout x R72 - Idrive x R71 = Idrive x R75
For simplicity of design,R71 and R75 have the same value
Ri:
Ri = Iout x R72 / (2 x Idrive)
Thepeakoutputcurrentisthepeakcurrentforthedeflection
coil used in the design (here peak Iout = 0.75A), Idrive is
the drive current from the TDA4851: 250µA. The value of
resistor R72 can be chosen freely within certain limits:
1. The power dissipation in the resistor may not exceed
the power rating of the resistor used;
2. ThevoltagedropacrossR72issubtracted fromthetotal
available peak to peak coil drive voltage;
3. The minimum resistance is limited bythe ground plane,
which introduces a tolerance that has to be minimised
with respect to the resistor value.
A good choice for R72 is 1Ω, the lowest available resistor
valueinthenormalrange. Thisleadstothefollowing result:
Ri = 0.75 x 1 / (2 x 250 x 10-6)
Ri = 1500 Ω
Amplitude control is realised by adjustment of the output of
the TDA4851.
0.8
0.6
70.0
64.0
60.0
56.748.0
35.5
37.831.5
0.1
0.15
0.2
0.3
0.4
1.0
1.5
2.0
3.0
4.0
FREQUENCY (kHz)
M36EDR311X170
M34EDC13X16
*
*
*
*
(uF)
E
C
N
A
T
I
C
A
P
A
C
Ibeam=0.1mA
X
XX
X
X
XX
X
EHT
[kV]
25
24
23
f [kHz]
60504030
Ibeam=0.0mA
X
XX
X
X
XX
Ibeam=0.4mA
X
XX
409
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Vertical shift, a user control, is possible by P52. Injecting
a dc current in one of the summation points, results in a dc
current through the deflection coil. The 0 to +12V from the
potmeterP52istranslatedintoadccurrentinR71bymeans
of resistors R73/74. Design considerations for these two
resistors are: a potmeter in middle position (the dc current
in the coil should be zero) and the maximum shift range.
Theresistorsvaluesinthecircuitdiagramallowashiftrange
of ±15 mm.
TheoutputoftheTDA4861isDCcoupledwiththedeflection
coil,resultinginabounce-freebehaviour. Togetherwiththe
fast response of the TDA4851 after a frequency change,
this combination offers a stable picture within two frames.
For stability reasons, the combination R70/C56 is added
between the output and the most negative supply voltage.
Ifno damping resistor is presenton the deflection coil,R69
should be added. Its value has to be determined
experimentally.
The supplyvoltages for theTDA4861 are ± 10.5V,allowing
simple dc coupling of the deflection coil. For flyback, an
extra supply voltage of +30V is connected to pin 8. This
results in a fast flyback of 300µs.
The vertical guard pulse, available on pin 9, is connected
totheVg1 circuit toprovidevertical blankingandprotection
in case no deflection coil is connected.
Diode D52 protects the TDA4861 in case the flyback
voltage is missing or drops faster than the +10.5V at
switching off of the circuit.
Fig. 16 Vertical Deflection Output Stage.
3.8 Miscellaneous II
3.8.1 EHT compensation
With the aid of the built-in EHT capacitor and focus / Vg2
divider in the LOT, the EHT voltage can be monitored in a
very easy way. When the time constant of these built-in
components is equal to the time constant of
(R128+P57)/C92, then at LOT pin 12 an exact (divided)
copy of the EHT can be found. This signal is buffered by
T66 and inverted by T67, see Fig. 17.
Due to the tolerance on the Focus / Vg2 bleeder an
adjustment is required (P57).
TheEHT informationsignalgoestoT68,theinvertedsignal
modulates the picture width driver, in order to compensate
the horizontal deflection for EHT variations. The
non-inverted output of T68 is fed to the vertical amplitude
control pin of the TDA4851 to compensate the vertical
deflection for EHT variations.
Fig. 17 EHT Compensation.
3.8.2 Beam current limiting
The long-term average anode current for the given tube is
700µA. This current is measured at the lower side of the
EHT winding of the LOT, L54. The anode current flows
through resistor R126, connected to the +12V rail. When
the voltage across this resistor increases (with increasing
anode current), the base voltage of T65 drops. With a high
contrast setting, 6V dc on pin 9 of connector 2, the beam
current limiter (BCL) will be activated at an average anode
current of:
Ia = { +Uv - Ucontrast + Ube(T65) + U(D74) } / R126 - Ibleeder
C93
R133R132
T68
P57
C92 D75 R131
R130
R129
T67
T66
R128
L54
TO VERTICAL AMPLITUDE ADJ.
TO PICTURE WIDTH MODULATOR
EHT comp.
Vg2
FOC
EHT
+155V
8
11
12
3
10
9
4
5
6
1
2
C24
BC548
1n5
BZX79
470n
560E
22K
2M2
10K
1K
MPSA92
BC549C
63V
1M
1M8
AT2090/01
+12V +30V
-10.5V
+10.5V
R69*)
IC51
R76
R75
R74
R73
R72
R71
P52
9
D52 R70
1
CON 6
51
C59
C58
C57
C56
OUTPUT
GUARD
FROM TDA4851
CURRENT INPUT
To DEFL. unit
+VDEFL
-VDEFL
V-shift
+HDEFL
-HDEFL
330E*)
TDA4861
15u
68u
68u
5E6
47n
1K5
1E
1K5
BYD31D
40V
16V
16V
8K2
82K
10K
10K
410
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Ia = { ( 12 - 6 + 0.65 + 0.5 ) / 15,000 }
- { 25,000 / ( 300 x 106 ) }
Ia = 394 µA
3.8.3 Vg1 supply
This monitor is equipped with an ac coupled video output
stage, using a supply voltage of 65V. After dc restoration,
the highest black level is approximately 45V, with respect
to ground. This implies that, with a tube requiring a cut-off
of 125V, Vg1 must be -80V.
At C84a negative flyback pulse isrectified (-130V). During
normal operation T58 is saturated and Vg1 will be -80V. If
T58 is not conducting, Vg1 will be -130V which will cut-off
the tube completely. This will be the case in the following
conditions:
vertical guard (failure in the vertical output stage;
absence of vertical supply (+10.5V); and
absenceof horizontal deflection (e.g.power switch off).
Vertical guard. When the vertical output stage generates
a vertical guard pulse, via D64 the base of T58 will become
high, which will turn off this transistor. Vg1 will be -130V.
Absence of vertical supply. When there is no vertical
supply, the vertical output stage can not generate a guard
pulse either. Therefore, the Vg1 circuit is connected to the
vertical supply rail. When the +10.5V supply is missing,
T58 cannot conduct, resulting in Vg1 = -130V.
Fig. 18 Vg1 Supply.
Absence of horizontal deflection. When there is no
horizontal deflection the line flyback pulse will be small or
not present at all. This line flyback pulse is peak-peak
rectified at C81 and thus keeping T59 blocked. When
flyback pulses disappear, caused by a failure in the Line
OutputStageoratswitch-off,T59will conduct,causingT58
to be blocked, and Vg1 will be -130V (C84 is large enough
to hold Vg1 on -130 Volts until the EHT is discharged).
3.8.4 Blanking for TDA4881
Horizontalblanking pulses arederived from theline flyback
pulses as delivered by the circuit around T54, see Fig. 2.
The cathode of D51 is connected to the collector of T54.
Tolimit the amplitude of theblanking pulses, D50 is added,
see Fig. 19.
Fig. 19 Blanking Pulse Generator.
3.8.5 Video supply
The ac coupled video output amplifiers require a supply
voltage of:
Vs = Vswing + Vmin + ( Vs - Vmax )
Vs = 50 + 10 + 5 V
Vs = 65 V
The secondary windings 3-4 and 5-6 of the LOT, L54, are
connectedinseriesandstackedonthe+10.5Vsupply. The
output voltage of rectifier diode D65 and capacitor C83 is
66V.
3.8.6 X-ray protection
A failure in the horizontal scan control section, could cause
a dangerous situation: the EHT might rise to an
unacceptable high level. The thyristor, consisting of
T60/61, see Fig. 20, is fired when the flyback voltage rises
to an unacceptable level. The flyback input pin 2 of the
TDA4851 is forced high. This causes the horizontal drive
output pin3 of theTDA4851 to beturned off (outputvoltage
is high). The line driver will be turned on, turning off the
line output transistor. The T-on driver will not be triggered
any more. The result is that the complete line output stage
stops working, so that the EHT will drop automatically.
Blanking is achieved, through the normal blanking circuit.
Furthermore, the Vg1 voltage will also drop, in order to
cut-off the tube.
+12V
D51
D50
R68
R67
C55
100n
10E
C5V6
BZX79
100V
BAW62
2K7
OUTPUT
BLANKING
T54
COLL.
R117
R116
C84
D71
D67
D66
C81
T59
T58
D64
D63D62
R113
R112
R111
R110
PIN 9 LOT
PIN 6 LOT
GUARD
VERTICAL
+10.5V
Vg1
C39
C39 10K
27K
10K 82K
1n5
C8V2
BZX79
15K
BC558
1K
BAW62
BYD31J
250V
BF423
BZX79
2u2
BYD31J
BZX79
411
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Fig. 20 X-ray Protection.
4. Oscillograms
The oscillograms given are meant as a guide-line in
debugging and aligning the circuit and together with the
abovetextitcanalsobeofhelpinunderstandingthecircuit.
All oscillogramsconcerning the horizontal syncprocessing
and deflection are given at two frequencies (31.5 &
56.7 kHz).
The relative position of the traces in the oscillograms with
respectto groundisnotgiven. It isassumedthat thereader
has enough knowledge of the circuits to understand them
without this indication.
In all the following figures trace 1 is at the top leading to
trace 4 at the bottom of each oscillogram.
R118
R119
R117
C84
D70D69
D68
C82
T61
T60
R115
R114
TO TDA4851 PIN 2
+12V
Vg1
200V
4u7
82K
C56C56C56
1K
1n
1K
BZX79 BZX79 BZX79
1K
1K BC548
BC558
31.5 kHz 56.7 kHz
Figs. 21 & 22 Horizontal Oscillator
Trace 1: H-sync pulse at pin 9 of TDA4851 (2V/div). Remark: Whenthesensitivityof trace3is enlargedsmall
Trace2: Sawtoothvoltage atpin19 ofTDA4851(2V/div). triangles can be seen (100 - 150mV) pointing
Trace 3: Φ1 voltage at pin 17 of TDA4851 (5V/div). downwards. These triangles coincide withthe horizontal
Trace 4: Drive voltage at pin 3 of TDA4851 (2V/div). syncpulse. Soawidersyncpulsegiveswider(andlarger)
Horizontal: 5µs/div. triangles.
412
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
31.5 kHz 56.7 kHz
Figs. 23 & 24 Frequency to Voltage Converter
Trace 1: H-sync pulse at pin 9 of TDA4851 (2V/div). Remark: The output voltage of pin 3 is integrated and is
Trace 2: Voltage at pin 2 of IC50 (5V/div). usedfor driving thehorizontal oscillator and S-correction
Trace 3: Output voltage at pin 3 of IC50 (5V/div). switches.
Horizontal: 5µs/div.
31.5 kHz 56.7 kHz
Figs. 25 & 26 Horizontal Driver
Trace 1: H drive pulse at pin 3 of TDA4851 (5V/div).
Trace 2: Drain voltage at the driver MOSFET BSN274 (200V/div).
Trace 3: Emitter voltage of the deflection transistor (500V/div).
Trace 4: Current in the base of the deflection transistor (2A/div).
Horizontal: 5µs/div.
413
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
31.5 kHz 56.7 kHz
Figs. 27 & 28 LOT & Deflection Transistor
Trace 1: Emitter voltage of the deflection transistor (500V/div).
Trace 2: Collector current of the deflection transistor (2A/div).
Trace 3: Primary current in the LOT (0.5A/div).
Trace 4: Primary voltage of the LOT (500V/div).
Horizontal: 5µs/div.
Remark: The switching of T64 causes ringing in the LOT. This ringing is suppressed by the damping network in series
with the LOT (R124, C88, L52). Some small ringing remains and is visible in the oscillograms.
31.5 kHz 56.7 kHz
Figs. 29 & 30 Ton Switch
Trace 1: Primary voltage LOT (500V/div).
Trace 2: Drain voltage T-on switch T64 (200V/div).
Trace 3: Drain current T-on switch T64 (0.5A/div).
Trace 4: Current in D73 (0.5A/div).
Horizontal: 5µs/div.
414
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
31.5 kHz 56.7 kHz
Figs. 31 & 32 Drive Ton Switch
Trace 1: Pin 4 IC54 (5V/div). Remark: At the negative edge of the drive pulse IC50 is
Trace 2: Pin 6 IC54 (5V/div). triggered. The output will give a fixed T-on (high) output.
Trace 3: Drain T64 (100V/div). To prevent this one shot working at half frequency
Horizontal: 5µs/div. (fdrive > 1/Ton), this one shot must be retriggerable.
31.5 kHz 56.7 kHz
Figs. 33 & 34 Deflection Stage
Trace 1: Emitter voltage T55 (500V/div)/
Trace 2: Voltage at pin 2 TDA4851 (2V/div).
Trace 3: Deflection current (5A/div).
Trace 4: Current in bridge coil L51 (5A/div).
Horizontal: 5µs/div).
415
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
31.5 kHz 56.7 kHz
Figs. 35 & 36 S-correction switches
Trace 1: Emitter voltage T55 (500V/div).
Trace 2: Anode voltage D61 (100V/div).
Trace 3: Drain voltage S-correction switch T70/71/72 when on (100V/div).
Trace 4: Drain voltage S-correction switch T70/71/72 when off (100V/div).
Horizontal: 5µs/div.
Remark: At 31.5 kHz all S-correction switches are conducting so in Fig. 35 trace 4 is missing.
31.5 kHz 56.7 kHz
Figs. 37 & 38 EW One Shot
Trace 1: Voltage at pin 2 IC55 (5V/div).
Trace 2: Voltage at pin 6 IC55 with max. pulse width (5V/div).
Trace 3: Voltage at pin 6 IC55 with min. pulse width (5V/div).
Horizontal: 5µs/div.
Remark: Pin 5 of IC55 is modulated with the EW parabola, see Fig. 43. The output pulse width is thus EW modulated
with the min/max pulse width given in traces 2 and 3.
416
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Figs. 39 & 40 Vg1 Switch Off
Trace 1: Emitter voltage T55 (500V/div). Remark: The time constant of the picture tube and EHT
Trace 2: Vg1 (50V/div). capacitor/bleeder is about 1.6s, so Vg1 is still -130V as
Horizontal left: 20µs/div. the picture tube is fully discharged. Vg1 will also blank
Horizontal right: 1s/div. the screen in case of vertical guard and failure of the
+10.5V supply.
Trace 1: Vertical deflection current (1A/div).
Trace 2: Sync pulse at pin 10 TDA4851 (2V/div).
Trace 3: Vertical oscillator voltage pin 16 TDA4851
(2V/div).
Trace 4: Output voltage at pin 5 TDA4861 (20V/div).
Horizontal: 2ms/div.
Remark: During vertical sync pin 8 will be 2V (not visible
in this oscillogram due to sampling effect of the DSO
used).
Fig. 41 Vertical Deflection
417
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Trace 1: Vertical sync pulse at pin 10TDA4851 (2V/div).
Trace 2: Voltage at pin 8 TDA4851 (2V/div).
Trace 3: Guard voltage at pin 9 TDA4861 (5V/div).
Trace 4: Vg1 (50V/div).
Horizontal: 2ms/div.
Fig. 42 Blanking and Guard
Trace 1: Vertical deflection current (1A/div).
Trace 2: EW output (pin 11) of TDA4851 (1V/div).
Trace 3: Pin 5 IC55 (1V/div).
Trace 4: Filtered voltage of pin 3 IC55 (0.5V/div).
Horizontal: 2ms/div.
Remark: Inthemiddle ofthe picturetubea whitebarwith
highintensitywasdisplayed. Attrace3thesuperimposed
EHTcompensationsignalfor picturewidthcorrectioncan
be seen. At trace 4 some line ripple is visible, see
Figs. 37&38.
Fig. 43 EW waveforms
418
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
Trace 1: VCE of the SMPS transistor (500V/div).
Trace 2: VBE of SMPS transistor (5V/div).
Trace 3: ICof SMPS transistor (1A/div).
Horizontal: 10µs/div.
Fig. 44 Power supply
5. Component Placement
The following recommendations are given for the design of
the PCB (see also reference).
1. Keep loop areas with high currents and sensitive loop
areas as small as possible.
2. Keep tracks that carry high voltage components and
sensitive tracks as short as possible.
3. Do not locate the asynchronous SMPS transformer
close to the TDA4850.
4. Use a star ground without ground loops!
5. Implement local supply filtering for an IC. On the
ground only peripheral components of this particular
IC may be grounded.
6. Try to ground sensitive components as close as
possible to the ground pin of its IC using a separate
ground track; for example, components of oscillator,
Φ1 and Φ2.
7. Especially critical in this circuit are the components
around the TDA4851 (IC52), belonging to the
horizontal part. Where possible, SMD types should
be used. In all other cases, connecting tracks should
be kept as short as possible.
Fig. 45 Examples of SMD component placement
+
+
+
PIN 1
IC52:TDA4851
R92
8K2
220n
C65 1n5
C66 220n
10n
C67
C68
120K
R96
PIN 1
IC50
C50
82n IC55
82n
C96 PIN 1
419
Televisions and Monitors Power Semiconductor Applications
Philips Semiconductors
8. IC50 and IC55, both NE555 timers, should be fitted
with an SMD supply bypass capacitor connected
directly acrossthe supply pins. The reason for this is,
to keep the current transient at switch-over of the
output as small as possible.
9. Thepulses from the F/Vconverter, IC50 inFig. 3, and
the picture width driver, IC55 in Fig. 7, must not
interfere with the sawtooth voltage of the horizontal
oscillator, IC52 in Fig. 1.
Examples of good layout solutions with SMD components
are shown in Fig. 45.
6. References
The information in this section has been extracted from the
following report:
A Versatile 30 - 64 kHz Autosync Monitor
Author: H.Misdom / H.Verhees
Report no.: ETV92003
12NC:
For a complete understanding of this application leading to
actual implementation of this design the above report
should be consulted. Other essential reference sources
are as follows:
Improvements on the 30 - 64 kHz Autosync Monitor
Author: H.Verhees
Report no.: ETV92008
12NC:
Full Mains Range 150W SMPS for TV and Monitors
Author: H.Simons
Report no.: ETV/AN92011
12NC.:
Advanced Monitor Deflection Controllers TDA4851
and TDA4852
Author: H.Verhees
Report no.: ETV93003
12NC:
Integrated SMPS Control Circuit TDA8380
Author:
Report no.:
12NC: 9398 358 40011
Specification of Bus Controlled Monitor
Author: J.Shy, T.H.Wu and J.Chiou
Report no.: Taiwan/AN9101
12NC:
Improvements on the 30 to 64 kHz Autosync Monitor
Author: H.Verhees
Report no.: ETV92008
12NC:
Electromagnetic Compatibility and PCB Constraints
Author: M.J.Coenen
Report no.: ESG89001
12NC: 9398 067 20011
420
Preface Power Semiconductor Applications
Philips Semiconductors
Acknowledgments
We are grateful for all the contributions from our colleagues within Philips and to the Application Laboratories in Eindhoven
and Hamburg.
We would also like to thank Dr.P.H.Mellor of the University of Sheffield for contributing the application note of section 3.1.5.
The authors thank Mrs.R.Hayes for her considerable help in the preparation of this book.
The authors also thank Mr.D.F.Haslam for his assistance in the formatting and printing of the manuscripts.
Contributing Authors
N.Bennett
M.Bennion
D.Brown
C.Buethker
L.Burley
G.M.Fry
R.P.Gant
J.Gilliam
D.Grant
N.J.Ham
C.J.Hammerton
D.J.Harper
W.Hettersheid
J.v.d.Hooff
J.Houldsworth
M.J.Humphreys
P.H.Mellor
R.Miller
H.Misdom
P.Moody
S.A.Mulder
E.B.G. Nijhof
J.Oosterling
N.Pichowicz
W.B.Rosink
D.C. de Ruiter
D.Sharples
H.Simons
T.Stork
D.Tebb
H.Verhees
F.A.Woodworth
T.van de Wouw
This book was originally prepared by the Power Semiconductor Applications Laboratory, of the Philips Semiconductors
product division, Hazel Grove:
M.J.Humphreys
C.J.Hammerton
D.Brown
R.Miller
L.Burley
It was revised and updated, in 1994, by:
N.J.Ham C.J.Hammerton D.Sharples
Preface Power Semiconductor Applications
Philips Semiconductors
Preface
This book was prepared by the Power Semiconductor Applications Laboratory of the Philips Semiconductors product
division, Hazel Grove. The book is intended as a guide to usingpower semiconductors both efficiently and reliably in power
conversion applications. It is made up of eight main chapters each of which contains a number of application notes aimed
at making it easier to select and use power semiconductors.
CHAPTER 1 forms an introduction to power semiconductors concentrating particularly on the two major power transistor
technologies, Power MOSFETs and High Voltage Bipolar Transistors.
CHAPTER 2 is devoted to Switched Mode Power Supplies. It begins with a basic description of the most commonly used
topologies and discusses the major issues surrounding the use of power semiconductors including rectifiers. Specific
design examples are given as well as a look at designing the magnetic components. The end of this chapter describes
resonant power supply technology.
CHAPTER 3 describes motion control in terms of ac, dc and stepper motor operation and control. This chapter looks only
at transistor controls, phase control using thyristors and triacs is discussed separately in chapter 6.
CHAPTER 4 looks at television and monitor applications. A description of the operation of horizontal deflection circuits is
givenfollowed bytransistorselection guidesfor both deflectionand power supplyapplications. Deflection andpower supply
circuitexamplesarealsogivenbasedoncircuitsdesignedbytheProductConceptandApplicationLaboratories(Eindhoven).
CHAPTER 5 concentrates on automotive electronics looking in detail at the requirementsfor the electronic switches taking
into consideration the harsh environment in which they must operate.
CHAPTER6reviewsthyristorandtriac applicationsfromthebasicsofdevice technologyand operationtothesimpledesign
rules which should be followed to achieve maximum reliability. Specific examples are given in this chapter for a number
of the common applications.
CHAPTER 7 looks at the thermal considerations for power semiconductors in terms of power dissipation and junction
temperaturelimits.Partofthischapterisdevotedto workedexamplesshowinghowjunctiontemperaturescan becalculated
to ensure the limits are not exceeded. Heatsink requirements and designs are also discussed in the second half of this
chapter.
CHAPTER 8 is an introduction to the use of high voltage bipolar transistors in electronic lighting ballasts. Many of the
possible topologies are described.
Contents Power Semiconductor Applications
Philips Semiconductors
Table of Contents
CHAPTER 1 Introduction to Power Semiconductors 1
General
3
1.1.1 An Introduction To Power Devices ............................................................ 5
Power MOSFET
17
1.2.1 PowerMOS Introduction ............................................................................. 19
1.2.2 Understanding Power MOSFET Switching Behaviour ............................... 29
1.2.3 Power MOSFET Drive Circuits .................................................................. 39
1.2.4 Parallel Operation of Power MOSFETs ..................................................... 49
1.2.5 Series Operation of Power MOSFETs ....................................................... 53
1.2.6 Logic Level FETS ...................................................................................... 57
1.2.7 Avalanche Ruggedness ............................................................................. 61
1.2.8 Electrostatic Discharge (ESD) Considerations .......................................... 67
1.2.9 Understanding the Data Sheet: PowerMOS .............................................. 69
High Voltage Bipolar Transistor
77
1.3.1 Introduction To High Voltage Bipolar Transistors ...................................... 79
1.3.2 Effects of Base Drive on Switching Times ................................................. 83
1.3.3 Using High Voltage Bipolar Transistors ..................................................... 91
1.3.4 Understanding The Data Sheet: High Voltage Transistors ....................... 97
CHAPTER 2 Switched Mode Power Supplies 103
Using Power Semiconductors in Switched Mode Topologies
105
2.1.1 An Introduction to Switched Mode Power Supply Topologies ................... 107
2.1.2 The Power Supply Designer’s Guide to High Voltage Transistors ............ 129
2.1.3 Base Circuit Design for High Voltage Bipolar Transistors in Power
Converters ........................................................................................................... 141
2.1.4 Isolated Power Semiconductors for High Frequency Power Supply
Applications ......................................................................................................... 153
Output Rectification
159
2.2.1 Fast Recovery Epitaxial Diodes for use in High Frequency Rectification 161
2.2.2 Schottky Diodes from Philips Semiconductors .......................................... 173
2.2.3 An Introduction to Synchronous Rectifier Circuits using PowerMOS
Transistors ........................................................................................................... 179
i
Contents Power Semiconductor Applications
Philips Semiconductors
Design Examples
185
2.3.1 Mains Input 100 W Forward Converter SMPS: MOSFET and Bipolar
Transistor Solutions featuring ETD Cores ...........................................................187
2.3.2 Flexible, Low Cost, Self-Oscillating Power Supply using an ETD34
Two-Part Coil Former and 3C85 Ferrite ..............................................................199
Magnetics Design
205
2.4.1 Improved Ferrite Materials and Core Outlines for High Frequency Power
Supplies ...............................................................................................................207
Resonant Power Supplies
217
2.5.1. An Introduction To Resonant Power Supplies ..........................................219
2.5.2. Resonant Power Supply Converters - The Solution For Mains Pollution
Problems ..............................................................................................................225
CHAPTER 3 Motor Control 241
AC Motor Control
243
3.1.1 Noiseless A.C. Motor Control: Introduction to a 20 kHz System ...............245
3.1.2 The Effect of a MOSFET’s Peak to Average Current Rating on Invertor
Efficiency .............................................................................................................251
3.1.3 MOSFETs and FREDFETs for Motor Drive Equipment .............................253
3.1.4 A Designers Guide to PowerMOS Devices for Motor Control ...................259
3.1.5 A 300V, 40A High Frequency Inverter Pole Using Paralleled FREDFET
Modules ...............................................................................................................273
DC Motor Control
283
3.2.1 Chopper circuits for DC motor control .......................................................285
3.2.2 A switched-mode controller for DC motors ................................................293
3.2.3 Brushless DC Motor Systems ....................................................................301
Stepper Motor Control
307
3.3.1 Stepper Motor Control ...............................................................................309
CHAPTER 4 Televisions and Monitors 317
Power Devices in TV & Monitor Applications (including selection
guides)
319
4.1.1 An Introduction to Horizontal Deflection ....................................................321
4.1.2 The BU25XXA/D Range of Deflection Transistors ....................................331
ii
Contents Power Semiconductor Applications
Philips Semiconductors
4.1.3 Philips HVT’s for TV & Monitor Applications .............................................. 339
4.1.4 TV and Monitor Damper Diodes ................................................................ 345
TV Deflection Circuit Examples
349
4.2.1 Application Information for the 16 kHz Black Line Picture Tubes .............. 351
4.2.2 32 kHz / 100 Hz Deflection Circuits for the 66FS Black Line Picture Tube 361
SMPS Circuit Examples
377
4.3.1 A 70W Full Performance TV SMPS Using The TDA8380 ......................... 379
4.3.2 A Synchronous 200W SMPS for 16 and 32 kHz TV .................................. 389
Monitor Deflection Circuit Example
397
4.4.1 A Versatile 30 - 64 kHz Autosync Monitor ................................................. 399
CHAPTER 5 Automotive Power Electronics 421
Automotive Motor Control (including selection guides)
423
5.1.1 Automotive Motor Control With Philips MOSFETS .................................... 425
Automotive Lamp Control (including selection guides)
433
5.2.1 Automotive Lamp Control With Philips MOSFETS .................................... 435
The TOPFET
443
5.3.1 An Introduction to the 3 pin TOPFET ......................................................... 445
5.3.2 An Introduction to the 5 pin TOPFET ......................................................... 447
5.3.3 BUK101-50DL - a Microcontroller compatible TOPFET ............................ 449
5.3.4 Protection with 5 pin TOPFETs ................................................................. 451
5.3.5 Driving TOPFETs ....................................................................................... 453
5.3.6 High Side PWM Lamp Dimmer using TOPFET ......................................... 455
5.3.7 Linear Control with TOPFET ...................................................................... 457
5.3.8 PWM Control with TOPFET ....................................................................... 459
5.3.9 Isolated Drive for TOPFET ........................................................................ 461
5.3.10 3 pin and 5 pin TOPFET Leadforms ........................................................ 463
5.3.11 TOPFET Input Voltage ............................................................................ 465
5.3.12 Negative Input and TOPFET ................................................................... 467
5.3.13 Switching Inductive Loads with TOPFET ................................................. 469
5.3.14 Driving DC Motors with TOPFET ............................................................. 471
5.3.15 An Introduction to the High Side TOPFET ............................................... 473
5.3.16 High Side Linear Drive with TOPFET ...................................................... 475
iii
Contents Power Semiconductor Applications
Philips Semiconductors
Automotive Ignition
477
5.4.1 An Introduction to Electronic Automotive Ignition ...................................... 479
5.4.2 IGBTs for Automotive Ignition .................................................................... 481
5.4.3 Electronic Switches for Automotive Ignition ............................................... 483
CHAPTER 6 Power Control with Thyristors and Triacs 485
Using Thyristors and Triacs
487
6.1.1 Introduction to Thyristors and Triacs ......................................................... 489
6.1.2 Using Thyristors and Triacs ....................................................................... 497
6.1.3 The Peak Current Handling Capability of Thyristors .................................. 505
6.1.4 Understanding Thyristor and Triac Data .................................................... 509
Thyristor and Triac Applications
521
6.2.1 Triac Control of DC Inductive Loads .......................................................... 523
6.2.2 Domestic Power Control with Triacs and Thyristors .................................. 527
6.2.3 Design of a Time Proportional Temperature Controller ............................. 537
Hi-Com Triacs
547
6.3.1 Understanding Hi-Com Triacs ................................................................... 549
6.3.2 Using Hi-Com Triacs .................................................................................. 551
CHAPTER 7 Thermal Management 553
Thermal Considerations
555
7.1.1 Thermal Considerations for Power Semiconductors ................................. 557
7.1.2 Heat Dissipation ......................................................................................... 567
CHAPTER 8 Lighting 575
Fluorescent Lamp Control
577
8.1.1 Efficient Fluorescent Lighting using Electronic Ballasts ............................. 579
8.1.2 Electronic Ballasts - Philips Transistor Selection Guide ............................ 587
8.1.3 An Electronic Ballast - Base Drive Optimisation ........................................ 589
iv
Index Power Semiconductor Applications
Philips Semiconductors
Index
Airgap, transformer core, 111, 113
Anti saturation diode, 590
Asynchronous, 497
Automotive
fans
see motor control
IGBT, 481, 483
ignition, 479, 481, 483
lamps, 435, 455
motor control, 425, 457, 459, 471, 475
resistive loads, 442
reverse battery, 452, 473, 479
screen heater, 442
seat heater, 442
solenoids, 469
TOPFET, 473
Avalanche, 61
Avalanche breakdown
thyristor, 490
Avalanche multiplication, 134
Baker clamp, 138, 187, 190
Ballast
electronic, 580
fluorescent lamp, 579
switchstart, 579
Base drive, 136
base inductor, 147
base inductor, diode assisted, 148
base resistor, 146
drive transformer, 145
drive transformer leakage inductance, 149
electronic ballast, 589
forward converter, 187
power converters, 141
speed-up capacitor, 143
Base inductor, 144, 147
Base inductor, diode assisted, 148
Boost converter, 109
continuous mode, 109
discontinuous mode, 109
output ripple, 109
Bootstrap, 303
Breakback voltage
diac, 492
Breakdown voltage, 70
Breakover current
diac, 492
Breakover voltage
diac, 492, 592
thyristor, 490
Bridge circuits
see Motor Control - AC
Brushless motor, 301, 303
Buck-boost converter, 110
Buck converter, 108 - 109
Burst firing, 537
Burst pulses, 564
Capacitance
junction, 29
Capacitor
mains dropper, 544
CENELEC, 537
Charge carriers, 133
triac commutation, 549
Choke
fluorescent lamp, 580
Choppers, 285
Clamp diode, 117
Clamp winding, 113
Commutation
diode, 164
Hi-Com triac, 551
thyristor, 492
triac, 494, 523, 529
Compact fluorescent lamp, 585
Continuous mode
see Switched Mode Power Supplies
Continuous operation, 557
Converter (dc-dc)
switched mode power supply, 107
Cookers, 537
Cooling
forced, 572
natural, 570
Crest factor, 529
Critical electric field, 134
Cross regulation, 114, 117
Current fed resonant inverter, 589
Current Mode Control, 120
Current tail, 138, 143
Damper Diodes, 345, 367
forward recovery, 328, 348
losses, 347
outlines, 345
picture distortion, 328, 348
selection guide, 345
Darlington, 13
Data Sheets
High Voltage Bipolar Transistor, 92,97,331
MOSFET, 69
i
Index Power Semiconductor Applications
Philips Semiconductors
dc-dc converter, 119
Depletion region, 133
Desaturation networks, 86
Baker clamp, 91, 138
dI/dt
triac, 531
Diac, 492, 500, 527, 530, 591
Diode, 6
double diffused, 162
epitaxial, 161
schottky, 173
structure, 161
Diode Modulator, 327, 367
Disc drives, 302
Discontinuous mode
see Switched Mode Power Supplies
Domestic Appliances, 527
Dropper
capacitive, 544
resistive, 544, 545
Duty cycle, 561
EFD core
see magnetics
Efficiency Diodes
see Damper Diodes
Electric drill, 531
Electronic ballast, 580
base drive optimisation, 589
current fed half bridge, 584, 587, 589
current fed push pull, 583, 587
flyback, 582
transistor selection guide, 587
voltage fed half bridge, 584, 588
voltage fed push pull, 583, 587
EMC, 260, 455
see RFI, ESD
TOPFET, 473
Emitter shorting
triac, 549
Epitaxial diode, 161
characteristics, 163
dI/dt, 164
forward recovery, 168
lifetime control, 162
operating frequency, 165
passivation, 162
reverse leakage, 169
reverse recovery, 162, 164
reverse recovery softness, 167
selection guide, 171
snap-off, 167
softness factor, 167
stored charge, 162
technology, 162
ESD, 67
see Protection, ESD
precautions, 67
ETD core
see magnetics
F-pack
see isolated package
Fall time, 143, 144
Fast Recovery Epitaxial Diode (FRED)
see epitaxial diode
FBSOA, 134
Ferrites
see magnetics
Flicker
fluorescent lamp, 580
Fluorescent lamp, 579
colour rendering, 579
colour temperature, 579
efficacy, 579, 580
triphosphor, 579
Flyback converter, 110, 111, 113
advantages, 114
clamp winding, 113
continuous mode, 114
coupled inductor, 113
cross regulation, 114
diodes, 115
disadvantages, 114
discontinuous mode, 114
electronic ballast, 582
leakage inductance, 113
magnetics, 213
operation, 113
rectifier circuit, 180
self oscillating power supply, 199
synchronous rectifier, 156, 181
transformer core airgap, 111, 113
transistors, 115
Flyback converter (two transistor), 111, 114
Food mixer, 531
Forward converter, 111, 116
advantages, 116
clamp diode, 117
conduction loss, 197
continuous mode, 116
core loss, 116
core saturation, 117
cross regulation, 117
diodes, 118
disadvantages, 117
duty ratio, 117
ferrite cores, 116
magnetics, 213
magnetisation energy, 116, 117
ii
Index Power Semiconductor Applications
Philips Semiconductors
operation, 116
output diodes, 117
output ripple, 116
rectifier circuit, 180
reset winding, 117
switched mode power supply, 187
switching frequency, 195
switching losses, 196
synchronous rectifier, 157, 181
transistors, 118
Forward converter (two transistor), 111, 117
Forward recovery, 168
FREDFET, 250, 253, 305
bridge circuit, 255
charge, 254
diode, 254
drive, 262
loss, 256
reverse recovery, 254
FREDFETs
motor control, 259
Full bridge converter, 111, 125
advantages, 125
diodes, 126
disadvantages, 125
operation, 125
transistors, 126
Gate
triac, 538
Gate drive
forward converter, 195
Gold doping, 162, 169
GTO, 11
Guard ring
schottky diode, 174
Half bridge, 253
Half bridge circuits
see also Motor Control - AC
Half bridge converter, 111, 122
advantages, 122
clamp diodes, 122
cross conduction, 122
diodes, 124
disadvantages, 122
electronic ballast, 584, 587, 589
flux symmetry, 122
magnetics, 214
operation, 122
synchronous rectifier, 157
transistor voltage, 122
transistors, 124
voltage doubling, 122
Heat dissipation, 567
Heat sink compound, 567
Heater controller, 544
Heaters, 537
Heatsink, 569
Heatsink compound, 514
Hi-Com triac, 519, 549, 551
commutation, 551
dIcom/dt, 552
gate trigger current, 552
inductive load control, 551
High side switch
MOSFET, 44, 436
TOPFET, 430, 473
High Voltage Bipolar Transistor, 8, 79, 91,
141, 341
‘bathtub’ curves, 333
avalanche breakdown, 131
avalanche multiplication, 134
Baker clamp, 91, 138
base-emitter breakdown, 144
base drive, 83, 92, 96, 136, 336, 385
base drive circuit, 145
base inductor, 138, 144, 147
base inductor, diode assisted, 148
base resistor, 146
breakdown voltage, 79, 86, 92
carrier concentration, 151
carrier injection, 150
conductivity modulation, 135, 150
critical electric field, 134
current crowding, 135, 136
current limiting values, 132
current tail, 138, 143
current tails, 86, 91
d-type, 346
data sheet, 92, 97, 331
depletion region, 133
desaturation, 86, 88, 91
device construction, 79
dI/dt, 139
drive transformer, 145
drive transformer leakage inductance, 149
dV/dt, 139
electric field, 133
electronic ballast, 581, 585, 587, 589
Fact Sheets, 334
fall time, 86, 99, 143, 144
FBSOA, 92, 99, 134
hard turn-off, 86
horizontal deflection, 321, 331, 341
leakage current, 98
limiting values, 97
losses, 92, 333, 342
Miller capacitance, 139
operation, 150
iii
Index Power Semiconductor Applications
Philips Semiconductors
optimum drive, 88
outlines, 332, 346
over current, 92, 98
over voltage, 92, 97
overdrive, 85, 88, 137, 138
passivation, 131
power limiting value, 132
process technology, 80
ratings, 97
RBSOA, 93, 99, 135, 138, 139
RC network, 148
reverse recovery, 143, 151
safe operating area, 99, 134
saturation, 150
saturation current, 79, 98, 341
secondary breakdown, 92, 133
smooth turn-off, 86
SMPS, 94, 339, 383
snubber, 139
space charge, 133
speed-up capacitor, 143
storage time, 86, 91, 92, 99, 138, 144, 342
sub emitter resistance, 135
switching, 80, 83, 86, 91, 98, 342
technology, 129, 149
thermal breakdown, 134
thermal runaway, 152
turn-off, 91, 92, 138, 142, 146, 151
turn-on, 91, 136, 141, 149, 150
underdrive, 85, 88
voltage limiting values, 130
Horizontal Deflection, 321, 367
base drive, 336
control ic, 401
d-type transistors, 346
damper diodes, 345, 367
diode modulator, 327, 347, 352, 367
drive circuit, 352, 365, 406
east-west correction, 325, 352, 367
line output transformer, 354
linearity correction, 323
operating cycle, 321, 332, 347
s-correction, 323, 352, 404
TDA2595, 364, 368
TDA4851, 400
TDA8433, 363, 369
test circuit, 321
transistors, 331, 341, 408
waveforms, 322
IGBT, 11, 305
automotive, 481, 483
clamped, 482, 484
ignition, 481, 483
Ignition
automotive, 479, 481, 483
darlington, 483
Induction heating, 53
Induction motor
see Motor Control - AC
Inductive load
see Solenoid
Inrush current, 528, 530
Intrinsic silicon, 133
Inverter, 260, 273
see motor control ac
current fed, 52, 53
switched mode power supply, 107
Irons, electric, 537
Isolated package, 154
stray capacitance, 154, 155
thermal resistance, 154
Isolation, 153
J-FET, 9
Junction temperature, 470, 557, 561
burst pulses, 564
non-rectangular pulse, 565
rectangular pulse, composite, 562
rectangular pulse, periodic, 561
rectangular pulse, single shot, 561
Lamp dimmer, 530
Lamps, 435
dI/dt, 438
inrush current, 438
MOSFET, 435
PWM control, 455
switch rate, 438
TOPFET, 455
Latching current
thyristor, 490
Leakage inductance, 113, 200, 523
Lifetime control, 162
Lighting
fluorescent, 579
phase control, 530
Logic Level FET
motor control, 432
Logic level MOSFET, 436
Magnetics, 207
100W 100kHz forward converter, 197
100W 50kHz forward converter, 191
50W flyback converter, 199
core losses, 208
core materials, 207
EFD core, 210
ETD core, 199, 207
iv
Index Power Semiconductor Applications
Philips Semiconductors
flyback converter, 213
forward converter, 213
half bridge converter, 214
power density, 211
push-pull converter, 213
switched mode power supply, 187
switching frequency, 215
transformer construction, 215
Mains Flicker, 537
Mains pollution, 225
pre-converter, 225
Mains transient, 544
Mesa glass, 162
Metal Oxide Varistor (MOV), 503
Miller capacitance, 139
Modelling, 236, 265
MOS Controlled Thyristor, 13
MOSFET, 9, 19, 153, 253
bootstrap, 303
breakdown voltage, 22, 70
capacitance, 30, 57, 72, 155, 156
capacitances, 24
characteristics, 23, 70 - 72
charge, 32, 57
data sheet, 69
dI/dt, 36
diode, 253
drive, 262, 264
drive circuit loss, 156
driving, 39, 250
dV/dt, 36, 39, 264
ESD, 67
gate-source protection, 264
gate charge, 195
gate drive, 195
gate resistor, 156
high side, 436
high side drive, 44
inductive load, 62
lamps, 435
leakage current, 71
linear mode, parallelling, 52
logic level, 37, 57, 305
loss, 26, 34
maximum current, 69
motor control, 259, 429
modelling, 265
on-resistance, 21, 71
package inductance, 49, 73
parallel operation, 26, 47, 49, 265
parasitic oscillations, 51
peak current rating, 251
Resonant supply, 53
reverse diode, 73
ruggedness, 61, 73
safe operating area, 25, 74
series operation, 53
SMPS, 339, 384
solenoid, 62
structure, 19
switching, 24, 29, 58, 73, 194, 262
switching loss, 196
synchronous rectifier, 179
thermal impedance, 74
thermal resistance, 70
threshold voltage, 21, 70
transconductance, 57, 72
turn-off, 34, 36
turn-on, 32, 34, 35, 155, 256
Motor, universal
back EMF, 531
starting, 528
Motor Control - AC, 245, 273
anti-parallel diode, 253
antiparallel diode, 250
carrier frequency, 245
control, 248
current rating, 262
dc link, 249
diode, 261
diode recovery, 250
duty ratio, 246
efficiency, 262
EMC, 260
filter, 250
FREDFET, 250, 259, 276
gate drives, 249
half bridge, 245
inverter, 250, 260, 273
line voltage, 262
loss, 267
MOSFET, 259
Parallel MOSFETs, 276
peak current, 251
phase voltage, 262
power factor, 262
pulse width modulation, 245, 260
ripple, 246
short circuit, 251
signal isolation, 250
snubber, 276
speed control, 248
switching frequency, 246
three phase bridge, 246
underlap, 248
Motor Control - DC, 285, 293, 425
braking, 285, 299
brushless, 301
control, 290, 295, 303
current rating, 288
v
Index Power Semiconductor Applications
Philips Semiconductors
drive, 303
duty cycle, 286
efficiency, 293
FREDFET, 287
freewheel diode, 286
full bridge, 287
half bridge, 287
high side switch, 429
IGBT, 305
inrush, 430
inverter, 302
linear, 457, 475
logic level FET, 432
loss, 288
MOSFET, 287, 429
motor current, 295
overload, 430
permanent magnet, 293, 301
permanent magnet motor, 285
PWM, 286, 293, 459, 471
servo, 298
short circuit, 431
stall, 431
TOPFET, 430, 457, 459, 475
topologies, 286
torque, 285, 294
triac, 525
voltage rating, 288
Motor Control - Stepper, 309
bipolar, 310
chopper, 314
drive, 313
hybrid, 312
permanent magnet, 309
reluctance, 311
step angle, 309
unipolar, 310
Mounting, transistor, 154
Mounting base temperature, 557
Mounting torque, 514
Parasitic oscillation, 149
Passivation, 131, 162
PCB Design, 368, 419
Phase angle, 500
Phase control, 546
thyristors and triacs, 498
triac, 523
Phase voltage
see motor control - ac
Power dissipation, 557
see High Voltage Bipolar Transistor loss,
MOSFET loss
Power factor correction, 580
active, boost converted, 581
Power MOSFET
see MOSFET
Proportional control, 537
Protection
ESD, 446, 448, 482
overvoltage, 446, 448, 469
reverse battery, 452, 473, 479
short circuit, 251, 446, 448
temperature, 446, 447, 471
TOPFET, 445, 447, 451
Pulse operation, 558
Pulse Width Modulation (PWM), 108
Push-pull converter, 111, 119
advantages, 119
clamp diodes, 119
cross conduction, 119
current mode control, 120
diodes, 121
disadvantages, 119
duty ratio, 119
electronic ballast, 582, 587
flux symmetry, 119, 120
magnetics, 213
multiple outputs, 119
operation, 119
output filter, 119
output ripple, 119
rectifier circuit, 180
switching frequency, 119
transformer, 119
transistor voltage, 119
transistors, 121
Qs (stored charge), 162
RBSOA, 93, 99, 135, 138, 139
Rectification, synchronous, 179
Reset winding, 117
Resistor
mains dropper, 544, 545
Resonant power supply, 219, 225
modelling, 236
MOSFET, 52, 53
pre-converter, 225
Reverse leakage, 169
Reverse recovery, 143, 162
RFI, 154, 158, 167, 393, 396, 497, 529, 530,
537
Ruggedness
MOSFET, 62, 73
schottky diode, 173
Safe Operating Area (SOA), 25, 74, 134, 557
forward biased, 92, 99, 134
reverse biased, 93, 99, 135, 138, 139
vi
Index Power Semiconductor Applications
Philips Semiconductors
Saturable choke
triac, 523
Schottky diode, 173
bulk leakage, 174
edge leakage, 174
guard ring, 174
reverse leakage, 174
ruggedness, 173
selection guide, 176
technology, 173
SCR
see Thyristor
Secondary breakdown, 133
Selection Guides
BU25XXA, 331
BU25XXD, 331
damper diodes, 345
EPI diodes, 171
horizontal deflection, 343
MOSFETs driving heaters, 442
MOSFETs driving lamps, 441
MOSFETs driving motors, 426
Schottky diodes, 176
SMPS, 339
Self Oscillating Power Supply (SOPS)
50W microcomputer flyback converter, 199
ETD transformer, 199
Servo, 298
Single ended push-pull
see half bridge converter
Snap-off, 167
Snubber, 93, 139, 495, 502, 523, 529, 549
active, 279
Softness factor, 167
Solenoid
TOPFET, 469, 473
turn off, 469, 473
Solid state relay, 501
SOT186, 154
SOT186A, 154
SOT199, 154
Space charge, 133
Speed-up capacitor, 143
Speed control
thyristor, 531
triac, 527
Starter
fluorescent lamp, 580
Startup circuit
electronic ballast, 591
self oscillating power supply, 201
Static Induction Thyristor, 11
Stepdown converter, 109
Stepper motor, 309
Stepup converter, 109
Storage time, 144
Stored charge, 162
Suppression
mains transient, 544
Switched Mode Power Supply (SMPS)
see also self oscillating power supply
100W 100kHz MOSFET forward converter,
192
100W 500kHz half bridge converter, 153
100W 50kHz bipolar forward converter, 187
16 & 32 kHz TV, 389
asymmetrical, 111, 113
base circuit design, 149
boost converter, 109
buck-boost converter, 110
buck converter, 108
ceramic output filter, 153
continuous mode, 109, 379
control ic, 391
control loop, 108
core excitation, 113
core loss, 167
current mode control, 120
dc-dc converter, 119
diode loss, 166
diode reverse recovery effects, 166
diode reverse recovery softness, 167
diodes, 115, 118, 121, 124, 126
discontinuous mode, 109, 379
epitaxial diodes, 112, 161
flux swing, 111
flyback converter, 92, 111, 113, 123
forward converter, 111, 116, 379
full bridge converter, 111, 125
half bridge converter, 111, 122
high voltage bipolar transistor, 94, 112, 115,
118, 121, 124, 126, 129, 339, 383, 392
isolated, 113
isolated packages, 153
isolation, 108, 111
magnetics design, 191, 197
magnetisation energy, 113
mains filter, 380
mains input, 390
MOSFET, 112, 153, 33, 384
multiple output, 111, 156
non-isolated, 108
opto-coupler, 392
output rectifiers, 163
parasitic oscillation, 149
power-down, 136
power-up, 136, 137, 139
power MOSFET, 153, 339, 384
pulse width modulation, 108
push-pull converter, 111, 119
vii
Index Power Semiconductor Applications
Philips Semiconductors
RBSOA failure, 139
rectification, 381, 392
rectification efficiency, 163
rectifier selection, 112
regulation, 108
reliability, 139
resonant
see resonant power supply
RFI, 154, 158, 167
schottky diode, 112, 154, 173
snubber, 93, 139, 383
soft start, 138
standby, 382
standby supply, 392
start-up, 391
stepdown, 109
stepup, 109
symmetrical, 111, 119, 122
synchronisation, 382
synchronous rectification, 156, 179
TDA8380, 381, 391
topologies, 107
topology output powers, 111
transformer, 111
transformer saturation, 138
transformers, 391
transistor current limiting value, 112
transistor mounting, 154
transistor selection, 112
transistor turn-off, 138
transistor turn-on, 136
transistor voltage limiting value, 112
transistors, 115, 118, 121, 124, 126
turns ratio, 111
TV & Monitors, 339, 379, 399
two transistor flyback, 111, 114
two transistor forward, 111, 117
Switching loss, 230
Synchronous, 497
Synchronous rectification, 156, 179
self driven, 181
transformer driven, 180
Temperature control, 537
Thermal
continuous operation, 557, 568
intermittent operation, 568
non-rectangular pulse, 565
pulse operation, 558
rectangular pulse, composite, 562
rectangular pulse, periodic, 561
rectangular pulse, single shot, 561
single shot operation, 561
Thermal capacity, 558, 568
Thermal characteristics
power semiconductors, 557
Thermal impedance, 74, 568
Thermal resistance, 70, 154, 557
Thermal time constant, 568
Thyristor, 10, 497, 509
’two transistor’ model, 490
applications, 527
asynchronous control, 497
avalanche breakdown, 490
breakover voltage, 490, 509
cascading, 501
commutation, 492
control, 497
current rating, 511
dI/dt, 490
dIf/dt, 491
dV/dt, 490
energy handling, 505
external commutation, 493
full wave control, 499
fusing I2t, 503, 512
gate cathode resistor, 500
gate circuits, 500
gate current, 490
gate power, 492
gate requirements, 492
gate specifications, 512
gate triggering, 490
half wave control, 499
holding current, 490, 509
inductive loads, 500
inrush current, 503
latching current, 490, 509
leakage current, 490
load line, 492
mounting, 514
operation, 490
overcurrent, 503
peak current, 505
phase angle, 500
phase control, 498, 527
pulsed gate, 500
resistive loads, 498
resonant circuit, 493
reverse characteristic, 489
reverse recovery, 493
RFI, 497
self commutation, 493
series choke, 502
snubber, 502
speed controller, 531
static switching, 497
structure, 489
switching, 489
viii
Index Power Semiconductor Applications
Philips Semiconductors
switching characteristics, 517
synchronous control, 497
temperature rating, 512
thermal specifications, 512
time proportional control, 497
transient protection, 502
trigger angle, 500
turn-off time, 494
turn-on, 490, 509
turn-on dI/dt, 502
varistor, 503
voltage rating, 510
Thyristor data, 509
Time proportional control, 537
TOPFET
3 pin, 445, 449, 461
5 pin, 447, 451, 457, 459, 463
driving, 449, 453, 461, 465, 467, 475
high side, 473, 475
lamps, 455
leadforms, 463
linear control, 451, 457
motor control, 430, 457, 459
negative input, 456, 465, 467
protection, 445, 447, 451, 469, 473
PWM control, 451, 455, 459
solenoids, 469
Transformer
triac controlled, 523
Transformer core airgap, 111, 113
Transformers
see magnetics
Transient thermal impedance, 559
Transient thermal response, 154
Triac, 497, 510, 518
400Hz operation, 489, 518
applications, 527, 537
asynchronous control, 497
breakover voltage, 510
charge carriers, 549
commutating dI/dt, 494
commutating dV/dt, 494
commutation, 494, 518, 523, 529, 549
control, 497
dc inductive load, 523
dc motor control, 525
dI/dt, 531, 549
dIcom/dt, 523
dV/dt, 523, 549
emitter shorting, 549
full wave control, 499
fusing I2t, 503, 512
gate cathode resistor, 500
gate circuits, 500
gate current, 491
gate requirements, 492
gate resistor, 540, 545
gate sensitivity, 491
gate triggering, 538
holding current, 491, 510
Hi-Com, 549, 551
inductive loads, 500
inrush current, 503
isolated trigger, 501
latching current, 491, 510
operation, 491
overcurrent, 503
phase angle, 500
phase control, 498, 527, 546
protection, 544
pulse triggering, 492
pulsed gate, 500
quadrants, 491, 510
resistive loads, 498
RFI, 497
saturable choke, 523
series choke, 502
snubber, 495, 502, 523, 529, 549
speed controller, 527
static switching, 497
structure, 489
switching, 489
synchronous control, 497
transformer load, 523
transient protection, 502
trigger angle, 492, 500
triggering, 550
turn-on dI/dt, 502
varistor, 503
zero crossing, 537
Trigger angle, 500
TV & Monitors
16 kHz black line, 351
30-64 kHz autosync, 399
32 kHz black line, 361
damper diodes, 345, 367
diode modulator, 327, 367
EHT, 352 - 354, 368, 409, 410
high voltage bipolar transistor, 339, 341
horizontal deflection, 341
picture distortion, 348
power MOSFET, 339
SMPS, 339, 354, 379, 389, 399
vertical deflection, 358, 364, 402
Two transistor flyback converter, 111, 114
Two transistor forward converter, 111, 117
Universal motor
back EMF, 531
ix
Index Power Semiconductor Applications
Philips Semiconductors
starting, 528
Vacuum cleaner, 527
Varistor, 503
Vertical Deflection, 358, 364, 402
Voltage doubling, 122
Water heaters, 537
Zero crossing, 537
Zero voltage switching, 537
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